LT1961 - Linear Technology - Farnell Element 14 - Revenir à l'accueil
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Sefram-SP270.pdf-PDF..> 29-Mar-2014 11:46 464K1 LT1961 1961fa FEATURES DESCRIPTIO U APPLICATIOU S TYPICAL APPLICATIO U 1.5A, 1.25MHz Step-Up Switching Regulator ■ 1.5A Switch in a Small MSOP Package ■ Constant 1.25MHz Switching Frequency ■ Wide Operating Voltage Range: 3V to 25V ■ High Efficiency 0.2Ω Switch ■ 1.2V Feedback Reference Voltage ■ ±2% Overall Output Voltage Tolerance ■ Uses Low Profile Surface Mount External Components ■ Low Shutdown Current: 6μA ■ Synchronizable from 1.5MHz to 2MHz ■ Current-Mode Loop Control ■ Constant Maximum Switch Current Rating at All Duty Cycles* ■ Thermally Enhanced Exposed Pad 8-Lead Plastic MSOP Package The LT®1961 is a 1.25MHz monolithic boost switching regulator. A high efficiency 1.5A, 0.2Ω switch is included on the die together with all the control circuitry required to complete a high frequency, current-mode switching regulator. Current-mode control provides fast transient response and excellent loop stability. New design techniques achieve high efficiency at high switching frequencies over a wide operating voltage range. A low dropout internal regulator maintains consistent performance over a wide range of inputs from 24V systems to Li-Ion batteries. An operating supply current of 1mA maintains high efficiency, especially at lower output currents. Shutdown reduces quiescent current to 6μA. Maximum switch current remains constant at all duty cycles. Synchronization allows an external logic level signal to increase the internal oscillator from 1.5MHz to 2MHz. The LT1961 is available in an exposed pad, 8-pin MSOP package. Full cycle-by-cycle switch current limit protection and thermal shutdown are provided. High frequency operation allows the reduction of input and output filtering components and permits the use of chip inductors. ■ DSL Modems ■ Portable Computers ■ Battery-Powered Systems ■ Distributed Power Efficiency vs Load Current 5V to 12V Boost Converter LT1961 VIN VOUT 12V 0.5A* VIN 5V 1961 TA01 6800pF 100pF 6.8k 10k 1% 90.9k UPS120 10μF CERAMIC 2.2μF CERAMIC VSW SHDN FB OPEN OR HIGH = ON SYNC GND VC *MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING. 6.8μH 2 6 8 3,4 7 5 1 LOAD CURRENT (mA) 0 EFFICIENCY (%) 90 85 80 75 70 65 60 100 200 300 400 1961 TA01a 500 VIN = 5V VOUT = 12V , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. *Patent Pending 2 LT1961 1961fa ABSOLUTE MAXIMUM RATINGS W W W U Input Voltage .......................................................... 25V Switch Voltage ......................................................... 35V SHDN Pin ............................................................... 25V FB Pin Current ....................................................... 1mA SYNC Pin Current .................................................. 1mA Operating Junction Temperature Range (Note 2) LT1961E, LT1961I ........................... – 40°C to 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C (Note 1) TJMAX = 125°C, θJA = 50°C/W GROUND PAD CONNECTED TO LARGE COPPER AREA 1234 VIN SW GND GND 8765 SYNC VC FB SHDN TOP VIEW MS8E PACKAGE 8-LEAD PLASTIC MSOP PI CO FIGURATIOU U U PARAMETER CONDITION MIN TYP MAX UNITS Recommended Operating Voltage ● 3 25 V Maximum Switch Current Limit ● 1.5 2 3 A Oscillator Frequency 3.3V < VIN < 25V ● 1 1.5 MHz Switch On Voltage Drop ISW = 1.5A ● 310 500 mV VIN Undervoltage Lockout (Note 3) ● 2.47 2.6 2.73 V VIN Supply Current ISW = 0A ● 0.9 1.3 mA VIN Supply Current/ISW ISW = 1.5A 27 mA/A Shutdown Supply Current VSHDN = 0V, VIN = 25V, VSW = 25V 6 20 μA ● 45 μA Feedback Voltage 3V < VIN < 25V, 0.4V < VC < 0.9V 1.182 1.2 1.218 V ● 1.176 1.224 V ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VC = 0.8V, SHDN, SYNC and switch open unless otherwise noted. LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT1961EMS8E#PBF LT1961EMS8E#TRPBF LTQY 8-Lead Plastic MSOP –40°C to 125°C LT1961IMS8E#PBF LT1961IMS8E#TRPBF LTQY 8-Lead Plastic MSOP –40°C to 125°C LEAD BASED FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT1961EMS8E LT1961EMS8E#TR LTQY 8-Lead Plastic MSOP –40°C to 125°C LT1961IMS8E LT1961IMS8E#TR LTQY 8-Lead Plastic MSOP –40°C to 125°C ORDER IUFORWATIOU Consult LTC Marketing for parts specified with wider operating temperature ranges. *Temperature grades are identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3 LT1961 1961fa ELECTRICAL CHARACTERISTICS PARAMETER CONDITION MIN TYP MAX UNITS FB Input Current ● 0 –0.2 –0.4 μA FB to VC Voltage Gain 0.4V < VC < 0.9V 150 350 FB to VC Transconductance ΔIVC = ±10μA ● 500 850 1300 μMho VC Pin Source Current VFB = 1V ● – 85 –120 –165 μA VC Pin Sink Current VFB = 1.4V ● 70 110 165 μA VC Pin to Switch Current Transconductance 2.4 A/V VC Pin Minimum Switching Threshold Duty Cycle = 0% 0.3 V VC Pin 1.5A ISW Threshold 0.9 V Maximum Switch Duty Cycle VC = 1.2V, ISW = 100mA ● 80 90 % VC = 1.2V, ISW = 1A, 25°C ≤ TA ≤ 125°C 75 80 % VC = 1.2V, ISW = 1A, TA ≤ 25°C 70 75 % SHDN Threshold Voltage ● 1.28 1.35 1.42 V SHDN Input Current (Shutting Down) SHDN = 60mV Above Threshold ● –7 –10 –13 μA SHDN Threshold Current Hysteresis SHDN = 100mV Below Threshold 4 7 10 μA SYNC Threshold Voltage 1.5 2.2 V SYNC Input Frequency 1.5 2 MHz SYNC Pin Resistance ISYNC = 1mA 20 kΩ The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VC = 0.8V, SHDN, SYNC and switch open unless otherwise noted. Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT1961E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT1961I is guaranteed over the – 40ºC to 125ºC operating junction temperature range. Note 3: Minimum input voltage is defined as the voltage where the internal regulator enters lockout. Actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. See Applications Information. 4 LT1961 1961fa TYPICAL PERFORMANCE CHARACTERISTICS U W FB vs Temperature Switch On Voltage Drop Oscillator Frequency SHDN Threshold vs Temperature SHDN Supply Current vs VIN SHDN IP Current vs Temperature TEMPERATURE (°C) –50 –25 0 25 50 75 100 125 FB VOLTAGE (V) 1961 G01 1.22 1.21 1.20 1.19 1.18 SWITCH CURRENT (A) 0 0.5 1 1.5 SWITCH VOLTAGE (mV) 1961 G02 400 350 300 250 200 150 100 50 0 125°C 25°C –40°C TEMPERATURE (°C) –50 –25 0 25 50 75 100 125 FREQUENCY (MHz) 1961 G03 1.5 1.4 1.3 1.2 1.1 TA = 25°C TEMPERATURE (°C) –50 –25 0 25 50 75 100 125 SHDN THRESHOLD (V) 1961 G04 1.40 1.38 1.36 1.34 1.32 1.30 VIN (V) 0 5 10 15 20 25 30 VIN CURRENT (μA) 1961 G05 7 6 5 4 3 2 1 0 TA = 25°C SHDN = 0V TEMPERATURE (°C) –50 –25 0 25 50 75 100 125 SHDN INPUT (μA) 1961G06 –12 –10 –8 –6 –4 –2 0 STARTING UP SHUTTING DOWN SHDN Supply Current Input Supply Current Current Limit Foldback SHUTDOWN VOLTAGE (V) 0 0.2 0.4 0.6 0.8 1 1.2 1.4 VIN CURRENT (μA) 1961 G07 300 250 200 150 100 50 0 TA = 25°C VIN = 15V INPUT VOLTAGE (V) 0 5 10 15 20 25 30 VIN CURRENT (μA) 1961 G08 1200 1000 800 600 400 200 0 MINIMUM INPUT VOLTAGE TA = 25°C FEEDBACK VOLTAGE (V) 0 0.2 0.4 0.6 0.8 1 1.2 SWITCH PEAK CURRENT (A) 1961 G09 2.0 1.5 1.0 0.5 0 FB INPUT CURRENT (μA) 40 30 20 10 0 FB CURRENT SWITCH CURRENT TA = 25°C 5 LT1961 1961fa FB: The feedback pin is used to set output voltage using an external voltage divider that generates 1.2V at the pin with the desired output voltage. If required, the current limit can be reduced during start up when the FB pin is below 0.5V (see the Current Limit Foldback graph in the Typical Performance Characteristics section). An impedance of less than 5kΩ at the FB pin is needed for this feature to operate. VIN: This pin powers the internal circuitry and internal regulator. Keep the external bypass capacitor close to this pin. GND: Short GND pins 3 and 4 and the exposed pad on the PCB. The GND is the reference for the regulated output, so load regulation will suffer if the “ground” end of the load is not at the same voltage as the GND of the IC. This condition occurs when the load current flows through the metal path between the GND pins and the load ground point. Keep the ground path short between the GND pins and the load and use a ground plane when possible. Keep the path between the input bypass and the GND pins short. The exposed pad should be attached to a large copper area to improve thermal resistance. VSW: The switch pin is the collector of the on-chip power NPN switch and has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. SYNC: The sync pin is used to synchronize the internal oscillator to an external signal. It is directly logic compatible and can be driven with any signal between 20% and 80% duty cycle. The synchronizing range is equal to initial operating frequency, up to 2MHz. See Synchronization section in Applications Information for details. When not in use, this pin should be grounded. SHDN: The shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. The 1.35V threshold can function as an accurate undervoltage lockout (UVLO), preventing the regulator from operating until the input voltage has reached a predetermined level. Float or pull high to put the regulator in the operating mode. VC: The VC pin is the output of the error amplifier and the input of the peak switch current comparator. It is normally used for frequency compensation, but can do double duty as a current clamp or control loop override. This pin sits at about 0.3V for very light loads and 0.9V at maximum load. PIN FUNCTIONSU U U 6 LT1961 1961fa amplifier commands current to be delivered to the output rather than voltage. A voltage fed system will have low phase shift up to the resonant frequency of the inductor and output capacitor, then an abrupt 180° shift will occur. The current fed system will have 90° phase shift at a much lower frequency, but will not have the additional 90° shift until well beyond the LC resonant frequency. This makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. A comparator connected to the shutdown pin disables the internal regulator, reducing supply current. The LT1961 is a constant frequency, current-mode boost converter. This means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. In addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. A switch cycle starts with an oscillator pulse which sets the RS flip-flop to turn the switch on. When switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. Output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. This technique means that the error Figure 1. Block Diagram BLOCK DIAGRAMW – + – + Σ VIN 2.5V BIAS REGULATOR 1.25MHz OSCILLATOR SW FB VC GND GND 1767 F01 SLOPE COMP 0.01Ω INTERNAL VCC CURRENT SENSE AMPLIFIER VOLTAGE GAIN = 40 SYNC SHDN SHUTDOWN COMPARATOR CURRENT COMPARATOR ERROR AMPLIFIER gm = 850μMho RS FLIP-FLOP DRIVER CIRCUITRY S R 0.3V Q1 POWER SWITCH 1.2V – + + – 1.35V 3μA 7μA 1 8 5 7 6 3 4 2 7 LT1961 1961fa APPLICATIONS INFORMATION W U U U FB RESISTOR NETWORK The suggested resistance (R2) from FB to ground is 10k 1%. This reduces the contribution of FB input bias current to output voltage to less than 0.2%. The formula for the resistor (R1) from VOUT to FB is: R R V R A OUT 1 2 12 1 2 2 0 2 = ( − ) − μ . . (. ) defines the pole frequency of the output stage, an X7R or X5R type ceramic, which have good temperature stability, is recommended. Tantalum capacitors are usually chosen for their bulk capacitance properties, useful in high transient load applications. ESR rather than absolute value defines output ripple at 1.25MHz. Values in the 22μF to 100μF range are generally needed to minimize ESR and meet ripple current ratings. Care should be taken to ensure the ripple ratings are not exceeded. Table 1. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current E Case Size ESR (Max, Ω) Ripple Current (A) AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1 AVX TAJ 0.7 to 0.9 0.4 D Case Size AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1 C Case Size AVX TPS 0.2 (typ) 0.5 (typ) INPUT CAPACITOR Unlike the output capacitor, RMS ripple current in the input capacitor is normally low enough that ripple current rating is not an issue. The current waveform is triangular, with an RMS value given by: I V V V L f V RIPPLE RMS IN OUT IN OUT ( )= ( )( − ) ( )( )( ) 0.29 At higher switching frequency, the energy storage requirement of the input capacitor is reduced so values in the range of 1μF to 4.7μF are suitable for most applications. Y5V or similar type ceramics can be used since the absolute value of capacitance is less important and has no significant effect on loop stability. If operation is required close to the minimum input voltage required by either the output or the LT1961, a larger value may be necessary. This is to prevent excessive ripple causing dips below the minimum operating voltage resulting in erratic operation. Figure 2. Feedback Network OUTPUT CAPACITOR Step-up regulators supply current to the output in pulses. The rise and fall times of these pulses are very fast. The output capacitor is required to reduce the voltage ripple this causes. The RMS ripple current can be calculated from: IRIPPLE(RMS) =IOUT (VOUT − VIN) / VIN The LT1961 will operate with both ceramic and tantalum output capacitors. Ceramic capacitors are generally chosen for their small size, very low ESR (effective series resistance), and good high frequency operation, reducing output ripple voltage. Their low ESR removes a useful zero in the loop frequency response, common to tantalum capacitors. To compensate for this, the VC loop compensation pole frequency must typically be reduced by a factor of 10. Typical ceramic output capacitors are in the 1μF to 10μF range. Since the absolute value of capacitance – + 1.2V VSW VC GND 1961 F02 R1 R2 10k OUTPUT ERROR AMPLIFIER FB LT1961 + 8 LT1961 1961fa APPLICATIONS INFORMATION W U U U INDUCTOR CHOICE AND MAXIMUM OUTPUT CURRENT When choosing an inductor, there are 2 conditions that limit the minimum inductance; required output current, and avoidance of subharmonic oscillation. The maximum output current for the LT1961 in a standard boost converter configuration with an infinitely large inductor is: I A V V OUT MAX IN OUT ( ) . • = 1 5 η Where η = converter efficiency (typically 0.87 at high current). As the value of inductance is reduced, ripple current increases and IOUT(MAX) is reduced. The minimum inductance for a required output current is given by: L V V V V f V I V MIN IN OUT IN OUT OUT OUT IN = ⎛ ⎝ ⎜ ⎞ ⎠ ⎟ ( – ) ( ) . – ( )( ) • 2 15 η The second condition, avoidance of subharmonic oscillation, must be met if the operating duty cycle is greater than 50%. The slope compensation circuit within the LT1961 prevents subharmonic oscillation for inductor ripple currents of up to 0.7AP-P, defining the minimum inductor value to be: L V V V V f MIN IN OUT IN OUT = ( – ) 0.7 ( ) These conditions define the absolute minimum inductance. However, it is generally recommended that to prevent excessive output noise, and difficulty in obtaining stability, the ripple current is no more than 40% of the average inductor current. Since inductor ripple is: I V V V V L f P P RIPPLE IN OUT IN OUT − = ( – ) ( )( ) The recommended minimum inductance is: L V V V V I f MIN IN OUT IN OUT OUT = ( ) ( – ) . ( ) ( )( ) 2 0 4 2 The inductor value may need further adjustment for other factors such as output voltage ripple and filtering requirements. Remember also, inductance can drop significantly with DC current and manufacturing tolerance. The inductor must have a rating greater than its peak operating current to prevent saturation resulting in efficiency loss. Peak inductor current is given by: I V I V V V V V L f LPEAK OUT OUT IN IN OUT IN OUT = ( )( ) + − • ( ) η 2 ( )( ) Also, consideration should be given to the DC resistance of the inductor. Inductor resistance contributes directly to the efficiency losses in the overall converter. Suitable inductors are available from Coilcraft, Coiltronics, Dale, Sumida, Toko, Murata, Panasonic and other manufactures. Table 2 PART NUMBER VALUE (uH) ISAT(DC) (Amps) DCR (Ω) HEIGHT (mm) Coiltronics TP1-2R2 2.2 1.3 0.188 1.8 TP2-2R2 2.2 1.5 0.111 2.2 TP3-4R7 4.7 1.5 0.181 2.2 TP4- 100 10 1.5 0.146 3.0 Murata LQH1C1R0M04 1.0 0.51 0.28 1.8 LQH3C1R0M24 1.0 1.0 0.06 2.0 LQH3C2R2M24 2.2 0.79 0.1 2.0 LQH4C1R5M04 1.5 1 0.09 2.6 Sumida CD73- 100 10 1.44 0.080 3.5 CDRH4D18-2R2 2.2 1.32 0.058 1.8 CDRH5D18-6R2 6.2 1.4 0.071 1.8 CDRH5D28-100 10 1.3 0.048 2.8 Coilcraft 1008PS-272M 2.7 1.3 0.14 2.7 LPO1704-222M 2.2 1.6 0.12 1.0 LPO1704-332M 3.3 1.3 0.16 1.0 9 LT1961 1961fa APPLICATIONS INFORMATION W U U U shutdown pin can be used. The threshold voltage of the shutdown pin comparator is 1.35V. A 3μA internal current source defaults the open pin condition to be operating (see Typical Performance Graphs). Current hysteresis is added above the SHDN threshold. This can be used to set voltage hysteresis of the UVLO using the following: R V V A R V V V R A H L H 1 7 2 1 35 1 35 1 3 = − μ = ( − ) + μ . . VH – Turn-on threshold VL – Turn-off threshold Example: switching should not start until the input is above 4.75V and is to stop if the input falls below 3.75V. VH = 4.75V VL = 3.75V R V V A k R V V V k A k 1 4 75 3 75 7 143 2 1 35 4 75 1 35 143 3 50 4 = − μ = = ( − ) + μ = . . . . . . Keep the connections from the resistors to the SHDN pin short and make sure that the interplane or surface capacitance to the switching nodes are minimized. If high resistor values are used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from the switch node. CATCH DIODE The suggested catch diode (D1) is a UPS120 or 1N5818 Schottky. It is rated at 1A average forward current and 20V/30V reverse voltage. Typical forward voltage is 0.5V at 1A. The diode conducts current only during switch off time. Peak reverse voltage is equal to regulator output voltage. Average forward current in normal operation is equal to output current. SHUTDOWN AND UNDERVOLTAGE LOCKOUT Figure 4 shows how to add undervoltage lockout (UVLO) to the LT1961. Typically, UVLO is used in situations where the input supply is current limited, or has a relatively high source resistance. A switching regulator draws constant power from the source, so source current increases as source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where these problems might occur. Figure 4. Undervoltage Lockout 1.35V GND INPUT R1 1961 F04 SHDN VCC IN LT1961 3μA C1 R2 7μA An internal comparator will force the part into shutdown below the minimum VIN of 2.6V. This feature can be used to prevent excessive discharge of battery-operated systems. If an adjustable UVLO threshold is required, the 10 LT1961 1961fa SYNCHRONIZATION The SYNC pin, is used to synchronize the internal oscillator to an external signal. The SYNC input must pass from a logic level low, through the maximum synchronization threshold with a duty cycle between 20% and 80%. The input can be driven directly from a logic level output. The synchronizing range is equal to initial operating frequency up to 2MHz. This means that minimum practical sync frequency is equal to the worst-case high self-oscillating frequency (1.5MHz), not the typical operating frequency of 1.25MHz. Caution should be used when synchronizing above 1.7MHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. Higher inductor values will tend to eliminate this problem. See Frequency Compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation. Application Note 19 has more details on the theory of slope compensation. LAYOUT CONSIDERATIONS As with all high frequency switchers, when considering layout, care must be taken to achieve optimal electrical, thermal and noise performance. For maximum efficiency, switch rise and fall times are typically in the nanosecond range. To prevent noise both radiated and conducted, the APPLICATIONS INFORMATION W U U U high speed switching current path, shown in Figure 5, must be kept as short as possible. This is implemented in the suggested layout of Figure 6. Shortening this path will also reduce the parasitic trace inductance of approximately 25nH/inch. At switch off, this parasitic inductance produces a flyback spike across the LT1961 switch. When operating at higher currents and output voltages, with poor layout, this spike can generate voltages across the LT1961 that may exceed its absolute maximum rating. A ground plane should always be used under the switcher circuitry to prevent interplane coupling and overall noise. The VC and FB components should be kept as far away as possible from the switch node. The LT1961 pinout has been designed to aid in this. The ground for these components should be separated from the switch current path. Failure to do so will result in poor stability or subharmonic like oscillation. Board layout also has a significant effect on thermal resistance. The exposed pad is the copper plate that runs under the LT1961 die. This is the best thermal path for heat out of the package. Soldering the pad onto the board will reduce die temperature and increase the power capability of the LT1961. Provide as much copper area as possible around this pad. Adding multiple solder filled feedthroughs under and around the pad to the ground plane will also help. Similar treatment to the catch diode and inductor terminations will reduce any additional heating effects. 1961 F05 VOUT L1 SW GND LT1961 D1 C1 C3 VIN HIGH FREQUENCY SWITCHING PATH LOAD Figure 5. High Speed Switching Path 11 LT1961 1961fa Figure 6. Typical Application and Suggested Layout (Topside Only Shown) LT1961 VIN OUTPUT 12V 0.5A* INPUT 5V C2 6800pF C4 R3 100pF 6.8k R2 10k 1% R1 90.9k D1 UPS120 C1 10μF CERAMIC C3 2.2μF CERAMIC VSW SHDN FB OPEN OR HIGH = ON SYNC GND VC *MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING. L1 6.8μH VOUT INPUT GND C3 C1 R2 R1 L1 D1 KELVIN SENSE VOUT MINIMIZE LT1961, C1, D1 LOOP KEEP FB AND VC COMPONENTS AWAY FROM HIGH FREQUENCY, HIGH INPUT COMPONENTS PLACE FEEDTHROUGHS AROUND GROUND PIN FOR GOOD THERMAL CONDUCTIVITY LT1961EMS8E GND C4 U1 SOLDER EXPOSED GROUND PAD TO BOARD R3 C2 APPLICATIONS INFORMATION W U U U 12 LT1961 1961fa APPLICATIONS INFORMATION W U U U THERMAL CALCULATIONS Power dissipation in the LT1961 chip comes from four sources: switch DC loss, switch AC loss, drive current, and input quiescent current. The following formulas show how to calculate each of these losses. These formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. DC duty cycle V V V I V I V OUT IN OUT SW OUT OUT IN , ( ) ( )( ) = − = Switch loss: PSW = (DC)(ISW)2(RSW)+ 17n(ISW)(VOUT )(f) VIN loss: P V I DC VIN mA V IN SW = + IN ( )( )( ) ( ) 50 1 RSW = Switch resistance (≈ 0.27Ω hot) Example: VIN = 5V, VOUT = 12V and IOUT = 0.5A Total power dissipation = 0.23 + 0.31 + 0.07 + 0.005 = 0.62W Thermal resistance for LT1961 package is influenced by the presence of internal or backside planes. With a full plane under the package, thermal resistance will be about 50°C/W. To calculate die temperature, use the appropriate thermal resistance number and add in worst-case ambient temperature: TJ = TA + θJA (PTOT) If a true die temperature is required, a measurement of the SYNC to GND pin resistance can be used. The SYNC pin resistance across temperature must first be calibrated, with no device power, in an oven. The same measurement can then be used in operation to indicate the die temperature. FREQUENCY COMPENSATION Loop frequency compensation is performed on the output of the error amplifier (VC pin) with a series RC network. The main pole is formed by the series capacitor and the output impedance (≈500kΩ) of the error amplifier. The pole falls in the range of 2Hz to 20Hz. The series resistor creates a “zero” at 1kHz to 5kHz, which improves loop stability and transient response. A second capacitor, typically one-tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the VC pin. VC pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor, VC pin ripple is: VC Pin Ripple = VRIPPLE = Output ripple (VP–P) gm = Error amplifier transconductance (≈850μmho) RC = Series resistor on VC pin VOUT = DC output voltage 1.2(VRIPPLE)(gm)(RC) (VOUT) To prevent irregular switching, VC pin ripple should be kept below 50mVP–P. Worst-case VC pin ripple occurs at maximum output load current and will also be increased if poor quality (high ESR) output capacitors are used. The addition of a 47pF capacitor on the VC pin reduces switching frequency ripple to only a few millivolts. A low value for RC will also reduce VC pin ripple, but loop phase margin may be inadequate. 13 LT1961 1961fa LT1961 FB VIN VC VIN 5V TO 10V *DALE LPE-4841-100MB GND LT1961 • TA02 S/S VSW P6KE-20A 1N4148 UPS140 UPS140 C1 4.7μF R2 10k 1% R1 115k 1% C2 2.2nF C3 100pF R3 10k –VOUT –15V VOUT 15V C4 47μF C5 47μF ON OFF 2, 3 8, 9 7 T1* 4 10 1 • • • + + + Dual Output Flyback Converter TYPICAL APPLICATIO SU LT1961 VIN GND VIN** 4V TO 9V VC FB LT1961 • TA03 S/S VSW C1 4.7μF 20V C4 2.2nF C5 100pF R1 10k R3 10k 1% R2 31.6k 1% VOUT † 5V C3 47μF 10V ON OFF L1A* 10μH • • L1B* 10μH C2 4.7μF BH ELECTRONICS 511-1012 INPUT VOLTAGE MAY BE GREATER OR LESS THAN OUTPUT VOLTAGE D1 UPS120 VIN 4V 5V 6V 7V 9V IOUT 0.59A 0.65A 0.70A 0.74A 0.80A †MAX IOUT * ** + + 4V-9VIN to 5VOUT SEPIC Converter** 14 LT1961 1961fa LT1961 VIN GND VC FB LT1961 • TA04 S/S VSW L1 4.7μH C1 10μF SINGLE Li-Ion CELL C4 47μF 10V C2 2.2nF C3 100pF R3 10k R2 10k 1% R1 31.6k 1% VOUT 5V D1 UPS120 ON OFF + + + VIN 2.7V 3.3V 3.6V IOUT 0.75A 0.93A 1.0A Single Li-Ion Cell to 5V TYPICAL APPLICATIO SU 15 LT1961 1961fa PACKAGE DESCRIPTIONU Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. MSOP (MS8E) 0307 REV D 0.53 ± 0.152 (.021 ± .006) SEATING PLANE NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.18 (.007) 0.254 (.010) 1.10 (.043) MAX 0.22 – 0.38 (.009 – .015) TYP 0.86 (.034) REF 0.65 (.0256) BSC 0° – 6° TYP DETAIL “A” DETAIL “A” GAUGE PLANE 1 2 3 4 4.90 ± 0.152 (.193 ± .006) 8 8 1 BOTTOM VIEW OF EXPOSED PAD OPTION 7 6 5 3.00 ± 0.102 (.118 ± .004) (NOTE 3) 3.00 ± 0.102 (.118 ± .004) (NOTE 4) 0.52 (.0205) REF 1.83 ± 0.102 (.072 ± .004) 2.06 ± 0.102 (.081 ± .004) 5.23 (.206) MIN 3.20 – 3.45 (.126 – .136) 2.083 ± 0.102 (.082 ± .004) 2.794 ± 0.102 (.110 ± .004) 0.889 ± 0.127 (.035 ± .005) RECOMMENDED SOLDER PAD LAYOUT 0.42 ± 0.038 (.0165 ± .0015) TYP 0.65 (.0256) BSC 0.1016 ± 0.0508 (.004 ± .002) MS8E Package 8-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1662 Rev D) 16 LT1961 1961fa PART NUMBER DESCRIPTION COMMENTS LT1308A 600kHz, 2A, Step-Up Regulator 30V Switch, VIN = 1V to 6V, Low Battery Comparator, S8 Package LT1310 4.5MHz, 1.5A Step-Up with Phase Lock Loop 34V Switch, VIN = 2.75V to 18V, VOUT up to 35V, MS10E Package LT1370 High Efficiency DC/DC Converter 42V Switch, 6A, 500kHz Switch, DD-Pak, TO-220 Package LT1371 High Efficiency DC/DC Converter 35V Switch, 3A, 500kHz Switch, DD-Pak, TO-220 Package LT1372/LT1377 500kHz and 1MHz High Efficiency 1.5A Switching Regulators Boost Topology, VIN(MIN) = 2.7V, S8 Package LT1946/LT1946A 1.2MHz/2.7MHz, 1.5A, Monolithic Step-Up Regulator VIN = 2.6V to 16V, VOUT up to 34V, Integrated SS, MS8 Package LTC3400/ 1.2MHz, 600mA, Synchronous Step-Up VIN = 0.85V to 5V, VOUT to 5.5V, Up to 95% Efficiency, LTC3400B ThinSOT Package LTC3401 Single Cell, High Current (1A), Micropower, Synchronous 3MHz VIN = 0.85V to 5V, VOUT to 5.5V, Up to 97% Efficiency Step-Up DC/DC Converter Synchronizable, Oscillator from 100kHz to 3MHz, MS10 Package LTC3402 Single Cell, High Current (2A), Micropower, Synchronous 3MHz VIN = 0.85V to 5V, VOUT to 5.5V, Up to 95% Efficiency Step-Up DC/DC Converter Synchronizable, Oscillator from 100kHz to 3MHz, MS10 Package LTC3405/ 1.5MHz High Efficiency, IOUT = 300mA, Monolithic Synchronous VIN = 2.5V to 5.5V, VOUT to 0.8V, Up to 95% Efficiency, 100% LTC3405A Step-Down Regulator Duty Cycle, IQ = 20μA, ThinSOT Package ThinSOT is a trademark of Linear Technology Corporation. LT 0707 REV A • PRINTED IN USA © LINEAR TECHNOLOGY CORPORATION 2001 RELATED PARTS Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LASER 190Ω 1% 1N4002 0.1μF (ALL) 10k VIN 10μF VC VIN FB GND 2.2μF VIN 12V TO 25V 150Ω MUR405 L2 10μH LT1961 L1 5 4 1 3 2 11 8 HV DIODES 1800pF 10kV 0.01μF 5kV 1800pF 10kV 47k 5W 2.2μF 0.47μF L1 = Q1, Q2 = 0.47μF = HV DIODES = LASER = COILTRONICS CTX02-11128 ZETEX ZTX849 WIMA 3X 0.15μF TYPE MKP-20 SEMTECH-FM-50 HUGHES 3121H-P 10k LT1961 • TA05 VSW Q1 Q2 + + + COILTRONICS (407) 241-7876 U TYPICAL APPLICATIO High Voltage Laser Power Supply LT3757/LT3757A 1 3757afd n Wide Input Voltage Range: 2.9V to 40V n Positive or Negative Output Voltage Programming with a Single Feedback Pin n Current Mode Control Provides Excellent Transient Response n Programmable Operating Frequency (100kHz to 1MHz) with One External Resistor n Synchronizable to an External Clock n Low Shutdown Current < 1μA n Internal 7.2V Low Dropout Voltage Regulator n Programmable Input Undervoltage Lockout with Hysteresis n Programmable Soft-Start n Small 10-Lead DFN (3mm × 3mm) and Thermally Enhanced 10-Pin MSOP Packages Typical Application Description Boost, Flyback, SEPIC and Inverting Controller The LT®3757/LT3757A are wide input range, current mode, DC/DC controllers which are capable of generating either positive or negative output voltages. They can be configured as either a boost, flyback, SEPIC or inverting converter. The LT3757/LT3757A drive a low side external N-channel power MOSFET from an internal regulated 7.2V supply. The fixed frequency, current-mode architecture results in stable operation over a wide range of supply and output voltages. The operating frequency of LT3757/LT3757A can be set with an external resistor over a 100kHz to 1MHz range, and can be synchronized to an external clock using the SYNC pin. A low minimum operating supply voltage of 2.9V, and a low shutdown quiescent current of less than 1μA, make the LT3757/LT3757A ideally suited for batteryoperated systems. The LT3757/LT3757A feature soft-start and frequency foldback functions to limit inductor current during start-up and output short-circuit. The LT3757A has improved load transient performance compared to the LT3757. High Efficiency Boost Converter Features Applications n Automotive and Industrial Boost, Flyback, SEPIC and Inverting Converters n Telecom Power Supplies n Portable Electronic Equipment Efficiency SENSE LT3757 VIN VIN 8V TO 16V 10μF 25V X5R VOUT 24V 2A 0.01 41.2k 300kHz GATE FBX GND INTVCC SHDN/UVLO SYNC RT SS VC 200k 43.2k 0.1μF 22k 6.8nF 10μH 3757 TA01a 226k 16.2k 4.7μF 10V X5R 10μF 25V X5R 47μF 35V ×2 + OUTPUT CURRENT (A) 0.001 EFFICIENCY (%) 30 50 40 60 70 80 90 100 0.01 0.1 1 3757 TA01b 10 VIN = 8V VIN = 16V L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and No RSENSE and ThinSOT are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. LT3757/LT3757A 2 3757afd Pin Configuration Absolute Maximum Ratings VIN, SHDN/UVLO (Note 6)..........................................40V INTVCC.....................................................VIN + 0.3V, 20V GATE......................................................... INTVCC + 0.3V SYNC...........................................................................8V VC, SS..........................................................................3V RT.............................................................................1.5V SENSE.....................................................................±0.3V FBX.................................................................. –6V to 6V (Note 1) TOP VIEW DD PACKAGE 10-LEAD (3mm × 3mm) PLASTIC DFN 10 9 6 7 8 4 5 3 11 2 1 VIN SHDN/UVLO INTVCC GATE SENSE VC FBX SS RT SYNC TJMAX = 125°C, θJA = 43°C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB 12345 VC FBX SS RT SYNC 10 9876 VIN SHDN/UVLO INTVCC GATE SENSE TOP VIEW MSE PACKAGE 10-LEAD PLASTIC MSOP 11 TJMAX = 150°C, θJA = 40°C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB Order Information LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3757EDD#PBF LT3757EDD#TRPBF LDYW 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LT3757IDD#PBF LT3757IDD#TRPBF LDYW 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LT3757EMSE#PBF LT3757EMSE#TRPBF LTDYX 10-Lead (3mm × 3mm) Plastic MSOP –40°C to 125°C LT3757IMSE#PBF LT3757IMSE#TRPBF LTDYX 10-Lead (3mm × 3mm) Plastic MSOP –40°C to 125°C LT3757HMSE#PBF LT3757HMSE#TRPBF LTDYX 10-Lead (3mm × 3mm) Plastic MSOP –40°C to 150°C LT3757MPMSE#PBF LT3757MPMSE#TRPBF LTDYX 10-Lead (3mm × 3mm) Plastic MSOP –55°C to 150°C LT3757AEDD#PBF LT3757AEDD#TRPBF LGGR 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LT3757AIDD#PBF LT3757AIDD#TRPBF LGGR 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LT3757AEMSE#PBF LT3757AEMSE#TRPBF LTGGM 10-Lead (3mm × 3mm) Plastic MSOP –40°C to 125°C LT3757AIMSE#PBF LT3757AIMSE#TRPBF LTGGM 10-Lead (3mm × 3mm) Plastic MSOP –40°C to 125°C LT3757AHMSE#PBF LT3757AHMSE#TRPBF LTGGM 10-Lead (3mm × 3mm) Plastic MSOP –40°C to 150°C LT3757AMPMSE#PBF LT3757AMPMSE#TRPBF LTGGM 10-Lead (3mm × 3mm) Plastic MSOP –55°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ Operating Temperature Range (Notes 2, 8) LT3757E/LT3757AE............................ –40°C to 125°C LT3757I/LT3757AI.............................. –40°C to 125°C LT3757H/LT3757AH............................ –40°C to 150°C LT3757MP/LT3757AMP...................... –55°C to 150°C Storage Temperature Range DFN..................................................... –65°C to 125°C MSOP................................................. –65°C to 150°C Lead Temperature (Soldering, 10 sec) MSOP................................................................300°C LT3757/LT3757A 3 3757afd E lectrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, SHDN/UVLO = 24V, SENSE = 0V, unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS VIN Operating Range 2.9 40 V VIN Shutdown IQ SHDN/UVLO = 0V SHDN/UVLO = 1.15V 0.1 1 6 μA μA VIN Operating IQ VC = 0.3V, RT = 41.2k 1.6 2.2 mA VIN Operating IQ with Internal LDO Disabled VC = 0.3V, RT = 41.2k, INTVCC = 7.5V 280 400 μA SENSE Current Limit Threshold l 100 110 120 mV SENSE Input Bias Current Current Out of Pin –65 μA Error Amplifier FBX Regulation Voltage (VFBX(REG)) VFBX > 0V (Note 3) VFBX < 0V (Note 3) l l 1.569 –0.816 1.6 –0.80 1.631 –0.784 V V FBX Overvoltage Lockout VFBX > 0V (Note 4) VFBX < 0V (Note 4) 6 7 8 11 10 14 % % FBX Pin Input Current VFBX = 1.6V (Note 3) VFBX = –0.8V (Note 3) –10 70 100 10 nA nA Transconductance gm (ΔIVC/ΔVFBX) (Note 3) 230 μS VC Output Impedance (Note 3) 5 MΩ VFBX Line Regulation [ΔVFBX /(ΔVIN • VFBX(REG))] VFBX > 0V, 2.9V < VIN < 40V (Notes 3, 7) VFBX < 0V, 2.9V < VIN < 40V (Notes 3, 7) 0.002 0.0025 0.056 0.05 %/V %/V VC Current Mode Gain (ΔVVC /ΔVSENSE) 5.5 V/V VC Source Current VFBX = 0V, VC = 1.5V –15 μA VC Sink Current VFBX = 1.7V VFBX = –0.85V 12 11 μA μA Oscillator Switching Frequency RT = 41.2k to GND, VFBX = 1.6V RT = 140k to GND, VFBX = 1.6V RT = 10.5k to GND, VFBX = 1.6V 270 300 100 1000 330 kHz kHz kHz RT Voltage VFBX = 1.6V 1.2 V Minimum Off-Time 220 ns Minimum On-Time 220 ns SYNC Input Low 0.4 V SYNC Input High 1.5 V SS Pull-Up Current SS = 0V, Current Out of Pin –10 μA Low Dropout Regulator INTVCC Regulation Voltage l 7 7.2 7.4 V INTVCC Undervoltage Lockout Threshold Falling INTVCC UVLO Hysteresis 2.6 2.7 0.1 2.8 V V INTVCC Overvoltage Lockout Threshold 16 17.5 V INTVCC Current Limit VIN = 40V VIN = 15V 30 40 95 55 mA mA INTVCC Load Regulation (ΔVINTVCC/ VINTVCC) 0 < IINTVCC < 20mA, VIN = 8V –0.9 –0.5 % INTVCC Line Regulation ΔVINTVCC/(VINTVCC • ΔVIN) 8V < VIN < 40V 0.008 0.03 %/V Dropout Voltage (VIN – VINTVCC) VIN = 6V, IINTVCC = 20mA 400 mV INTVCC Current in Shutdown SHDN/UVLO = 0V, INTVCC = 8V 16 μA LT3757/LT3757A 4 3757afd TEMPERATURE (°C) –75 –50 1580 1585 REGULATED FEEDBACK VOLTAGE (mV) 1590 1605 1600 0 50 75 1595 –25 25 100 125 150 3757 G01 VIN = 40V VIN = 24V VIN = 8V VIN = INTVCC = 2.9V SHDN/UVLO = 1.33V TEMPERATURE (°C) REGULATED FEEDBACK VOLTAGE (mV) –802 –800 –798 –788 –790 –792 –794 –804 –796 3757 G02 –75 –50 –25 0 25 50 75 100 125 150 VIN = 40V VIN = 24V VIN = 8V VIN = INTVCC = 2.9V SHDN/UVLO = 1.33V Typical Performance Characteristics Positive Feedback Voltage vs Temperature, VIN Negative Feedback Voltage vs Temperature, VIN Quiescent Current vs Temperature, VIN TA = 25°C, unless otherwise noted. E lectrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, SHDN/UVLO = 24V, SENSE = 0V, unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS INTVCC Voltage to Bypass Internal LDO 7.5 V Logic Inputs SHDN/UVLO Threshold Voltage Falling VIN = INTVCC = 8V l 1.17 1.22 1.27 V SHDN/UVLO Input Low Voltage I(VIN) Drops Below 1μA 0.4 V SHDN/UVLO Pin Bias Current Low SHDN/UVLO = 1.15V 1.7 2 2.5 μA SHDN/UVLO Pin Bias Current High SHDN/UVLO = 1.30V 10 100 nA Gate Driver tr Gate Driver Output Rise Time CL = 3300pF (Note 5), INTVCC = 7.5V 22 ns tf Gate Driver Output Fall Time CL = 3300pF (Note 5), INTVCC = 7.5V 20 ns Gate VOL 0.05 V Gate VOH INTVCC –0.05 V Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3757E/LT3757AE are guaranteed to meet performance specifications from the 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3757I/LT3757AI are guaranteed over the full –40°C to 125°C operating junction temperature range. The LT3757H/LT3757AH are guaranteed over the full –40°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C. The LT3757MP/LT3757AMP are 100% tested and guaranteed over the full –55°C to 150°C operating junction temperature range. Note 3: The LT3757/LT3757A are tested in a feedback loop which servos VFBX to the reference voltages (1.6V and –0.8V) with the VC pin forced to 1.3V. Note 4: FBX overvoltage lockout is measured at VFBX(OVERVOLTAGE) relative to regulated VFBX(REG). Note 5: Rise and fall times are measured at 10% and 90% levels. Note 6: For VIN below 6V, the SHDN/UVLO pin must not exceed VIN. Note 7: SHDN/UVLO = 1.33V when VIN = 2.9V. Note 8: The LT3757/LT3757A include overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating junction temperature when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. –75 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1.4 QUIESCENT CURRENT (mA) 1.6 1.8 1.5 1.7 3757 G03 VIN = 40V VIN = 24V VIN = INTVCC = 2.9V LT3757/LT3757A 5 3757afd Typical Performance Characteristics Switching Frequency vs Temperature SENSE Current Limit Threshold vs Temperature SENSE Current Limit Threshold vs Duty Cycle SHDN/UVLO Threshold vs Temperature SHDN/UVLO Current vs Voltage SHDN/UVLO Hysteresis Current vs Temperature Dynamic Quiescent Current vs Switching Frequency RT vs Switching Frequency Normalized Switching Frequency vs FBX TA = 25°C, unless otherwise noted. FBX VOLTAGE (V) –0.8 0 NORMALIZED FREQUENCY (%) 20 40 60 80 120 –0.4 0 0.4 0.8 3757 G06 1.2 1.6 100 –75 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 100 SENSE THRESHOLD (mV) 105 110 115 120 3757 G08 DUTY CYCLE (%) 0 95 SENSE THRESHOLD (mV) 105 20 40 60 80 115 100 110 100 3757 G09 SHDN/UVLO VOLTAGE (V) 0 0 SHDN/UVLO CURRENT (μA) 20 10 20 30 40 10 30 40 3757 G11 –75 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1.6 ISHDN/UVLO (μA) 1.8 2.0 2.2 2.4 3757 G12 SWITCHING FREQUENCY (KHz) 0 0 IQ(mA) 15 20 35 300 500 600 700 10 5 25 30 100 200 400 800 900 1000 3757 G04 CL = 3300pF SWITCHING FREQUENCY (KHz) 0 10 RT (k ) 100 1000 100 200 300 400 500 600 700 800 900 1000 3757 G05 –75 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 270 SWITCHING FREQUENCY (kHz) 280 290 300 310 330 3757 G07 320 RT = 41.2K –75 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1.18 SHDN/UVLO VOLTAGE (V) 1.22 1.24 1.26 1.28 1.20 3757 G10 SHDN/UVLO FALLING SHDN/UVLO RISING LT3757/LT3757A 6 3757afd Typical Performance Characteristics INTVCC Line Regulation INTVCC Dropout Voltage vs Current, Temperature Gate Drive Rise and Fall Time vs INTVCC Typical Start-Up Waveforms INTVCC vs Temperature INTVCC Minimum Output Current vs VIN INTVCC Load Regulation TA = 25°C, unless otherwise noted. Gate Drive Rise and Fall Time vs CL FBX Frequency Foldback Waveforms During Overcurrent –75 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 7.0 INTVCC (V) 7.1 7.2 7.3 7.4 3757 G13 VIN (V) 0 INTVCC VOLTAGE (V) 35 7.25 7.20 5 10 15 20 25 30 40 7.15 7.10 7.30 3757 G16 CL (nF) 0 TIME (ns) 60 70 80 50 40 5 10 15 20 25 30 10 0 30 90 20 3757 G18 RISE TIME INTVCC = 7.2V FALL TIME INTVCC (V) 3 TIME (ns) 20 25 15 10 6 9 12 15 5 0 30 3757 G19 CL = 3300pF RISE TIME FALL TIME 2ms/DIV VOUT 5V/DIV IL1A + IL1B 5A/DIV 3757 G20 VIN = 12V PAGE 31 CIRCUIT 50μs/DIV PAGE 31 CIRCUIT VOUT 10V/DIV VSW 20V/DIV IL1A + IL1B 5A/DIV 3757 G21 VIN = 12V INTVCC LOAD (mA) 0 6.8 7 7.1 7.2 7.3 20 40 50 60 6.9 10 30 70 3757 G15 INTVCC VOLTAGE (V) VIN = 8V INTVCC LOAD (mA) 0 DROPOUT VOLTAGE (mV) 500 600 300 400 200 5 10 15 20 100 0 700 3757 G17 150°C 125°C 25°C 0°C –55°C 75°C VIN = 6V VIN (V) 0 INTVCC CURRENT (mA) 50 60 70 40 3757 G14 40 30 0 10 5 10 15 20 25 30 35 20 90 80 TJ = 150°C INTVCC = 6V INTVCC = 4.5V LT3757/LT3757A 7 3757afd Pin Functions VC (Pin 1): Error Amplifier Compensation Pin. Used to stabilize the voltage loop with an external RC network. FBX (Pin 2): Positive and Negative Feedback Pin. Receives the feedback voltage from the external resistor divider across the output. Also modulates the frequency during start-up and fault conditions when FBX is close to GND. SS (Pin 3): Soft-Start Pin. This pin modulates compensation pin voltage (VC) clamp. The soft-start interval is set with an external capacitor. The pin has a 10μA (typical) pull-up current source to an internal 2.5V rail. The soft-start pin is reset to GND by an undervoltage condition at SHDN/ UVLO, an INTVCC undervoltage or overvoltage condition or an internal thermal lockout. RT (Pin 4): Switching Frequency Adjustment Pin. Set the frequency using a resistor to GND. Do not leave this pin open. SYNC (Pin 5): Frequency Synchronization Pin. Used to synchronize the switching frequency to an outside clock. If this feature is used, an RT resistor should be chosen to program a switching frequency 20% slower than the SYNC pulse frequency. Tie the SYNC pin to GND if this feature is not used. SYNC is ignored when FBX is close to GND. SENSE (Pin 6): The Current Sense Input for the Control Loop. Kelvin connect this pin to the positive terminal of the switch current sense resistor in the source of the N-channel MOSFET. The negative terminal of the current sense resistor should be connected to GND plane close to the IC. GATE (Pin 7): N-Channel MOSFET Gate Driver Output. Switches between INTVCC and GND. Driven to GND when IC is shut down, during thermal lockout or when INTVCC is above or below the OV or UV thresholds, respectively. INTVCC (Pin 8): Regulated Supply for Internal Loads and Gate Driver. Supplied from VIN and regulated to 7.2V (typical). INTVCC must be bypassed with a minimum of 4.7μF capacitor placed close to pin. INTVCC can be connected directly to VIN, if VIN is less than 17.5V. INTVCC can also be connected to a power supply whose voltage is higher than 7.5V, and lower than VIN, provided that supply does not exceed 17.5V. SHDN/UVLO (Pin 9): Shutdown and Undervoltage Detect Pin. An accurate 1.22V (nominal) falling threshold with externally programmable hysteresis detects when power is okay to enable switching. Rising hysteresis is generated by the external resistor divider and an accurate internal 2μA pull-down current. An undervoltage condition resets sort-start. Tie to 0.4V, or less, to disable the device and reduce VIN quiescent current below 1μA. VIN (Pin 10): Input Supply Pin. Must be locally bypassed with a 0.22μF, or larger, capacitor placed close to the pin. Exposed Pad (Pin 11): Ground. This pin also serves as the negative terminal of the current sense resistor. The Exposed Pad must be soldered directly to the local ground plane. LT3757/LT3757A 8 3757afd Block Diagram Figure 1. LT3757 Block Diagram Working as a SEPIC Converter L1 R1 R4 R3 M1 L2 R2 FBX 1.22V 2.5V CDC D1 CIN VOUT COUT2 COUT1 CVCC INTVCC VIN RSENSE VISENSE • + + VIN IS1 2μA 10 8 7 1 9 SHDN/UVLO INTERNAL REGULATOR AND UVLO TSD 165°C A10 Q3 VC VC 17.5V 2.7V UP 2.6V DOWN A8 UVLO IS2 10μA IS3 CC1 CC2 RC DRIVER SLOPE SENSE GND GATE 108mV SR1 + – + – CURRENT LIMIT RAMP GENERATOR 7.2V LDO • + – + – R O S 2.5V G1 RT RT SS CSS SYNC 1.25V 1.25V FBX FBX 1.6V –0.8V + – + – + – 2 3 5 4 + – + – 6 11 RAMP PWM COMPARATOR FREQUENCY FOLDBACK 100kHz-1MHz OSCILLATOR FREQ FOLDBACK FREQ PROG 3757 F01 – ++ Q1 A1 A2 1.72V –0.88V + – + – A11 A12 A3 A4 A5 A6 G5 G2 G6 A7 A9 Q2 D2 R5 8k D3 G4 G3 LT3757/LT3757A 9 3757afd Applications Information Main Control Loop The LT3757 uses a fixed frequency, current mode control scheme to provide excellent line and load regulation. Operation can be best understood by referring to the Block Diagram in Figure 1. The start of each oscillator cycle sets the SR latch (SR1) and turns on the external power MOSFET switch M1 through driver G2. The switch current flows through the external current sensing resistor RSENSE and generates a voltage proportional to the switch current. This current sense voltage VISENSE (amplified by A5) is added to a stabilizing slope compensation ramp and the resulting sum (SLOPE) is fed into the positive terminal of the PWM comparator A7. When SLOPE exceeds the level at the negative input of A7 (VC pin), SR1 is reset, turning off the power switch. The level at the negative input of A7 is set by the error amplifier A1 (or A2) and is an amplified version of the difference between the feedback voltage (FBX pin) and the reference voltage (1.6V or –0.8V, depending on the configuration). In this manner, the error amplifier sets the correct peak switch current level to keep the output in regulation. The LT3757 has a switch current limit function. The current sense voltage is input to the current limit comparator A6. If the SENSE pin voltage is higher than the sense current limit threshold VSENSE(MAX) (110mV, typical), A6 will reset SR1 and turn off M1 immediately. The LT3757 is capable of generating either positive or negative output voltage with a single FBX pin. It can be configured as a boost, flyback or SEPIC converter to generate positive output voltage, or as an inverting converter to generate negative output voltage. When configured as a SEPIC converter, as shown in Figure 1, the FBX pin is pulled up to the internal bias voltage of 1.6V by a voltage divider (R1 and R2) connected from VOUT to GND. Comparator A2 becomes inactive and comparator A1 performs the inverting amplification from FBX to VC. When the LT3757 is in an inverting configuration, the FBX pin is pulled down to –0.8V by a voltage divider connected from VOUT to GND. Comparator A1 becomes inactive and comparator A2 performs the noninverting amplification from FBX to VC. The LT3757 has overvoltage protection functions to protect the converter from excessive output voltage overshoot during start-up or recovery from a short-circuit condition. An overvoltage comparator A11 (with 20mV hysteresis) senses when the FBX pin voltage exceeds the positive regulated voltage (1.6V) by 8% and provides a reset pulse. Similarly, an overvoltage comparator A12 (with 10mV hysteresis) senses when the FBX pin voltage exceeds the negative regulated voltage (–0.8V) by 11% and provides a reset pulse. Both reset pulses are sent to the main RS latch (SR1) through G6 and G5. The power MOSFET switch M1 is actively held off for the duration of an output overvoltage condition. Programming Turn-On and Turn-Off Thresholds with the SHDN/UVLO Pin The SHDN/UVLO pin controls whether the LT3757 is enabled or is in shutdown state. A micropower 1.22V reference, a comparator A10 and a controllable current source IS1 allow the user to accurately program the supply voltage at which the IC turns on and off. The falling value can be accurately set by the resistor dividers R3 and R4. When SHDN/UVLO is above 0.7V, and below the 1.22V threshold, the small pull-down current source IS1 (typical 2μA) is active. The purpose of this current is to allow the user to program the rising hysteresis. The Block Diagram of the comparator and the external resistors is shown in Figure 1. The typical falling threshold voltage and rising threshold voltage can be calculated by the following equations: VVIN,FALLING = 1.22 • (R3 +R4) R4 VVIN,RISING = 2μA •R3+ VIN,FALLING For applications where the SHDN/UVLO pin is only used as a logic input, the SHDN/UVLO pin can be connected directly to the input voltage VIN for always-on operation. LT3757/LT3757A 10 3757afd Applications Information INTVCC Regulator Bypassing and Operation An internal, low dropout (LDO) voltage regulator produces the 7.2V INTVCC supply which powers the gate driver, as shown in Figure 1. If a low input voltage operation is expected (e.g., supplying power from a lithium-ion battery or a 3.3V logic supply), low threshold MOSFETs should be used. The LT3757 contains an undervoltage lockout comparator A8 and an overvoltage lockout comparator A9 for the INTVCC supply. The INTVCC undervoltage (UV) threshold is 2.7V (typical), with 100mV hysteresis, to ensure that the MOSFETs have sufficient gate drive voltage before turning on. The logic circuitry within the LT3757 is also powered from the internal INTVCC supply. The INTVCC overvoltage (OV) threshold is set to be 17.5V (typical) to protect the gate of the power MOSFET. When INTVCC is below the UV threshold, or above the OV threshold, the GATE pin will be forced to GND and the soft-start operation will be triggered. The INTVCC regulator must be bypassed to ground immediately adjacent to the IC pins with a minimum of 4.7μF ceramic capacitor. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate driver. In an actual application, most of the IC supply current is used to drive the gate capacitance of the power MOSFET. The on-chip power dissipation can be a significant concern when a large power MOSFET is being driven at a high frequency and the VIN voltage is high. It is important to limit the power dissipation through selection of MOSFET and/ or operating frequency so the LT3757 does not exceed its maximum junction temperature rating. The junction temperature TJ can be estimated using the following equations: TJ = TA + PIC • θJA TA = ambient temperature θJA = junction-to-ambient thermal resistance PIC = IC power consumption = VIN • (IQ + IDRIVE) IQ = VIN operation IQ = 1.6mA IDRIVE = average gate drive current = f • QG f = switching frequency QG = power MOSFET total gate charge The LT3757 uses packages with an Exposed Pad for enhanced thermal conduction. With proper soldering to the Exposed Pad on the underside of the package and a full copper plane underneath the device, thermal resistance (θJA) will be about 43°C/W for the DD package and 40°C/W for the MSE package. For an ambient board temperature of TA = 70°C and maximum junction temperature of 125°C, the maximum IDRIVE (IDRIVE(MAX)) of the DD package can be calculated as: IDRIVE(MAX) = (TJ − TA) (θJA • VIN) −IQ = 1.28W VIN − 1.6mA The LT3757 has an internal INTVCC IDRIVE current limit function to protect the IC from excessive on-chip power dissipation. The IDRIVE current limit decreases as the VIN increases (see the INTVCC Minimum Output Current vs VIN graph in the Typical Performance Characteristics section). If IDRIVE reaches the current limit, INTVCC voltage will fall and may trigger the soft-start. Based on the preceding equation and the INTVCC Minimum Output Current vs VIN graph, the user can calculate the maximum MOSFET gate charge the LT3757 can drive at a given VIN and switch frequency. A plot of the maximum QG vs VIN at different frequencies to guarantee a minimum 4.5V INTVCC is shown in Figure 2. As illustrated in Figure 2, a trade-off between the operating frequency and the size of the power MOSFET may be needed in order to maintain a reliable IC junction temperature. Figure 2. Recommended Maximum QG vs VIN at Different Frequencies to Ensure INTVCC Higher Than 4.5V VIN (V) 0 QG (nC) 200 250 150 100 5 10 15 20 25 30 35 40 50 0 300 3757 F02 300kHz 1MHz LT3757/LT3757A 11 3757afd Applications Information Prior to lowering the operating frequency, however, be sure to check with power MOSFET manufacturers for their most recent low QG, low RDS(ON) devices. Power MOSFET manufacturing technologies are continually improving, with newer and better performance devices being introduced almost yearly. An effective approach to reduce the power consumption of the internal LDO for gate drive is to tie the INTVCC pin to an external voltage source high enough to turn off the internal LDO regulator. If the input voltage VIN does not exceed the absolute maximum rating of both the power MOSFET gate-source voltage (VGS) and the INTVCC overvoltage lockout threshold voltage (17.5V), the INTVCC pin can be shorted directly to the VIN pin. In this condition, the internal LDO will be turned off and the gate driver will be powered directly from the input voltage, VIN. With the INTVCC pin shorted to VIN, however, a small current (around 16μA) will load the INTVCC in shutdown mode. For applications that require the lowest shutdown mode input supply current, do not connect the INTVCC pin to VIN. In SEPIC or flyback applications, the INTVCC pin can be connected to the output voltage VOUT through a blocking diode, as shown in Figure 3, if VOUT meets the following conditions: 1. VOUT < VIN (pin voltage) 2. VOUT < 17.5V 3. VOUT < maximum VGS rating of power MOSFET A resistor RVCC can be connected, as shown in Figure 3, to limit the inrush current from VOUT. Regardless of whether or not the INTVCC pin is connected to an external voltage source, it is always necessary to have the driver circuitry bypassed with a 4.7μF low ESR ceramic capacitor to ground immediately adjacent to the INTVCC and GND pins. Figure 3. Connecting INTVCC to VOUT CVCC 4.7μF VOUT 3757 F03 INTVCC GND LT3757 RVCC DVCC Operating Frequency and Synchronization The choice of operating frequency may be determined by on-chip power dissipation, otherwise it is a trade-off between efficiency and component size. Low frequency operation improves efficiency by reducing gate drive current and MOSFET and diode switching losses. However, lower frequency operation requires a physically larger inductor. Switching frequency also has implications for loop compensation. The LT3757 uses a constant-frequency architecture that can be programmed over a 100kHz to 1000kHz range with a single external resistor from the RT pin to ground, as shown in Figure 1. The RT pin must have an external resistor to GND for proper operation of the LT3757. A table for selecting the value of RT for a given operating frequency is shown in Table 1. Table 1. Timing Resistor (RT) Value OSCILLATOR FREQUENCY (kHz) RT (kΩ) 100 140 200 63.4 300 41.2 400 30.9 500 24.3 600 19.6 700 16.5 800 14 900 12.1 1000 10.5 The operating frequency of the LT3757 can be synchronized to an external clock source. By providing a digital clock signal into the SYNC pin, the LT3757 will operate at the SYNC clock frequency. If this feature is used, an RT resistor should be chosen to program a switching frequency 20% slower than SYNC pulse frequency. The SYNC pulse should have a minimum pulse width of 200ns. Tie the SYNC pin to GND if this feature is not used. LT3757/LT3757A 12 3757afd Applications Information Duty Cycle Consideration Switching duty cycle is a key variable defining converter operation. As such, its limits must be considered. Minimum on-time is the smallest time duration that the LT3757 is capable of turning on the power MOSFET. This time is generally about 220ns (typical) (see Minimum On-Time in the Electrical Characteristics table). In each switching cycle, the LT3757 keeps the power switch off for at least 220ns (typical) (see Minimum Off-Time in the Electrical Characteristics table). The minimum on-time and minimum off-time and the switching frequency define the minimum and maximum switching duty cycles a converter is able to generate: Minimum duty cycle = minimum on-time • frequency Maximum duty cycle = 1 – (minimum off-time • frequency) Programming the Output Voltage The output voltage (VOUT) is set by a resistor divider, as shown in Figure 1. The positive and negative VOUT are set by the following equations: VOUT,POSITIVE = 1.6V • 1+ R2 R1 VOUT,NEGATIVE = –0.8V • 1+ R2 R1 The resistors R1 and R2 are typically chosen so that the error caused by the current flowing into the FBX pin during normal operation is less than 1% (this translates to a maximum value of R1 at about 158k). In the applications where VOUT is pulled up by an external positive power supply, the FBX pin is also pulled up through the R2 and R1 network. Make sure the FBX does not exceed its absolute maximum rating (6V). The R5, D2, and D3 in Figure 1 provide a resistive clamp in the positive direction. To ensure FBX is lower than 6V, choose sufficiently large R1 and R2 to meet the following condition: 6V • 1+ R2 R1 + 3.5V • R2 8kΩ > VOUT(MAX) where VOUT(MAX) is the maximum VOUT that is pulled up by an external power supply. Soft-Start The LT3757 contains several features to limit peak switch currents and output voltage (VOUT) overshoot during start-up or recovery from a fault condition. The primary purpose of these features is to prevent damage to external components or the load. High peak switch currents during start-up may occur in switching regulators. Since VOUT is far from its final value, the feedback loop is saturated and the regulator tries to charge the output capacitor as quickly as possible, resulting in large peak currents. A large surge current may cause inductor saturation or power switch failure. The LT3757 addresses this mechanism with the SS pin. As shown in Figure 1, the SS pin reduces the power MOSFET current by pulling down the VC pin through Q2. In this way the SS allows the output capacitor to charge gradually toward its final value while limiting the start-up peak currents. The typical start-up waveforms are shown in the Typical Performance Characteristics section. The inductor current IL slewing rate is limited by the soft-start function. Besides start-up, soft-start can also be triggered by the following faults: 1. INTVCC > 17.5V 2. INTVCC < 2.6V 3. Thermal lockout Any of these three faults will cause the LT3757 to stop switching immediately. The SS pin will be discharged by Q3. When all faults are cleared and the SS pin has been discharged below 0.2V, a 10μA current source IS2 starts charging the SS pin, initiating a soft-start operation. The soft-start interval is set by the soft-start capacitor selection according to the equation: TSS =CSS • 1.25V 10μA LT3757/LT3757A 13 3757afd Applications Information FBX Frequency Foldback When VOUT is very low during start-up or a short-circuit fault on the output, the switching regulator must operate at low duty cycles to maintain the power switch current within the current limit range, since the inductor current decay rate is very low during switch off time. The minimum on-time limitation may prevent the switcher from attaining a sufficiently low duty cycle at the programmed switching frequency. So, the switch current will keep increasing through each switch cycle, exceeding the programmed current limit. To prevent the switch peak currents from exceeding the programmed value, the LT3757 contains a frequency foldback function to reduce the switching frequency when the FBX voltage is low (see the Normalized Switching Frequency vs FBX graph in the Typical Performance Characteristics section). The typical frequency foldback waveforms are shown in the Typical Performance Characteristics section. The frequency foldback function prevents IL from exceeding the programmed limits because of the minimum on-time. During frequency foldback, external clock synchronization is disabled to prevent interference with frequency reducing operation. Thermal Lockout If LT3757 die temperature reaches 165°C (typical), the part will go into thermal lockout. The power switch will be turned off. A soft-start operation will be triggered. The part will be enabled again when the die temperature has dropped by 5°C (nominal). Loop Compensation Loop compensation determines the stability and transient performance. The LT3757/LT3757A use current mode control to regulate the output which simplifies loop compensation. The LT3757A improves the no-load to heavy load transient response, when compared to the LT3757. New internal circuits ensure that the transient from not switching to switching at high current can be made in a few cycles. The optimum values depend on the converter topology, the component values and the operating conditions (including the input voltage, load current, etc.). To compensate the feedback loop of the LT3757/LT3757A, a series resistorcapacitor network is usually connected from the VC pin to GND. Figure 1 shows the typical VC compensation network. For most applications, the capacitor should be in the range of 470pF to 22nF, and the resistor should be in the range of 5k to 50k. A small capacitor is often connected in parallel with the RC compensation network to attenuate the VC voltage ripple induced from the output voltage ripple through the internal error amplifier. The parallel capacitor usually ranges in value from 10pF to 100pF. A practical approach to design the compensation network is to start with one of the circuits in this data sheet that is similar to your application, and tune the compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. SENSE Pin Programming For control and protection, the LT3757 measures the power MOSFET current by using a sense resistor (RSENSE) between GND and the MOSFET source. Figure 4 shows a typical waveform of the sense voltage (VSENSE) across the sense resistor. It is important to use Kelvin traces between the SENSE pin and RSENSE, and to place the IC GND as close as possible to the GND terminal of the RSENSE for proper operation. Figure 4. The Sense Voltage During a Switching Cycle 3757 F04 VSENSE(PEAK) ΔVSENSE = χ • VSENSE(MAX) VSENSE DT t S VSENSE(MAX) TS LT3757/LT3757A 14 3757afd Applications Information Due to the current limit function of the SENSE pin, RSENSE should be selected to guarantee that the peak current sense voltage VSENSE(PEAK) during steady state normal operation is lower than the SENSE current limit threshold (see the Electrical Characteristics table). Given a 20% margin, VSENSE(PEAK) is set to be 80mV. Then, the maximum switch ripple current percentage can be calculated using the following equation: c = ΔVSENSE 80mV − 0.5 • ΔVSENSE c is used in subsequent design examples to calculate inductor value. ΔVSENSE is the ripple voltage across RSENSE. The LT3757 switching controller incorporates 100ns timing interval to blank the ringing on the current sense signal immediately after M1 is turned on. This ringing is caused by the parasitic inductance and capacitance of the PCB trace, the sense resistor, the diode, and the MOSFET. The 100ns timing interval is adequate for most of the LT3757 applications. In the applications that have very large and long ringing on the current sense signal, a small RC filter can be added to filter out the excess ringing. Figure 5 shows the RC filter on SENSE pin. It is usually sufficient to choose 22Ω for RFLT and 2.2nF to 10nF for CFLT. Keep RFLT’s resistance low. Remember that there is 65μA (typical) flowing out of the SENSE pin. Adding RFLT will affect the SENSE current limit threshold: VSENSE_ILIM = 108mV – 65μA • RFLT Application Circuits The LT3757 can be configured as different topologies. The first topology to be analyzed will be the boost converter, followed by the flyback, SEPIC and inverting converters. Boost Converter: Switch Duty Cycle and Frequency The LT3757 can be configured as a boost converter for the applications where the converter output voltage is higher than the input voltage. Remember that boost converters are not short-circuit protected. Under a shorted output condition, the inductor current is limited only by the input supply capability. For applications requiring a step-up converter that is short-circuit protected, please refer to the Applications Information section covering SEPIC converters. The conversion ratio as a function of duty cycle is VOUT VIN = 1 1−D in continuous conduction mode (CCM). For a boost converter operating in CCM, the duty cycle of the main switch can be calculated based on the output voltage (VOUT) and the input voltage (VIN). The maximum duty cycle (DMAX) occurs when the converter has the minimum input voltage: DMAX = VOUT − VIN(MIN) VOUT Discontinuous conduction mode (DCM) provides higher conversion ratios at a given frequency at the cost of reduced efficiencies and higher switching currents. Figure 5. The RC Filter on SENSE Pin CFLT 3757 F05 LT3757 RFLT RSENSE M1 SENSE GATE GND LT3757/LT3757A 15 3757afd Applications Information Boost Converter: Inductor and Sense Resistor Selection For the boost topology, the maximum average inductor current is: IL(MAX) =IO(MAX) • 1 1−DMAX Then, the ripple current can be calculated by: ΔIL = c •IL(MAX) = c •IO(MAX) • 1 1−DMAX The constant c in the preceding equation represents the percentage peak-to-peak ripple current in the inductor, relative to IL(MAX). The inductor ripple current has a direct effect on the choice of the inductor value. Choosing smaller values of ΔIL requires large inductances and reduces the current loop gain (the converter will approach voltage mode). Accepting larger values of ΔIL provides fast transient response and allows the use of low inductances, but results in higher input current ripple and greater core losses. It is recommended that c fall within the range of 0.2 to 0.6. Given an operating input voltage range, and having chosen the operating frequency and ripple current in the inductor, the inductor value of the boost converter can be determined using the following equation: L = VIN(MIN) ΔIL • f •DMAX The peak and RMS inductor current are: IL(PEAK) =IL(MAX) • 1+ c 2 IL(RMS) =IL(MAX) • 1+ c2 12 Based on these equations, the user should choose the inductors having sufficient saturation and RMS current ratings. Set the sense voltage at IL(PEAK) to be the minimum of the SENSE current limit threshold with a 20% margin. The sense resistor value can then be calculated to be: RSENSE = 80mV IL(PEAK) Boost Converter: Power MOSFET Selection Important parameters for the power MOSFET include the drain-source voltage rating (VDS), the threshold voltage (VGS(TH)), the on-resistance (RDS(ON)), the gate to source and gate to drain charges (QGS and QGD), the maximum drain current (ID(MAX)) and the MOSFET’s thermal resistances (RθJC and RθJA). The power MOSFET will see full output voltage, plus a diode forward voltage, and any additional ringing across its drain-to-source during its off-time. It is recommended to choose a MOSFET whose BVDSS is higher than VOUT by a safety margin (a 10V safety margin is usually sufficient). The power dissipated by the MOSFET in a boost converter is: PFET = I2 L(MAX) • RDS(ON) • DMAX + 2 • V2 OUT • IL(MAX) • CRSS • f /1A The first term in the preceding equation represents the conduction losses in the device, and the second term, the switching loss. CRSS is the reverse transfer capacitance, which is usually specified in the MOSFET characteristics. For maximum efficiency, RDS(ON) and CRSS should be minimized. From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following equation: TJ = TA + PFET • θJA = TA + PFET • (θJC + θCA) TJ must not exceed the MOSFET maximum junction temperature rating. It is recommended to measure the MOSFET temperature in steady state to ensure that absolute maximum ratings are not exceeded. LT3757/LT3757A 16 3757afd Applications Information Figure 6. The Output Ripple Waveform of a Boost Converter VOUT (AC) tON ΔVESR RINGING DUE TO TOTAL INDUCTANCE (BOARD + CAP) ΔVCOUT 3757 F05 tOFF Boost Converter: Output Diode Selection To maximize efficiency, a fast switching diode with low forward drop and low reverse leakage is desirable. The peak reverse voltage that the diode must withstand is equal to the regulator output voltage plus any additional ringing across its anode-to-cathode during the on-time. The average forward current in normal operation is equal to the output current, and the peak current is equal to: ID(PEAK) =IL(PEAK) = 1+ c 2 •IL(MAX) It is recommended that the peak repetitive reverse voltage rating VRRM is higher than VOUT by a safety margin (a 10V safety margin is usually sufficient). The power dissipated by the diode is: PD = IO(MAX) • VD and the diode junction temperature is: TJ = TA + PD • RθJA The RθJA to be used in this equation normally includes the RθJC for the device plus the thermal resistance from the board to the ambient temperature in the enclosure. TJ must not exceed the diode maximum junction temperature rating. Boost Converter: Output Capacitor Selection Contributions of ESR (equivalent series resistance), ESL (equivalent series inductance) and the bulk capacitance must be considered when choosing the correct output capacitors for a given output ripple voltage. The effect of The choice of component(s) begins with the maximum acceptable ripple voltage (expressed as a percentage of the output voltage), and how this ripple should be divided between the ESR step ΔVESR and the charging/discharging ΔVCOUT. For the purpose of simplicity, we will choose 2% for the maximum output ripple, to be divided equally between ΔVESR and ΔVCOUT. This percentage ripple will change, depending on the requirements of the application, and the following equations can easily be modified. For a 1% contribution to the total ripple voltage, the ESR of the output capacitor can be determined using the following equation: ESRCOUT ≤ 0.01• VOUT ID(PEAK) these three parameters (ESR, ESL and bulk C) on the output voltage ripple waveform for a typical boost converter is illustrated in Figure 6. LT3757/LT3757A 17 3757afd Applications Information For the bulk C component, which also contributes 1% to the total ripple: COUT ≥ IO(MAX) 0.01• VOUT • f The output capacitor in a boost regulator experiences high RMS ripple currents, as shown in Figure 6. The RMS ripple current rating of the output capacitor can be determined using the following equation: IRMS(COUT) ≥IO(MAX) • DMAX 1−DMAX Multiple capacitors are often paralleled to meet ESR requirements. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the required RMS current rating. Additional ceramic capacitors in parallel are commonly used to reduce the effect of parasitic inductance in the output capacitor, which reduces high frequency switching noise on the converter output. Boost Converter: Input Capacitor Selection The input capacitor of a boost converter is less critical than the output capacitor, due to the fact that the inductor is in series with the input, and the input current waveform is continuous. The input voltage source impedance determines the size of the input capacitor, which is typically in the range of 10μF to 100μF. A low ESR capacitor is recommended, although it is not as critical as for the output capacitor. The RMS input capacitor ripple current for a boost converter is: IRMS(CIN) = 0.3 • ΔIL Flyback Converter Applications The LT3757 can be configured as a flyback converter for the applications where the converters have multiple outputs, high output voltages or isolated outputs. Figure 7 shows a simplified flyback converter. The flyback converter has a very low parts count for multiple outputs, and with prudent selection of turns ratio, can have high output/input voltage conversion ratios with a desirable duty cycle. However, it has low efficiency due to the high peak currents, high peak voltages and consequent power loss. The flyback converter is commonly used for an output power of less than 50W. The flyback converter can be designed to operate either in continuous or discontinuous mode. Compared to continuous mode, discontinuous mode has the advantage of smaller transformer inductances and easy loop compensation, and the disadvantage of higher peak-to-average current and lower efficiency. In the high output voltage applications, the flyback converters can be designed to operate in discontinuous mode to avoid using large transformers. Figure 7. A Simplified Flyback Converter RSENSE NP:NS VIN CIN CSN VSN LP D SUGGESTED RCD SNUBBER ID ISW VDS 3757 F06 GATE GND LT3757 SENSE LS M + – + – RSN DSN – + + COUT + LT3757/LT3757A 18 3757afd Applications Information Flyback Converter: Switch Duty Cycle and Turns Ratio The flyback converter conversion ratio in the continuous mode operation is: VOUT VIN = NS NP • D 1−D where NS/NP is the second to primary turns ratio. Figure 8 shows the waveforms of the flyback converter in discontinuous mode operation. During each switching period TS, three subintervals occur: DTS, D2TS, D3TS. During DTS, M is on, and D is reverse-biased. During D2TS, M is off, and LS is conducting current. Both LP and LS currents are zero during D3TS. The flyback converter conversion ratio in the discontinuous mode operation is: VOUT VIN = NS NP • D D2 According to the preceding equations, the user has relative freedom in selecting the switch duty cycle or turns ratio to suit a given application. The selections of the duty cycle and the turns ratio are somewhat iterative processes, due to the number of variables involved. The user can choose either a duty cycle or a turns ratio as the start point. The following trade-offs should be considered when selecting the switch duty cycle or turns ratio, to optimize the converter performance. A higher duty cycle affects the flyback converter in the following aspects: • Lower MOSFET RMS current ISW(RMS), but higher MOSFET VDS peak voltage • Lower diode peak reverse voltage, but higher diode RMS current ID(RMS) • Higher transformer turns ratio (NP/NS) The choice, D D+D2 = 1 3 (for discontinuous mode operation with a given D3) gives the power MOSFET the lowest power stress (the product of RMS current and peak voltage). However, in the high output voltage applications, a higher duty cycle may be adopted to limit the large peak reverse voltage of the diode. The choice, D D+D2 = 2 3 (for discontinuous mode operation with a given D3) gives the diode the lowest power stress (the product of RMS current and peak voltage). An extreme high or low duty cycle results in high power stress on the MOSFET or diode, and reduces efficiency. It is recommended to choose a duty cycle, D, between 20% and 80%. Figure 8. Waveforms of the Flyback Converter in Discontinuous Mode Operation 3757 F07 ISW VDS ID DTS D2TS D3TS t ISW(MAX) ID(MAX) TS LT3757/LT3757A 19 3757afd Applications Information Flyback Converter: Transformer Design for Discontinuous Mode Operation The transformer design for discontinuous mode of operation is chosen as presented here. According to Figure 8, the minimum D3 (D3MIN) occurs when the converter has the minimum VIN and the maximum output power (POUT). Choose D3MIN to be equal to or higher than 10% to guarantee the converter is always in discontinuous mode operation (choosing higher D3 allows the use of low inductances, but results in a higher switch peak current). The user can choose a DMAX as the start point. Then, the maximum average primary currents can be calculated by the following equation: ILP(MAX) =ISW(MAX) = POUT(MAX) DMAX • VIN(MIN) • h where h is the converter efficiency. If the flyback converter has multiple outputs, POUT(MAX) is the sum of all the output power. The maximum average secondary current is: ILS(MAX) =ID(MAX) = IOUT(MAX) D2 where: D2 = 1 – DMAX – D3 the primary and secondary RMS currents are: ILP(RMS) = 2 •ILP(MAX) • DMAX 3 ILS(RMS) = 2 •ILS(MAX) • D2 3 According to Figure 8, the primary and secondary peak currents are: ILP(PEAK) = ISW(PEAK) = 2 • ILP(MAX) ILS(PEAK) = ID(PEAK) = 2 • ILS(MAX) The primary and second inductor values of the flyback converter transformer can be determined using the following equations: LP = D2 MAX • V2 IN(MIN) • h 2 • POUT(MAX) • f LS = D22 • (VOUT + VD) 2 • IOUT(MAX) • f The primary to second turns ratio is: NP NS = LP LS Flyback Converter: Snubber Design Transformer leakage inductance (on either the primary or secondary) causes a voltage spike to occur after the MOSFET turn-off. This is increasingly prominent at higher load currents, where more stored energy must be dissipated. In some cases a snubber circuit will be required to avoid overvoltage breakdown at the MOSFET’s drain node. There are different snubber circuits, and Application Note 19 is a good reference on snubber design. An RCD snubber is shown in Figure 7. The snubber resistor value (RSN) can be calculated by the following equation: RSN = 2 • V2 SN − VSN • VOUT • NP NS I2 SW(PEAK) •LLK • f LT3757/LT3757A 20 3757afd Applications Information where VSN is the snubber capacitor voltage. A smaller VSN results in a larger snubber loss. A reasonable VSN is 2 to 2.5 times of: VOUT •NP NS LLK is the leakage inductance of the primary winding, which is usually specified in the transformer characteristics. LLK can be obtained by measuring the primary inductance with the secondary windings shorted. The snubber capacitor value (CCN) can be determined using the following equation: CCN = VSN ΔVSN •RCN • f where ΔVSN is the voltage ripple across CCN. A reasonable ΔVSN is 5% to 10% of VSN. The reverse voltage rating of DSN should be higher than the sum of VSN and VIN(MAX). Flyback Converter: Sense Resistor Selection In a flyback converter, when the power switch is turned on, the current flowing through the sense resistor (ISENSE) is: ISENSE = ILP Set the sense voltage at ILP(PEAK) to be the minimum of the SENSE current limit threshold with a 20% margin. The sense resistor value can then be calculated to be: RSENSE = 80mV ILP(PEAK) Flyback Converter: Power MOSFET Selection For the flyback configuration, the MOSFET is selected with a VDC rating high enough to handle the maximum VIN, the reflected secondary voltage and the voltage spike due to the leakage inductance. Approximate the required MOSFET VDC rating using: BVDSS > VDS(PEAK) where: VDS(PEAK) = VIN(MAX) + VSN The power dissipated by the MOSFET in a flyback converter is: PFET = I2 M(RMS) • RDS(ON) + 2 • V2 DS(PEAK) • IL(MAX) • CRSS • f /1A The first term in this equation represents the conduction losses in the device, and the second term, the switching loss. CRSS is the reverse transfer capacitance, which is usually specified in the MOSFET characteristics. From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following equation: TJ = TA + PFET • θJA = TA + PFET • (θJC + θCA) TJ must not exceed the MOSFET maximum junction temperature rating. It is recommended to measure the MOSFET temperature in steady state to ensure that absolute maximum ratings are not exceeded. LT3757/LT3757A 21 3757afd Applications Information Flyback Converter: Output Diode Selection The output diode in a flyback converter is subject to large RMS current and peak reverse voltage stresses. A fast switching diode with a low forward drop and a low reverse leakage is desired. Schottky diodes are recommended if the output voltage is below 100V. Approximate the required peak repetitive reverse voltage rating VRRM using: VRRM > NS NP • VIN(MAX) + VOUT The power dissipated by the diode is: PD = IO(MAX) • VD and the diode junction temperature is: TJ = TA + PD • RθJA The RθJA to be used in this equation normally includes the RθJC for the device, plus the thermal resistance from the board to the ambient temperature in the enclosure. TJ must not exceed the diode maximum junction temperature rating. Flyback Converter: Output Capacitor Selection The output capacitor of the flyback converter has a similar operation condition as that of the boost converter. Refer to the Boost Converter: Output Capacitor Selection section for the calculation of COUT and ESRCOUT. The RMS ripple current rating of the output capacitors in discontinuous operation can be determined using the following equation: IRMS(COUT),DISCONTINUOUS ≥ IO(MAX) • 4 − (3 •D2) 3 •D2 Flyback Converter: Input Capacitor Selection The input capacitor in a flyback converter is subject to a large RMS current due to the discontinuous primary current. To prevent large voltage transients, use a low ESR input capacitor sized for the maximum RMS current. The RMS ripple current rating of the input capacitors in discontinuous operation can be determined using the following equation: IRMS(CIN),DISCONTINUOUS ≥ POUT(MAX) VIN(MIN) • h • 4 − (3 •DMAX ) 3 •DMAX SEPIC Converter Applications The LT3757 can be configured as a SEPIC (single-ended primary inductance converter), as shown in Figure 1. This topology allows for the input to be higher, equal, or lower than the desired output voltage. The conversion ratio as a function of duty cycle is: VOUT + VD VIN = D 1−D in continuous conduction mode (CCM). In a SEPIC converter, no DC path exists between the input and output. This is an advantage over the boost converter for applications requiring the output to be disconnected from the input source when the circuit is in shutdown. Compared to the flyback converter, the SEPIC converter has the advantage that both the power MOSFET and the output diode voltages are clamped by the capacitors (CIN, CDC and COUT), therefore, there is less voltage ringing across the power MOSFET and the output diodes. The SEPIC converter requires much smaller input capacitors than those of the flyback converter. This is due to the fact that, in the SEPIC converter, the inductor L1 is in series with the input, and the ripple current flowing through the input capacitor is continuous. LT3757/LT3757A 22 3757afd Applications Information Figure 9. The Switch Current Waveform of the SEPIC Converter 3757 F08 ΔISW = χ • ISW(MAX) ISW DT t S ISW(MAX) TS SEPIC Converter: Switch Duty Cycle and Frequency For a SEPIC converter operating in CCM, the duty cycle of the main switch can be calculated based on the output voltage (VOUT), the input voltage (VIN) and the diode forward voltage (VD). The maximum duty cycle (DMAX) occurs when the converter has the minimum input voltage: DMAX = VOUT + VD VIN(MIN) + VOUT + VD SEPIC Converter: Inductor and Sense Resistor Selection As shown in Figure 1, the SEPIC converter contains two inductors: L1 and L2. L1 and L2 can be independent, but can also be wound on the same core, since identical voltages are applied to L1 and L2 throughout the switching cycle. For the SEPIC topology, the current through L1 is the converter input current. Based on the fact that, ideally, the output power is equal to the input power, the maximum average inductor currents of L1 and L2 are: IL1(MAX) = IIN(MAX) = IO(MAX) • DMAX 1− DMAX IL2(MAX) = IO(MAX) In a SEPIC converter, the switch current is equal to IL1 + IL2 when the power switch is on, therefore, the maximum average switch current is defined as: ISW(MAX) =IL1(MAX) +IL2(MAX) =IO(MAX) • 1 1−DMAX and the peak switch current is: ISW(PEAK) = 1+ c 2 •IO(MAX) • 1 1−DMAX The constant c in the preceding equations represents the percentage peak-to-peak ripple current in the switch, relative to ISW(MAX), as shown in Figure 9. Then, the switch ripple current ΔISW can be calculated by: ΔISW = c • ISW(MAX) The inductor ripple currents ΔIL1 and ΔIL2 are identical: ΔIL1 = ΔIL2 = 0.5 • ΔISW The inductor ripple current has a direct effect on the choice of the inductor value. Choosing smaller values of ΔIL requires large inductances and reduces the current loop gain (the converter will approach voltage mode). Accepting larger values of ΔIL allows the use of low inductances, but results in higher input current ripple and greater core losses. It is recommended that c falls in the range of 0.2 to 0.4. LT3757/LT3757A 23 3757afd Given an operating input voltage range, and having chosen the operating frequency and ripple current in the inductor, the inductor value (L1 and L2 are independent) of the SEPIC converter can be determined using the following equation: L1=L2 = VIN(MIN) 0.5 • ΔISW • f •DMAX For most SEPIC applications, the equal inductor values will fall in the range of 1μH to 100μH. By making L1 = L2, and winding them on the same core, the value of inductance in the preceding equation is replaced by 2L, due to mutual inductance: L = VIN(MIN) ΔISW • f •DMAX This maintains the same ripple current and energy storage in the inductors. The peak inductor currents are: IL1(PEAK) = IL1(MAX) + 0.5 • ΔIL1 IL2(PEAK) = IL2(MAX) + 0.5 • ΔIL2 The RMS inductor currents are: IL1(RMS) =IL1(MAX) • 1+ c2 L1 12 where: cL1 = ΔIL1 IL1(MAX) IL2(RMS) =IL2(MAX) • 1+ c2 L2 12 where: cL2 = ΔIL2 IL2 (MAX) Based on the preceding equations, the user should choose the inductors having sufficient saturation and RMS current ratings. In a SEPIC converter, when the power switch is turned on, the current flowing through the sense resistor (ISENSE) is the switch current. Set the sense voltage at ISENSE(PEAK) to be the minimum of the SENSE current limit threshold with a 20% margin. The sense resistor value can then be calculated to be: RSENSE = 80mV ISW(PEAK) SEPIC Converter: Power MOSFET Selection For the SEPIC configuration, choose a MOSFET with a VDC rating higher than the sum of the output voltage and input voltage by a safety margin (a 10V safety margin is usually sufficient). The power dissipated by the MOSFET in a SEPIC converter is: PFET = I2 SW(MAX) • RDS(ON) • DMAX + 2 • (VIN(MIN) + VOUT)2 • IL(MAX) • CRSS • f /1A The first term in this equation represents the conduction losses in the device, and the second term, the switching loss. CRSS is the reverse transfer capacitance, which is usually specified in the MOSFET characteristics. For maximum efficiency, RDS(ON) and CRSS should be minimized. From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following equation: TJ = TA + PFET • θJA = TA + PFET • (θJC + θCA) TJ must not exceed the MOSFET maximum junction temperature rating. It is recommended to measure the MOSFET temperature in steady state to ensure that absolute maximum ratings are not exceeded. Applications Information LT3757/LT3757A 24 3757afd Applications Information Figure 10. A Simplified Inverting Converter RSENSE CDC VIN CIN L1 D1 COUT VOUT 3757 F09 GATE + GND LT3757 SENSE L2 M1 + – + – + SEPIC Converter: Output Diode Selection To maximize efficiency, a fast switching diode with a low forward drop and low reverse leakage is desirable. The average forward current in normal operation is equal to the output current, and the peak current is equal to: ID(PEAK) = 1+ c 2 •IO(MAX) • 1 1−DMAX It is recommended that the peak repetitive reverse voltage rating VRRM is higher than VOUT + VIN(MAX) by a safety margin (a 10V safety margin is usually sufficient). The power dissipated by the diode is: PD = IO(MAX) • VD and the diode junction temperature is: TJ = TA + PD • RθJA The RθJA used in this equation normally includes the RθJC for the device, plus the thermal resistance from the board, to the ambient temperature in the enclosure. TJ must not exceed the diode maximum junction temperature rating. SEPIC Converter: Output and Input Capacitor Selection The selections of the output and input capacitors of the SEPIC converter are similar to those of the boost converter. Please refer to the Boost Converter, Output Capacitor Selection and Boost Converter, Input Capacitor Selection sections. SEPIC Converter: Selecting the DC Coupling Capacitor The DC voltage rating of the DC coupling capacitor (CDC, as shown in Figure 1) should be larger than the maximum input voltage: VCDC > VIN(MAX) CDC has nearly a rectangular current waveform. During the switch off-time, the current through CDC is IIN, while approximately –IO flows during the on-time. The RMS rating of the coupling capacitor is determined by the following equation: IRMS(CDC) > IO(MAX) • VOUT + VD VIN(MIN) A low ESR and ESL, X5R or X7R ceramic capacitor works well for CDC. Inverting Converter Applications The LT3757 can be configured as a dual-inductor inverting topology, as shown in Figure 10. The VOUT to VIN ratio is: VOUT − VD VIN = − D 1−D in continuous conduction mode (CCM). LT3757/LT3757A 25 3757afd Inverting Converter: Switch Duty Cycle and Frequency For an inverting converter operating in CCM, the duty cycle of the main switch can be calculated based on the negative output voltage (VOUT) and the input voltage (VIN). The maximum duty cycle (DMAX) occurs when the converter has the minimum input voltage: DMAX = VOUT − VD VOUT − VD − VIN(MIN) Inverting Converter: Inductor, Sense Resistor, Power MOSFET, Output Diode and Input Capacitor Selections The selections of the inductor, sense resistor, power MOSFET, output diode and input capacitor of an inverting converter are similar to those of the SEPIC converter. Please refer to the corresponding SEPIC converter sections. Inverting Converter: Output Capacitor Selection The inverting converter requires much smaller output capacitors than those of the boost, flyback and SEPIC converters for similar output ripples. This is due to the fact that, in the inverting converter, the inductor L2 is in series with the output, and the ripple current flowing through the output capacitors are continuous. The output ripple voltage is produced by the ripple current of L2 flowing through the ESR and bulk capacitance of the output capacitor: ΔVOUT(P–P) = ΔIL2 • ESRCOUT + 1 8 • f •COUT After specifying the maximum output ripple, the user can select the output capacitors according to the preceding equation. The ESR can be minimized by using high quality X5R or X7R dielectric ceramic capacitors. In many applications, ceramic capacitors are sufficient to limit the output voltage ripple. The RMS ripple current rating of the output capacitor needs to be greater than: IRMS(COUT) > 0.3 • ΔIL2 Inverting Converter: Selecting the DC Coupling Capacitor The DC voltage rating of the DC coupling capacitor (CDC, as shown in Figure 10) should be larger than the maximum input voltage minus the output voltage (negative voltage): VCDC > VIN(MAX) – VOUT CDC has nearly a rectangular current waveform. During the switch off-time, the current through CDC is IIN, while approximately –IO flows during the on-time. The RMS rating of the coupling capacitor is determined by the following equation: IRMS(CDC) >IO(MAX) • DMAX 1−DMAX A low ESR and ESL, X5R or X7R ceramic capacitor works well for CDC. Applications Information LT3757/LT3757A 26 3757afd Applications Information Figure 11. 8V to 16V Input, 24V/2A Output Boost Converter Suggested Layout VIN 3757 F10 VOUT L1 VIAS TO GROUND PLANE C D1 COUT2 OUT1 1 2 8 7 3 4 6 5 M1 CIN R4 RC R1 R2 RSS RT R3 CVCC CC1 CC2 LT3757 1 2 3 4 5 9 10 6 7 8 RS Board Layout The high speed operation of the LT3757 demands careful attention to board layout and component placement. The Exposed Pad of the package is the only GND terminal of the IC, and is important for thermal management of the IC. Therefore, it is crucial to achieve a good electrical and thermal contact between the Exposed Pad and the ground plane of the board. For the LT3757 to deliver its full output power, it is imperative that a good thermal path be provided to dissipate the heat generated within the package. It is recommended that multiple vias in the printed circuit board be used to conduct heat away from the IC and into a copper plane with as much area as possible. To prevent radiation and high frequency resonance problems, proper layout of the components connected to the IC is essential, especially the power paths with higher di/ dt. The following high di/dt loops of different topologies should be kept as tight as possible to reduce inductive ringing: • In boost configuration, the high di/dt loop contains the output capacitor, the sensing resistor, the power MOSFET and the Schottky diode. • In flyback configuration, the high di/dt primary loop contains the input capacitor, the primary winding, the power MOSFET and the sensing resistor. The high di/ dt secondary loop contains the output capacitor, the secondary winding and the output diode. • In SEPIC configuration, the high di/dt loop contains the power MOSFET, sense resistor, output capacitor, Schottky diode and the coupling capacitor. • In inverting configuration, the high di/dt loop contains power MOSFET, sense resistor, Schottky diode and the coupling capacitor. LT3757/LT3757A 27 3757afd Table 2. Recommended Component Manufacturers VENDOR COMPONENTS WEB ADDRESS AVX Capacitors avx.com BH Electronics Inductors, Transformers bhelectronics.com Coilcraft Inductors coilcraft.com Cooper Bussmann Inductors bussmann.com Diodes, Inc Diodes diodes.com Fairchild MOSFETs fairchildsemi.com General Semiconductor Diodes generalsemiconductor.com International Rectifier MOSFETs, Diodes irf.com IRC Sense Resistors irctt.com Kemet Capacitors kemet.com Magnetics Inc Toroid Cores mag-inc.com Microsemi Diodes microsemi.com Murata-Erie Inductors, Capacitors murata.co.jp Nichicon Capacitors nichicon.com On Semiconductor Diodes onsemi.com Panasonic Capacitors panasonic.com Sanyo Capacitors sanyo.co.jp Sumida Inductors sumida.com Taiyo Yuden Capacitors t-yuden.com TDK Capacitors, Inductors component.tdk.com Thermalloy Heat Sinks aavidthermalloy.com Tokin Capacitors nec-tokinamerica.com Toko Inductors tokoam.com United Chemi-Con Capacitors chemi-con.com Vishay/Dale Resistors vishay.com Vishay/Siliconix MOSFETs vishay.com Vishay/Sprague Capacitors vishay.com Würth Elektronik Inductors we-online.com Zetex Small-Signal Discretes zetex.com Applications Information Check the stress on the power MOSFET by measuring its drain-to-source voltage directly across the device terminals (reference the ground of a single scope probe directly to the source pad on the PC board). Beware of inductive ringing, which can exceed the maximum specified voltage rating of the MOSFET. If this ringing cannot be avoided, and exceeds the maximum rating of the device, either choose a higher voltage device or specify an avalancherated power MOSFET. The small-signal components should be placed away from high frequency switching nodes. For optimum load regulation and true remote sensing, the top of the output voltage sensing resistor divider should connect independently to the top of the output capacitor (Kelvin connection), staying away from any high dV/dt traces. Place the divider resistors near the LT3757 in order to keep the high impedance FBX node short. Figure 11 shows the suggested layout of the 8V to 16V Input, 24V/2A Output Boost Converter. Recommended Component Manufacturers Some of the recommended component manufacturers are listed in Table 2. LT3757/LT3757A 28 3757afd Typical Applications 3.3V Input, 5V/10A Output Boost Converter Boost Preregulator for Automotive Stop-Start/Idle Efficiency vs Output Current Transient VIN and VOUT Waveforms SENSE LT3757 VIN VIN 3.3V CIN 22μF 6.3V ×2 VOUT 5V 10A 0.004 1W M1 41.2k 300kHz GATE FBX GND INTVCC SHDN/UVLO SYNC RT SS VC 49.9k 34k 0.1μF 6.8k 22nF 2.2nF 22 L1 0.5μH D1 3757 TA02a 34k 1% 15.8k 1% COUT1 150μF 6.3V ×4 COUT2 22μF 6.3V X5R ×4 + CVCC 4.7μF 10V X5R CIN: TAIYO YUDEN JMK325BJ226MM COUT1: PANASONIC EEFUEOJ151R COUT2: TAIYO YUDEN JMK325BJ226MM D1: MBRB2515L L1: VISHAY SILICONIX IHLP-5050FD-01 M1: VISHAY SILICONIX SI4448DY OUTPUT CURRENT (A) EFFICIENCY (%) 3757 TA02b 0.001 20 30 40 50 60 70 80 90 100 0.01 0.1 1 10 SENSE LT3757A VIN VIN 3V TO 36V 10μF 50V X5R ×2 VOUT 9VMIN 2A 41.2k 300kHz GATE FBX GND INTVCC SHDN/UVLO SYNC RT SS VC 1M 698k 0.1μF 10k 10nF 4.7μF L1 3.3μH D1 3757 TA03a M1 75k 8m 16.2k C1 10μF 50V ×4 + 10μF 50V X5R L1: COILTRONIX DR127-3R3 M1: VISHAY SILICONIX Si7848BDP D1: VISHAY SILICONIX 50SQ04FN C1: KEMET T495X106K050A 10ms/DIV VOUT 5V/DIV VIN 5V/DIV 0V 3757 TA03b OUTPUT POWER = 10W LT3757/LT3757A 29 3757afd Typical Applications 8V to 16V Input, 24V/2A Output Boost Converter Efficiency vs Output Current Load Step Response at VIN = 12V SENSE LT3757 VIN VIN 8V TO 16V CIN 10μF 25V X5R CVCC 4.7μF 10V X5R VOUT 24V 2A RS 0.01 1W M1 RT 41.2k 300kHz GATE FBX GND INTVCC SHDN/UVLO SYNC RT SS VC R3 200k R4 43.2k CSS 0.1μF CC2 100pF RC 22k CC1 6.8nF L1 10μH D1 3757 TA04a R2 226k 1% R1 16.2k 1% COUT1 47μF 35V ×4 COUT2 10μF 25V X5R + CIN, COUT2: MURATA GRM31CR61E106KA12 COUT1: KEMET T495X476K035AS D1: ON SEMI MBRS340T3G L1: VISHAY SILICONIX IHLP-5050FD-01 10μH M1: VISHAY SILICONIX Si4840BDP OUTPUT CURRENT (A) 0.001 EFFICIENCY (%) 30 50 40 60 70 80 90 100 0.01 0.1 1 3757 TA04b 10 VIN = 8V VIN = 16V 500μs/DIV VOUT 500mV/DIV (AC) 1.6A 0.4A IOUT 1A/DIV 3757 TA04c LT3757/LT3757A 30 3757afd 2ms/DIV VOUT 100V/DIV 3757 TA05b 5μs/DIV VOUT 5V/DIV (AC) VSW 20V/DIV 3757 TA05c Typical Applications High Voltage Flyback Power Supply Start-Up Waveforms Switching Waveforms SENSE LT3757 VIN VSW VIN 5V TO 12V CIN 47μF 16V ×4 INTVCC COUT 68nF ×2 VOUT 350V 10mA 0.02 22 M1 140k 100kHz GATE GND FBX SHDN/UVLO DANGER! HIGH VOLTAGE OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY SYNC RT SS VC • 105k • 46.4k 0.1μF 220pF 100pF 6.8k 22nF T1 1:10 D1 CIN: MURATA GRM32ER61C476K COUT: TDK C3225X7R2J683K D1: VISHAY SILICONIX GSD2004S DUAL DIODE CONNECTED IN SERIES M1: VISHAY SILICONIX Si7850DP T1: TDK DCT15EFD-U44S003 3757 TA05a 1M 1% 1M 1% 1.50M 1% 16.2k 1% 10nF CVCC 47μF 25V X5R 22 LT3757/LT3757A 31 3757afd Typical Applications 5.5V to 36V Input, 12V/2A Output SEPIC Converter Efficiency vs Output Current Load Step Waveforms Start-Up Waveforms Frequency Foldback Waveforms When Output Short-Circuits SENSE LT3757A VIN VIN 5.5V TO 36V CIN1 4.7μF 50V ×2 CDC 4.7μF 50V, X5R, ×2 4.7μF 10V X5R VOUT 12V 2A 0.01 1W M1 41.2k 300kHz GATE FBX GND INTVCC SHDN/UVLO SYNC RT SS CIN2 4.7μF 50V ×2 • • 105k 46.4k 0.1μF 6.8nF 10k L1A IL1B L1B D1 CIN1, CDC: TAIYO YUDEN UMK316BJ475KL CIN2: KEMET T495X475K050AS COUT1: KEMET T495X476K020AS COUT2: TAIYO YUDEN TMK432BJ106MM D1: ON SEMI MBRS360T3G L1A, L1B: COILTRONICS DRQ127-4R7 (*COUPLED INDUCTORS) M1: VISHAY SILICONIX Si7460DP 3757 TA06a 105k 1% 15.8k 1% COUT1 47μF 20V ×4 COUT2 10μF 25V X5R + VSW IL1A VC + 2ms/DIV VOUT 5V/DIV IL1A + IL1B 5A/DIV 3757 TA06d VIN = 12V 50μs/DIV VOUT 10V/DIV VSW 20V/DIV IL1A + IL1B 5A/DIV 3757 TA06e VIN = 12V OUTPUT CURRENT (A) 0.001 20 EFFICIENCY (%) 30 40 50 60 70 80 90 100 0.01 0.1 1 3757 TA06b 10 VIN = 16V VIN = 8V 500μs/DIV VOUT 2V/DIV AC-COUPLED IOUT 2A/DIV 0A 2A 3757 TA06c VIN = 12V LT3757/LT3757A 32 3757afd Typical Applications 5V to 12V Input, ±12V/0.4A Output SEPIC Converter Nonisolated Inverting SLIC Supply SENSE LT3757 VIN VIN 5V TO 12V CIN1 1μF 16V, X5R CIN2 47μF 16V CDC1 4.7μF 16V, X5R CDC2 4.7μF 16V X5R COUT2 4.7μF 16V, X5R ×3 VOUT1 12V 0.4A VOUT2 –12V 0.4A COUT2 4.7μF 16V, X5R ×3 CVCC 4.7μF 10V X5R 0.02 M1 30.9k 400kHz D1, D2: MBRS140T3 T1: COILTRONICS VP1-0076 (*PRIMARY = 4 WINDINGS IN PARALLEL) M1: SILICONIX/VISHAY Si4840BDY GATE FBX GND INTVCC SHDN/UVLO SYNC RT SS + 105k • 46.4k 0.1μF 100pF 22k 6.8nF T1 1,2,3,4 D1 GND 1.05k 1% 158 1% D2 5 6 • • 3757 TA07 VC SENSE LT3757 VIN VIN 5V TO 16V CIN 22μF 25V, X5R ×2 C2 10μF 50V X5R D1 DFLS160 CVCC 4.7μF 10V, X5R C3 22μF 25V X5R C4 22μF 25V X5R COUT 3.3μF 100V GND C5 22μF 25V X5R VOUT1 –24V 200mA VOUT1 –72V 200mA 0.012 0.5W M1 Si7850DP 63.4k 200kHz GATE FBX GND INTVCC SHDN/UVLO SYNC RT SS • • • R2 • 105k R1 46.4k 0.1μF 100pF 15.8k 464k 9.1k 10nF T1 1,2,3 4 D2 DFLS160 5 D3 DFLS160 6 VP5-0155 (PRIMARY = 3 WINDINGS IN PARALLEL) 3757 TA08 VC LT3757/LT3757A 33 3757afd Package Description 3.00 ±0.10 (4 SIDES) NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 0.40 ± 0.10 BOTTOM VIEW—EXPOSED PAD 1.65 ± 0.10 (2 SIDES) 0.75 ±0.05 R = 0.125 TYP 2.38 ±0.10 (2 SIDES) 5 1 6 10 PIN 1 TOP MARK (SEE NOTE 6) 0.200 REF 0.00 – 0.05 (DD) DFN REV C 0310 0.25 ± 0.05 2.38 ±0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS 1.65 ±0.05 2.15 ±0.05 (2 SIDES) 0.50 BSC 0.70 ±0.05 3.55 ±0.05 PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC DD Package 10-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1699 Rev C) PIN 1 NOTCH R = 0.20 OR 0.35 × 45° CHAMFER LT3757/LT3757A 34 3757afd Package Description MSOP (MSE) 0911 REV H 0.53 ±0.152 (.021 ±.006) SEATING PLANE 0.18 (.007) 1.10 (.043) MAX 0.17 – 0.27 (.007 – .011) TYP 0.86 (.034) REF 0.50 (.0197) BSC 1 2 3 4 5 4.90 ±0.152 (.193 ±.006) 0.497 ±0.076 (.0196 ±.003) REF 10 9 8 10 1 7 6 3.00 ±0.102 (.118 ±.004) (NOTE 3) 3.00 ±0.102 (.118 ±.004) (NOTE 4) NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 0.254 (.010) 0° – 6° TYP DETAIL “A” DETAIL “A” GAUGE PLANE 5.23 (.206) MIN 3.20 – 3.45 (.126 – .136) 0.889 ±0.127 (.035 ±.005) RECOMMENDED SOLDER PAD LAYOUT 1.68 ±0.102 (.066 ±.004) 1.88 ±0.102 (.074 ±.004) 0.50 (.0197) BSC 0.305 ± 0.038 (.0120 ±.0015) TYP BOTTOM VIEW OF EXPOSED PAD OPTION 1.68 (.066) 1.88 (.074) 0.1016 ±0.0508 (.004 ±.002) DETAIL “B” DETAIL “B” CORNER TAIL IS PART OF THE LEADFRAME FEATURE. FOR REFERENCE ONLY NO MEASUREMENT PURPOSE 0.05 REF 0.29 REF MSE Package 10-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1664 Rev H) LT3757/LT3757A 35 3757afd Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. Revision History REV DATE DESCRIPTION PAGE NUMBER B 3/10 Deleted Bullet from Features and Last Line of Description Updated Entire Page to Add H-Grade and Military Grade Updated Electrical Characteristics Notes and Typical Performance Characteristics for H-Grade and Military Grade Revised TA04a and Replaced TA04c in Typical Applications Updated Related Parts 1 2 4 to 6 30 36 C 5/11 Revised MP-grade temperature range in Absolute Maximum Ratings and Order Information sections Revised Note 2 Revised formula in Applications Information Updated Typical Application drawing TA04a values Revised Typical Application title TA06 2 4 19 30 32 D 07/12 Added LT3757A version Throughout Updated Block Diagram 8 Updated Programming the Output Voltage section 12 Updated Loop Compensation section 13 Added an application circuit in the Typical Applications section 28 Updated the schematic and Load Step Waveforms in the Typical Applications section 31 (Revision history begins at Rev B) LT3757/LT3757A 36 3757afd Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2008 LT 0712 REV D • PRINTED IN USA Related Parts Typical Application PART NUMBER DESCRIPTION COMMENTS LT3758A Boost, Flyback, SEPIC and Inverting Controller 5.5V ≤ VIN ≤ 100V, Current Mode Control, 100kHz to 1MHz Programmable Operation Frequency, 3mm × 3mm DFN-10 and MSOP-10E Packages LT3759 Boost, SEPIC and Inverting Controller 1.6V ≤ VIN ≤ 42V, Current Mode Control, 100kHz to 1MHz Programmable Operation Frequency, MSOP-12E Packages LT3957A Boost, Flyback, SEPIC and Inverting Controller with 5A, 40V Switch 3V ≤ VIN ≤ 40V, Current Mode Control, 100kHz to 1MHz Programmable Operation Frequency, 5mm × 6mm QFN Package LT3958 Boost, Flyback, SEPIC and Inverting Controller with 3.3A, 84V Switch 5V ≤ VIN ≤ 80V, Current Mode Control, 100kHz to 1MHz Programmable Operation Frequency, 5mm × 6mm QFN Package LT3573/LT3574/ LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch LT3798 Offline Isolated No Opto-Coupler Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components, MSOP-16 Package LT3799/LT3799-1 Offline Isolated Flyback LED Controllers with Active PFC VIN and VOUT Limited Only by External Components, MSOP-16 Package High Efficiency Inverting Power Supply Efficiency vs Output Current OUTPUT CURRENT (A) 0.001 10 EFFICIENCY (%) 20 30 40 50 60 70 80 90 100 0.01 0.1 1 3757 TA09b 10 VIN = 16V VIN = 5V SENSE LT3757 VIN VIN 5V TO 15V CIN 47μF 16V X5R CDC 47μF 25V, X5R VOUT –5V 3A to 5A 0.006 1W M1 Si7848BDP 41.2k 300kHz GATE FBX GND INTVCC SHDN/UVLO SYNC RT SS • R2 • 105k R1 46.4k 0.1μF 9.1k 10nF L1 L2 D1 MBRD835L L1, L2: COILTRONICS DRQ127-3R3 (*COUPLED INDUCTORS) 3757 TA09a 84.5k CVCC 16k 4.7μF 10V X5R COUT 100μF 6.3V, X5R ×2 VC Photoelectric proximity switches HGA Photoelectric proximity switches ener. Photoelectric proximity switches V Photoelectric reflex switches T W 9-2: A Versatile, Complete and Compact Series D A T A S H E E T The W 9-2 series is as versatile as the tasks in automation. The standardized, compact housing model makes it possible to use high-performance sensors that operate reliably even in cramped mounting conditions. All W 9-2 models have red light transmitters as a standard feature. The sensor can be aligned on the object quickly and precisely using the visible light spot. In the models with Teach-In function, the sensor optimizes its sensitivity automatically to the given operating conditions at the push of a button. Depending on the job, the most suitable sensor can be selected from the W 9-2 series. Overview of the sensors: WT 9-2, with adjustable background suppression, max. scanning distance 250 mm, WT 9-2, energetic, max. scanning distance 450 mm, WT 9-2, V model, max. scanning distance 20 mm, WL 9-2, basic model, max. scanning range 4 m, WL 9-2, Teach-In model, max. scanning range 4 m, WL 9-2, focus, max. scanning range 0.4 m. There are multifaceted applications in the targeted main branches thanks to this great variety of products: Storage and handling engineering Packaging industry Electronics industry Elevator construction. 2 SENSICK WT 9-2 Photoelectric Proximity Switch with Background Suppression Setting options Dimension illustration LED light source, visible red light Background suppression Scanning distance adjustable Switching frequency 1500/s Outputs short-circuit protected Scanning distance 30 ... 250 mm 12 22 40 20 3 3 1.5 1 3 2 3 18.5 10.5 11 4 5 7 Axis of the sender optics Axis of the receiver optics Mounting hole Ø 3.2 mm LED signal strength indicator Plug M 12 or M 8, 4 pin, 2 m connection cable or 120 mm cable with plug M 12, 4 pin Scanning distance adjuster Standard direction of the material to be scanned 1 2 3 4 5 Photoelectric proximity switch 6 7 WT 9-2P130 WT 9-2P430 WT 9-2N130 WT 9-2N430 4 6 Cable receptacles Adapter plate Mounting bracket Accessories Connection type L+ Q Q M brn wht blu blk 4 pin, M 12 WT 9-2P330 WT 9-2P630 WT 9-2P430 WT 9-2N430 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 8 1 L+ Q Q 4 2 3 M brn wht blu blk 4 x 0,14 mm2 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 12 with 120 mm cable WT 9-2P330 WT 9-2P130 WT 9-2N130 WT 9-2P630 Scanning distance adjustable 1) 30 ... 250 mm Scanning range 5 ... 250 mm Supply voltage VS 2) DC 10 ... 30 V Ripple 3) ≤ 5 VPP Current consumption 4) ≤ 40 mA Light source LED, visible red light 5) Light spot diameter 15 x 15 mm at a distance of 200 mm Switching outputs Q and Q– PNP NPN Signal voltage HIGH VS – 2.9 V VS Signal voltage LOW 6) Approx. 0 V ≤ 1.5 V Output current IA max. ≤ 100 mA Response time 7) ≤ 333 μs Switching frequency max. 8) 1500/s Connection technology Connection cable, 2 m Cable, 120 mm, with plug M 12, 4 pin Plug M 12, 4 pin Plug M 8, 4 pin VDE protection class M 12 9) VDE protection class M 8 9) III Protection type IP 67 Protection circuits 10) A, B, C Ambient temperature 11) Operation –40 ... +60 °C Storage –40 ... +75 °C Weight with connection cable 2 m/120 mm Approx. 80 g with equipment plug M 12/M 8, 4 pin Approx. 20 g SENSICK 3 WT 9-2 Scanning distance Ordering information Technical data WT 9-2 P130 P430 N130 N430 P330 P630 1) Object with 90% reflectance (referred to standard white DIN 5033) 2) Limit values 3) Must be within VS tolerances 4) Without load 5) Average service life at room temperature 100,000 h 6) At TU = +25 °C and 100 mA output current 7) With resistive load 8) With light/dark ratio 1:1 9) Withstand voltage 50 V 10) A = supply connections reverse polarity protected B = outputs short-circuit protected C = interference suppression 11) Do not distort cable below 0 °C Type WT 9-2P130 WT 9-2P430 WT 9-2N130 WT 9-2N430 WT 9-2P330 WT 9-2P630 Order no. 1 018 293 1 018 295 1 018 294 1 018 296 1 019 026 1 019 272 (mm) 50 100 150 200 250 30 15 20 25 10 0 5 % of scanning distance 1 3 2 WT 9-2 HGA 90%/90% 18%/90% 6%/90% Scanning range on gray, white background, Black = 6% reflectance 1 Scanning range on black ), white background, 2 White = 90% reflectance Scanning range on white, white background, Gray = 18% reflectance 3 0(mm) 50 100 150 200 250 3 1 2 Operating distance 30 150 30 220 30 250 4 SENSICK WT 9-2 Photoelectric Proximity Switch, Energetic, Teach-In Setting options Dimension illustration Red-light emitter LED as alignment aid Scanning distance adjustable Switching frequency 800/s Outputs short-circuit protected Teach-In Scanning distance 18 ... 450 mm 12 22 40 20 3 3 1.5 1 3 2 3 25.55 6.5 11 4 5 Axis of the receiver optics Axis of the sender optics Mounting hole Ø 3.2 mm LED signal strength indicator Plug M 12 or M 8, 4 pin, 2 m connection cable or 120 mm cable with plug M 12, 4 pin Scanning distance adjuster, teachable 1 2 3 4 5 Photoelectric proximity switch 6 WT 9-2P151 WT 9-2P451 WT 9-2N151 WT 9-2N451 4 6 Cable receptacles Adapter plate Mounting bracket Accessories Connection type L+ Q Q M brn wht blu blk 4 pin, M 12 WT 9-2P351 WT 9-2P651 WT 9-2P451 WT 9-2N451 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 8 1 L+ Q Q 4 2 3 M brn wht blu blk 4 x 0,14 mm2 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 12 with 120 mm cable WT 9-2P351 WT 9-2P151 WT 9-2N151 WT 9-2P651 SENSICK 5 Scanning distance adjustable 1) 10 ... 450 mm Supply voltage VS 2) DC 10 ... 30 V Ripple 3) ≤ 5 VPP Current consumption 4) ≤ 30 mA Light source LED, visible red light 5) Light spot diameter 80 x 80 mm at a distance of 500 mm Switching outputs Q and Q– PNP NPN Signal voltage HIGH VS – 2.9 V VS Signal voltage LOW6) Approx. 0 V ≤ 2.9 V Output current IA max. ≤ 100 mA Response time 7) ≤ 625 μs Switching frequency max. 8) 800/s Connection technology Connection cable, 2 m Cable, 120 mm, with plug M 12, 4 pin Plug M 12, 4 pin Plug M 8, 4 pin VDE protection class M 12 9) VDE protection class M 8 9) III Protection type IP 67 Protection circuits 10) A, B, C Ambient temperature 11) Operation –40 ... +60 °C Storage –40 ... +75 °C Weight with connection cable 2 m/120 mm Approx. 80 g with equipment plug M 12/M 8, 4 pin Approx. 20 g WT 9-2 Scanning distance Ordering information Technical data WT 9-2 P151 P451 N151 N451 P351 P651 1) Object with 90% reflectance (referred to standard white DIN 5033) 2) Limit values 3) Must be within VS tolerances 4) Without load 5) Average service life at room temperature 50,000 h 6) At TU = +25 °C and 100 mA output current 7) With resistive load 8) With light/dark ratio 1:1 9) Withstand voltage 50 V 10) A = supply connections reverse polarity protected B = outputs short-circuit protected C = interference suppression 11) Do not distort cable below 0 °C Type WT 9-2P151 WT 9-2P451 WT 9-2N151 WT 9-2N451 WT 9-2P351 WT 9-2P651 Order no. 1 018 297 1 018 299 1 018 298 1 018 300 1 019 027 1 019 273 (mm) 100 200 1000 10 100 1 300 400 500 Function reserve Operating distance Limiting scanning distance WT 9-2 energetic 3 90% 2 18% 1 6% Programming via Teach-In button. Simple programming: Position object in the beam and push the button: finished; LED confirms the Teach-In procedure. Teach-In values can be stored. Teach-In function Two operating modes: Default setting: short Teach-In time (< 8 s); for standard applications; approx. double reserve via switching threshold; LED lights continuously. Precise setting: long Teach-In time (> 8 s); for precise applications; small switching hysteresis; LED blinks. Scanning range on white, 90 % reflectance Scanning range on gray, 18% reflectance 1 Scanning range on black, 6% reflectance 2 3 0(mm) 100 200 300 400 500 1 2 3 Operating distance Limiting scanning distance 10 180 220 10/100 130 10 350 450 6 SENSICK WT 9-2 Photoelectric Proximity Switch, V-type, Teach-In Setting options Dimension illustration Red-light emitter LED as alignment aid Scanning distance adjustable Switching frequency 800/s Outputs short-circuit protected Teach-In Scanning distance 10 ... 20 mm 12 22 40 20 3 3 1.5 1 3 2 3 26.45 4.7 11 4 5 Axis of the receiver optics Axis of the receiver optics Mounting hole Ø 3.2 mm LED signal strength indicator Plug M 12 or M 8, 4 pin, 2 m connection cable or 120 mm cable with plug M 12, 4 pin Scanning distance adjuster, teachable 1 2 3 4 5 Photoelectric proximity switch 6 WT 9-2P141 WT 9-2P441 WT 9-2N141 WT 9-2N441 4 6 Cable receptacles Adapter plate Mounting bracket Accessories Connection type L+ Q Q M brn wht blu blk 4 pin, M 12 WT 9-2P341 WT 9-2P641 WT 9-2P441 WT 9-2N441 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 8 1 L+ Q Q 4 2 3 M brn wht blu blk 4 x 0,14 mm2 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 12 with 120 mm cable WT 9-2P341 WT 9-2P141 WT 9-2N141 WT 9-2P641 SENSICK 7 Scanning distance adjustable 1) 10 ... 20 mm Supply voltage VS 2) DC 10 ... 30 V Ripple 3) ≤ 5 VPP Current consumption 4) ≤ 30 mA Light source LED, visible red light 5) Light spot diameter 3 mm at a distance of 20 mm Switching outputs Q and Q– PNP NPN Signal voltage HIGH VS – 2.9 V VS Signal voltage LOW6) Approx. 0 V ≤ 2.9 V Output current IA max. ≤ 100 mA Response time 7) ≤ 625 μs Switching frequency max. 8) 800/s Connection technology Connection cable, 2 m Cable, 120 mm, with plug M 12, 4 pin Plug M 12, 4 pin Plug M 8, 4 pin VDE protection class M 12 9) VDE protection class M 8 9) III Protection type IP 67 Protection circuits 10) A, B, C Ambient temperature 11) Operation –40 ... +60 °C Storage –40 ... +75 °C Weight with connection cable 2 m/120 mm Approx. 80 g with equipment plug M 12/M 8, 4 pin Approx. 20 g WT 9-2 Scanning distance Ordering information Technical data WT 9-2 P141 P441 N141 N441 P341 P641 1) Object with 90% reflectance (referred to standard white DIN 5033) 2) Limit values 3) Must be within VS tolerances 4) Without load 5) Average service life at room temperature 100,000 h 6) At TU = +25 °C and 100 mA output current 7) With resistive load 8) With light/dark ratio 1:1 9) Withstand voltage 50 V 10) A = supply connections reverse polarity protected B = outputs short-circuit protected C = interference suppression 11) Do not distort cable below 0 °C Type WT 9-2P141 WT 9-2P441 WT 9-2N141 WT 9-2N441 WT 9-2P341 WT 9-2P641 Order no. 1 018 301 1 018 303 1 018 302 1 018 304 1 019 274 1 019 275 Programming via Teach-In button. Simple programming: Position object in the beam and push the button: finished; LED confirms the Teach-In procedure. Teach-In values can be stored. Teach-In function Two operating modes: Default setting: short Teach-In time (< 8 s); for standard applications; approx. double reserve via switching threshold; LED lights continuously. Precise setting: long Teach-In time (> 8 s); for precise applications; small switching hysteresis; LED blinks. (mm) 4 1 10 100 8 12 16 20 24 28 Function reserve 1 3 2 6% 18% 90% Operating distance WT 9-2 0(mm) 10 20 30 1 2 3 Scanning distance 10 22 10 20 10 24 Scanning range on white, 90 % reflectance Scanning range on gray, 18% reflectance 1 Scanning range on black, 6% reflectance 2 3 8 SENSICK WL 9-2 Photoelectric Reflex Switch, Standard Without setting options Dimension illustration Red-light emitter LED as alignment aid Switching frequency 800/s Outputs short-circuit protected Scanning range 0 ... 4 m 12 22 40 20 3 3 1.5 1 2 2 29.5 11 3 4 Middle of optic axis Mounting hole Ø 3.2 mm LED signal strength indicator Plug M 12 or M 8, 4 pin, 2 m connection cable or 120 mm cable with plug M 12, 4 pin 1 2 3 4 Photoelectric reflex switch WL 9-2P130 WL 9-2P430 WL 9-2N130 WL 9-2N430 3 Cable receptacles Adapter plate Mounting bracket Reflectors Accessories Connection type L+ Q Q M brn wht blu blk 4 pin, M 12 WT 9-2P330 WT 9-2P630 WT 9-2P430 WT 9-2N430 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 8 1 L+ Q Q 4 2 3 M brn wht blu blk 4 x 0,14 mm2 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 12 with 120 mm cable WT 9-2P330 WT 9-2P130 WT 9-2N130 WT 9-2P630 SENSICK 9 Scanning range typ. max./on reflector 4 m/PL 80 A Supply voltage VS 1) DC 10 ... 30 V Ripple 2) ≤ 5 VPP Current consumption 3) ≤ 30 mA Light source LED, visible red light 4) Angle of dispersion 2.5° Light spot diameter 120 x 120 mm at a distance of 3 m Switching outputs Q and Q– PNP NPN Signal voltage HIGH VS – 2.9 V VS Signal voltage LOW5) Approx. 0 V ≤ 2.9 V Output current IA max. ≤ 100 mA Response time 6) ≤ 625 μs Max. switching frequency 7) 800/s Connection technology Connection cable, 2 m Cable, 120 mm, with plug M 12, 4 pin Plug M 12, 4 pin Plug M 8, 4 pin VDE protection class M 12 8) VDE protection class M 8 8) III Protection type IP 67 Protection circuits 9) A, B, C Ambient temperature 10) Operation –40 ... +60 °C Storage –40 ... +75 °C Weight with connection cable 2 m/120 mm Approx. 80 g with equipment plug M 12/M 8, 4 pin Approx. 20 g WL 9-2 Scanning range Ordering information Technical data WL 9-2 P130 P430 N130 N430 P330 P630 1) Limit values 2) Must be within VS tolerances 3) Without load 4) Average service life at room temperature 100,000 h 5) At TU = +25 °C and 100 mA output current 6) With resistive load 7) With light/dark ratio 1:1 8) Withstand voltage 50 V 19) A = supply connections reverse polarity protected B = outputs short-circuit protected C = interference suppression 10) Do not distort cable below 0 °C Type WL 9-2P130 WL 9-2P430 WL 9-2N130 WL 9-2N430 WL 9-2P330 WL 9-2P630 Order no. 1 018 281 1 018 283 1 018 282 1 018 284 1 019 024 1 019 268 (m) 1 2 3 4 5 100 10 1 Function reserve 1 3 2 Operating range WL 9-2 Limiting scanning range 0(m) 1 2 3 4 5 1 2 3 Operating range Scanning range typ. max. 0 3.0 4.0 0 2.0 3.0 0 0.6/1.0 Reflective tape 0 ... 0.6 m Diamond Grade* 3 2 PL 40 A 0 ... 2 m 1 PL 80 A 0 ... 3 m Reflector type Operating range * 100 x 100 mm2 10 SENSICK WL 9-2 Photoelectric Reflex Switch, Standard, Teach-In Setting options Dimension illustration Red-light emitter LED as alignment aid Switching frequency 800/s Outputs short-circuit protected Teach-In Scanning range 0 ... 4 m 12 22 40 20 3 3 1.5 1 2 2 29.5 11 3 4 Middle of optic axis Mounting hole Ø 3.2 mm LED signal strength indicator Plug M 12 or M 8, 4 pin, 2 m connection cable or 120 mm cable with plug M 12, 4 pin Sensitivity control, teachable 1 2 3 4 5 Photoelectric reflex switch WL 9-2P131 WL 9-2P431 WL 9-2N131 WL 9-2N431 3 5 Cable receptacles Adapter plate Mounting bracket Reflectors Accessories Connection type L+ Q Q M brn wht blu blk 4 pin, M 12 WT 9-2P331 WT 9-2P631 WT 9-2P431 WT 9-2N431 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 8 1 L+ Q Q 4 2 3 M brn wht blu blk 4 x 0,14 mm2 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 12 with 120 mm cable WT 9-2P331 WT 9-2P131 WT 9-2N131 WT 9-2P631 SENSICK 11 Scanning range typ. max./on reflector 4 m/PL 80 A Supply voltage VS 1) DC 10 ... 30 V Ripple 2) ≤ 5 VPP Current consumption 3) ≤ 30 mA Light source LED, visible red light 4) Angle of dispersion 2.5° Light spot diameter 120 x 120 mm at a distance of 3 m Switching outputs Q and Q– PNP NPN Signal voltage HIGH VS – 2.9 V VS Signal voltage LOW5) Approx. 0 V ≤ 2.9 V Output current IA max. ≤ 100 mA Response time 6) ≤ 625 μs Max. switching frequency 7) 800/s Connection technology Connection cable, 2 m Cable, 120 mm, with plug M 12, 4 pin Plug M 12, 4 pin Plug M 8, 4 pin VDE protection class M 12 8) VDE protection class M 8 8) III Protection type IP 67 Protection circuits 9) A, B, C Ambient temperature 10) Operation –40 ... +60 °C Storage –40 ... +75 °C Weight with connection cable 2 m/120 mm Approx. 80 g with equipment plug M 12/M 8, 4 pin Approx. 20 g WL 9-2 Scanning range Ordering information Technical data WL 9-2 P131 P431 N131 N431 P331 P631 1) Limit values 2) Must be within VS tolerances 3) Without load 4) Average service life at room temperature 100,000 h 5) At TU = +25 °C and 100 mA output current 6) With resistive load 7) With light/dark ratio 1:1 8) Withstand voltage 50 V 19) A = supply connections reverse polarity protected B = outputs short-circuit protected C = interference suppreasion 10) Do not distort cable below 0 °C Type WL 9-2P131 WL 9-2P431 WL 9-2N131 WL 9-2N431 WL 9-2P331 WL 9-2P631 Order no. 1 018 285 1 018 287 1 018 286 1 018 288 1 019 025 1 019 269 Programming via Teach-In button. Simple programming: Position reflector in the beam and push the button: finished; LED confirms the Teach-In procedure. Teach-In values can be stored. Teach-In function Two operating modes: Default setting: short Teach-In time (< 8 s); for standard applications; approx. double reserve via switching threshold; LED lights continuously. Precise setting: long Teach-In time (> 8 s); for precise applications; small switching hysteresis; LED blinks. (m) 1 2 3 4 5 100 10 1 Function reserve 1 3 2 Operating range WL 9-2 Limiting scanning range 0(m) 1 2 3 4 5 1 2 3 Operating range Scanning range typ. max. 0 3.0 4.0 0 2.0 3.0 0 0.6/1.0 Reflective tape 0 ... 0.6 m Diamond Grade* 3 2 PL 40 A 0 ... 2 m 1 PL 80 A 0 ... 3 m Reflector type Operating range * 100 x 100 mm2 12 SENSICK WL 9-2 Photoelectric Reflex Switch, Focus 35 mm, Teach-In Setting options Dimension illustration LED light source, visible red light Sensitivity adjustment using the Teach-In method Switching frequency 800/s Outputs short-circuit protected Scanning range 0 ... 0.4 m 12 22 40 20 3 3 1.5 1 2 2 29.5 11 3 4 Middle of optic axis Mounting hole Ø 3.2 mm LED signal strength indicator Plug M 12 or M 8, 4 pin, 2 m connection cable or 120 mm cable with plug M 12, 4 pin Sensitivity control, teachable 1 2 3 4 5 Photoelectric reflex switch WL 9-2P121 WL 9-2P421 WL 9-2N121 WL 9-2N421 3 5 Cable receptacles Adapter plate Mounting bracket Reflectors Accessories Connection type L+ Q Q M brn wht blu blk 4 pin, M 12 WT 9-2P321 WT 9-2P621 WT 9-2P421 WT 9-2N421 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 8 1 L+ Q Q 4 2 3 M brn wht blu blk 4 x 0,14 mm2 1 L+ Q Q 4 2 3 M brn wht blu blk 4 pin, M 12 with 120 mm cable WT 9-2P321 WT 9-2P121 WT 9-2N121 WT 9-2P621 SENSICK 13 Scanning range typ. max./on reflector 0.4 m/PL 80 A Supply voltage VS 1) DC 10 ... 30 V Ripple 2) ≤ 5 VPP Current consumption 3) ≤ 30 mA Light source LED, visible red light 4) Light spot diameter 1.5 x 1.5 mm at a distance of 35 mm Switching outputs Q and Q– PNP NPN Signal voltage HIGH VS – 2.9 V VS Signal voltage LOW5) Approx. 0 V ≤ 2.9 V Output current IA max. ≤ 100 mA Response time 6) ≤ 625 μs Max. switching frequency 7) 800/s Connection technology Connection cable, 2 m Cable, 120 mm, with plug M 12, 4 pin Plug M 12, 4 pin Plug M 8, 4 pin VDE protection class M 12 8) VDE protection class M 8 8) III Protection type IP 67 Protection circuits 9) A, B, C Ambient temperature 10) Operation –40 ... +60 °C Storage –40 ... +75 °C Weight with connection cable 2 m/120 mm Approx. 80 g with equipment plug M 12/M 8, 4 pin Approx. 20 g WL 9-2 Scanning range Ordering information Technical data WL 9-2 P121 P421 N121 N421 P321 P621 1) Limit values 2) Must be within VS tolerances 3) Without load 4) Average service life at room temperature 100,000 h 5) At TU = +25 °C and 100 mA output current 6) With resistive load 7) With light/dark ratio 1:1 8) Withstand voltage 50 V 19) A = supply connections reverse polarity protected B = outputs short-circuit protected C = interference suppression 10) Do not distort cable below 0 °C Type WL 9-2P121 WL 9-2P421 WL 9-2N121 WL 9-2N421 WL 9-2P321 WL 9-2P621 Order no. 1 018 289 1 018 291 1 018 290 1 018 292 1 019 270 1 019 271 Programming via Teach-In button. Simple programming: Position reflector in the beam and push the button: finished; LED confirms the Teach-In procedure. Teach-In values can be stored. Teach-In function Two operating modes: Default setting: short Teach-In time (< 8 s); for standard applications; approx. double reserve via switching threshold; LED lights continuously. Precise setting: long Teach-In time (> 8 s); for precise applications; small switching hysteresis; LED blinks. (m) 0.1 0.2 0.3 0.4 0.5 0.6 100 10 1 Function reserve Limiting scanning range Operating range WL 9-2 2 1 3 0(m) 0.1 0.2 0.3 0.4 0.5 1 2 3 Operating range Limiting scanning range 0 0.3 0.4 0 0.2 0.3 0 0,.1 0.2 Reflective tape 0 ... 0.25 m Diamond Grade* 3 2 PL 40 A 0 ... 0.3 m 1 PL 80 A 0 ... 0.5 m Reflector type Operating range * 100 x 100 mm2 14 Accessoires SENSICK Dimension illustrations of reflectors Reflector 20 x 40 mm Order no. 1 012 719 Type PL 20 A Reflector 30 x 50 mm Order no. 1 002 314 Type PL 30 A 15 18 38 ø8 ø4.6 50 60 4.2 7.3 3.4 ø4.5 ø8 71 82 29.8 7.2 Reflector 40 x 60 mm Order no. 1 012 720 Type PL 40 A Reflector hexagonal, SW 48 mm Order no. 1 000 132 Type PL 50 A 34 38 7.8 40.2 52 56.6 59.8 ø8.5 ø4.5 8 78 68 59 Reflector 80 x 80 mm Order no. 1 003 865 Type PL 80 A Reflector ø 83 mm, center hole mounting Order no. 5 304 549 Type C 110 84 68 71 84 4.5 8 8.5 2.5 ø4.8 83 9 Also available as heatable model: Continuous heating: PL 50HK, Order no. 1 001 545 Regulated heating: PL 50HS, Order no. 1 009 871 Reflective tape fabricated sheet 749 x 914 mm Order no. 4 019 634 5 304 334 Type REF-DG-K REF-DG Dimension illustrations and ordering information 15 Accessoires SENSICK Contact assignments according to EN 50044 DC coding Dimension illustrations of cable receptacles Cable receptacles M 12, 4 pin, straight Order no. 6 007 302 Cable lengths – 5 ø18 M12x1 54 5 ø10.5 ø8.8 1.5 12 M12x1 14.5 27 25.5 42 Rmin 571) Rmin 571) 38.3 12 45° M12x1 26.5 14.5 12 1.5 ø8.8 ø10.5 Pin assignments Pin 1 = brown Pin 2 = white Pin 3 = blue Pin 4 = black 3 2 4 1 Pins 4 Type DOS-1204-G Cable receptacles M 12, 4 pin, angled Order no. 6 007 303 Cable lengths – 36 25 5 14.8 M12x1 ø18 36 5 20.5 Pins 4 Type DOS-1204-W Cable receptacles M 12, 4 pin, straight Pins 4 4 4 Type DOS-1204-G02M DOS-1204-G05M DOS-1204-G10M Order no. 6 009 382 6 009 866 6 010 543 Cable lengths 2 m 5 m 10 m Cable receptacles M 12, 4 pin, angled Pins 4 4 4 Type DOS-1204-W02M DOS-1204-W05M DOS-1204-W10M Order no. 6 009 383 6 009 867 6 010 541 Cable lengths 2 m 5 m 10 m Can be self-made for cables Ø 4.5 to 6.5 mm 1) Minimum bending radius with dynamic use Can be self-made for cables Ø 4.5 to 6.5 mm 1) Minimum bending radius with dynamic use Dimension illustrations and ordering information 16 Accessoires SENSICK Dimension illustrations and ordering information ø 11.6 M 8x1 38.4 Cable diameter max. 5.0 mm 28.0 ø 11.6 M 8x1 12.5 Cable diameter max. 5.0 mm SENSICK circular screwing system, M 8 plug, 4 pin, enclosure rating IP 67 M 8 cable receptacle, 4 pin, straight Type DOS-0804-G Order no. 6 009 974 M 8 cable receptacles, 4 pin, angled Type DOS-0804-W Order no. 6 009 975 2/wht 1/brn 4/blk 3/blu ø 10 30.5 Rmin1) M 8x1 3/blu 6 26 M 8x1 16.5 ø 10 Rmin1) 1/brn 4/blk 2/wht M 8 cable receptacle, 4 pin, straight M 8 cable receptacles, 4 pin, angled Cable diameter 5 mm, 4 x 0.25 mm2, PVC coating Cable diameter 5 mm, 4 x 0.25 mm2, PVC coating Cable length 2 m 5 m 10 m Type DOL-0804-G02M DOL-0804-G05M DOL-0804-G10M Order no. 6 009 870 6 009 872 6 010 754 Cable length 2 m 5 m 10 m Type DOL-0804-W02M DOL-0804-W05M DOL-0804-W10M Order no. 6 009 871 6 009 873 6 010 755 1) Minimum bending radius with dynamic use Rmin= 20x cable diameter SENSICK 17 Dimension illustration adapter plate Adapter plate Order no. 4 033 145 Type BEF-AP-W9 22 63.25 1 8.25 5 3.25 20 3 M 3 ø 3.2 Dimension illustration mounting bracket Mounting bracket Order no. 4 033 146 Type BEF-WN-W9-2 44 1 4 4 3.5 5 6 14.8 12 6.4 8 14.8 16 17 17 Accessoires Dimension illustrations and ordering information 8 008 988.0700 HJS • SM • Printed in Germany • We reserve the right to make changes Contact: Au s t r a l i a Phone +61 3 94 97 41 00 0 08 33 48 02 – toll free Fax +61 3 94 97 11 87 Au s t r i a Phone +43 2 23 66 22 88-0 Fax +43 2 23 66 22 88-5 Bel g i u m / Luxembourg Phone +32 24 66 55 66 Fax +32 24 63 35 07 Br a z i l Phone +55 11 55 61 26 83 Fax +55 11 5 35 41 53 C h i n a / Ho n g Kong Phone +8 52 27 63 69 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9 52 9 41-92 87 Representatives and agencies in all major industrial nations. SICK AG • Industrial Sensors • Sebastian-Kneipp-Straße 1 • D-79183 Waldkirch Phone +49/76 81/2 02-0 • Fax +49/76 81/2 02-36 09 • www.sick.de 1. Product profile 1.1 General description The BGA7124 MMIC is a one-stage amplifier, available in a low-cost leadless surface-mount package. It delivers 25 dBm output power at 1 dB gain compression and superior performance up to 2700 MHz. Its power saving features include easy quiescent current adjustment enabling class-AB operation and logic-level shutdown control to reduce the supply current to 4 μA. 1.2 Features and benefits 400 MHz to 2700 MHz frequency operating range 16 dB small signal gain at 2 GHz 25 dBm output power at 1 dB gain compression Integrated active biasing External matching allows broad application optimization of the electrical performance 3.3 V or 5 V single supply operation All pins ESD protected 1.3 Applications 1.4 Quick reference data [1] The supply current is adjustable; see Section 8.1 “Supply current adjustment”. [2] Operation outside this range is possible but not guaranteed. [3] PL = 11 dBm per tone; spacing = 1 MHz. BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier Rev. 3 — 9 September 2010 Product data sheet Wireless infrastructure (base station, repeater, backhaul systems) E-metering Broadband CPE/MoCA Satellite Master Antenna TV (SMATV) Industrial applications WLAN/ISM/RFID Table 1. Quick reference data Input and output impedances matched to 50 Ω, SHDN = HIGH (shutdown disabled). Typical values at VCC = 5 V; ICC = 130 mA; Tcase = 25 °C; unless otherwise specified. Symbol Parameter Conditions Min Typ Max Unit ICC supply current VCC = 5.0 V [1] 50 - 170 mA f frequency [2] 400 - 2700 MHz Gp power gain f = 2140 MHz 14.5 16 17.5 dB PL(1dB) output power at 1 dB gain compression f = 2140 MHz 23.5 24.5 - dBm IP3O output third-order intercept point f = 2140 MHz [3] 34.5 37.5 - dBm BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 2 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 2. Pinning information 2.1 Pinning 2.2 Pin description [1] This pin is DC-coupled and requires an external DC-blocking capacitor. [2] RF decoupled. [3] The center metal base of the SOT908-1 also functions as heatsink for the power amplifier. 3. Ordering information Fig 1. HVSON8 package pin configuration 014aab046 VCC(BIAS) VCC(RF) SHDN VCC(RF) RF_IN ICQ_ADJ GND PAD n.c. Transparent top view 4 5 3 6 2 7 1 8 terminal 1 index area BGA7124 n.c. Table 2. Pin description Symbol Pin Description n.c. 1, 4 not connected VCC(RF) 2, 3 RF output for the power amplifier and DC supply input for the RF transistor collector [1] VCC(BIAS) 5 bias supply voltage [2] SHDN 6 shutdown control function enabled/disabled RF_IN 7 RF input for the power amplifier [1] ICQ_ADJ 8 quiescent collector current adjustment controlled by an external resistor GND GND pad RF and DC ground[3] Table 3. Ordering information Type number Package Name Description Version BGA7124 HVSON8 plastic thermal enhanced very thin small outline package; no leads; 8 terminals; body 3 × 3 × 0.85 mm SOT908-1 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 3 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 4. Functional diagram 5. Shutdown control Fig 2. Functional diagram BANDGAP INPUT MATCH OUTPUT MATCH BIAS ENABLE V/I CONVERTER RF_OUT GND R1 R2 RF_IN SHDN VCC ICQ_ADJ 6 7 5 8 2, 3 014aab047 VCC(BIAS) VCC(RF) Table 4. Shutdown control settings Mode Mode description Function description Pin SHDN Vctrl(sd) (V) Ictrl(sd) (μA) Min Max Min Max Idle medium power MMIC fully off; minimal supply current shutdown control enabled 0 0 0.7 - 2 TX medium power MMIC transmit mode shutdown control disabled 1 2.5 VCC(BIAS)- 9 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 4 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 6. Limiting values [1] See Figure 3 for safe operating area. [2] The supply current is adjustable; see Section 8.1 “Supply current adjustment”. [3] If Vctrl(sd) exceeds VCC(BIAS), the internal ESD circuit can be damaged. To prevent this, it is recommended that the Ictrl(sd) is limited to 20 mA. If the SHDN function is not used, the SHDN pin should be connected to the VCC(BIAS) pin. Table 5. Limiting values In accordance with the Absolute Maximum Rating System (IEC 60134). Symbol Parameter Conditions Min Max Unit VCC(RF) RF supply voltage [1]- 6.0 V VCC(BIAS) bias supply voltage [1]- 6.0 V ICC supply current [1][2] 50 200 mA Vctrl(sd) shutdown control voltage [3] 0.0 VCC(BIAS) V Pi(RF) RF input power - 20 dBm Tcase case temperature −40 +85 °C Tj junction temperature - 150 °C VESD electrostatic discharge voltage Human Body Model (HBM); According JEDEC standard 22-A114E - 2000 V Charged Device Model (CDM); According JEDEC standard 22-C101B - 500 V Exceeding the safe operating area limits may cause serious damage to the product. The impact on ICC due to the spread of the external ICQ resistor (R2) should be taken into account. The product-spread on ICC should be taken into account (see Section 8 “Static characteristics”). Fig 3. BGA7124 DC safe operating area VCC(RF) (V) 2 3 4 5 6 7 014aab048 150 100 200 250 ICC (mA) 50 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 5 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 7. Thermal characteristics [1] defined as thermal resistance from junction to GND paddle. 8. Static characteristics [1] The supply current is adjustable; see Section 8.1 “Supply current adjustment”. [2] See Section 12 “Application information”. 8.1 Supply current adjustment The supply current can be adjusted by changing the value of external ICQ resistor (R2); (see Figure 4). Table 6. Thermal characteristics Symbol Parameter Conditions Typ Max Unit Rth(j-mb) thermal resistance from junction to mounting base Tcase = 85 °C; VCC = 5 V; ICC = 130 mA [1] 32 - K/W Table 7. Characteristics Input and output impedances matched to 50 Ω, pin SHDN = HIGH (shutdown disabled). Typical values at VCC = 3.3 V or VCC = 5 V; Tcase = 25°C; unless otherwise specified. Symbol Parameter Conditions Min Typ Max Unit ICC supply current VCC = 3.3 V [1] 50 - 200 mA R1 = 0 Ω; R2 = 1330 Ω [2] 115 130 145 mA R1 = 2.2 Ω; R2 = 1070 Ω [2] 135 160 185 mA VCC = 5.0 V [1] 50 - 170 mA R1 = 0 Ω; R2 = 1960 Ω [2] 110 130 150 mA R1 = 2.2 Ω; R2 = 1650 Ω [2] 125 150 175 mA during shutdown; pin SHDN = LOW (shutdown enabled) - 4 6 μA BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 6 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 9. Dynamic characteristics a. 5 V supply voltage. b. 3.3 V supply voltage Fig 4. Supply current as a function of the value of R2 VCC = 5 V; R1 = 0 R2 (kΩ) 1.6 2.0 2.4 2.8 3.2 3.6 4.0 4.4 014aab049 90 130 170 ICC (mA) 50 VCC = 3.3 V; R1 = 0 R2 (kΩ) 0.9 1.4 1.9 2.4 2.9 3.4 014aab050 110 140 80 170 200 ICC (mA) 50 Table 8. Characteristics at VCC = 5 V Input and output impedances matched to 50 Ω, pin SHDN = HIGH (shutdown disabled). Typical values at VCC = 5 V; ICC = 130 mA; Tcase = 25°C; see Section 12 “Application information”; unless otherwise specified. Symbol Parameter Conditions Min Typ Max Unit f frequency [1] 400 - 2700 MHz Gp power gain for small signals f = 940 MHz - 22.7 - dB f = 1960 MHz - 16.4 - dB f = 2140 MHz 14.5 16.0 17.5 dB f = 2445 MHz [2] - 14.2 - dB PL(1dB) output power at 1 dB gain compression f = 940 MHz - 25.0 - dBm f = 1960 MHz - 24.5 - dBm f = 2140 MHz 23.5 24.5 - dBm f = 2445 MHz [2] - 23.5 - dBm IP3O output third-order intercept point f = 940 MHz [3] - 38.5 - dBm f = 1960 MHz [3] - 38.0 - dBm f = 2140 MHz [3] 34.5 37.5 - dBm f = 2445 MHz [2][3] - 36.0 - dBm NF noise figure f = 940 MHz [4]- 5.2 - dB f = 1960 MHz [4]- 4.6 - dB f = 2140 MHz [4]- 4.8 6.5 dB f = 2445 MHz [2][4]- 5.4 - dB BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 7 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier [1] Operation outside this range is possible but not guaranteed. [2] ICC = 150 mA; see Section 12 “Application information”. [3] PL = 11 dBm per tone; spacing = 1 MHz. [4] Defined at Pi = −40 dBm; small signal conditions. RLin input return loss f = 940 MHz - −15 - dB f = 1960 MHz - −11 - dB f = 2140 MHz - −17 - dB f = 2445 MHz [2] - −13 - dB RLout output return loss f = 940 MHz - −8 - dB f = 1960 MHz - −12 - dB f = 2140 MHz - −15 - dB f = 2445 MHz [2] - −25 - dB Table 8. Characteristics at VCC = 5 V …continued Input and output impedances matched to 50 Ω, pin SHDN = HIGH (shutdown disabled). Typical values at VCC = 5 V; ICC = 130 mA; Tcase = 25°C; see Section 12 “Application information”; unless otherwise specified. Symbol Parameter Conditions Min Typ Max Unit BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 8 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier [1] Operation outside this range is possible but not guaranteed. [2] ICC = 160 mA; see Section 12 “Application information”. [3] PL= 11 dBm per tone; spacing = 1 MHz. [4] Defined at Pi = −40 dBm; small signal conditions. Table 9. Characteristics at VCC = 3.3 V Input and output impedances matched to 50 Ω, pin SHDN = HIGH (shutdown disabled). Typical values at VCC = 3.3 V; ICC = 130 mA; Tcase = 25°C, see Section 12 “Application information”; unless otherwise specified. Symbol Parameter Conditions Min Typ Max Unit f frequency [1] 400 - 2700 MHz Gp power gain for small signals f = 940 MHz - 22.5 - dB f = 2445 MHz [2]- 13.8 - dB PL(1dB) output power at 1 dB gain compression f = 940 MHz - 23.5 - dBm f = 2445 MHz [2]- 22.0 - dBm IP3O output third-order intercept point f = 940 MHz [3]- 36.4 - dBm f = 2445 MHz [2][3]- 35.2 - dBm NF noise figure f = 940 MHz [4]- 5.5 - dB f = 2445 MHz [2][4]- 5.5 - dB RLin input return loss f = 940 MHz - −15 - dB f = 2445 MHz [2] - −10 - dB RLout output return loss f = 940 MHz - −9 - dB f = 2445 MHz [2] - −25 - dB BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 9 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 9.1 Scattering parameters Table 10. Scattering parameters at 5 V, MMIC only VCC = 5 V; ICC = 130mA; Tcase = 25°C. f (MHz) s11 s21 s12 s22 Magnitude (ratio) Angle (degree) Magnitude (ratio) Angle (degree) Magnitude (ratio) Angle (degree) Magnitude (ratio) Angle (degree) 400 0.85 161.56 22.94 82.35 0.01 17.02 0.46 −156.50 500 0.90 159.44 11.82 82.58 0.01 27.08 0.63 176.13 600 0.90 152.15 9.98 73.86 0.01 24.10 0.64 169.61 700 0.89 145.75 8.59 66.00 0.01 21.41 0.64 164.34 800 0.88 139.33 7.55 58.86 0.02 18.47 0.65 159.29 900 0.87 133.19 6.74 51.66 0.02 14.00 0.65 154.44 1000 0.87 127.07 6.14 45.11 0.02 11.25 0.65 149.58 1100 0.87 120.67 5.61 38.20 0.02 7.99 0.65 144.25 1200 0.87 114.18 5.19 31.60 0.02 4.20 0.64 139.60 1300 0.86 107.68 4.82 25.08 0.02 0.31 0.64 134.85 1400 0.86 100.86 4.51 18.49 0.02 −4.01 0.63 130.13 1500 0.86 94.14 4.23 11.74 0.02 −8.65 0.63 125.02 1600 0.86 87.48 3.99 5.25 0.03 −13.15 0.63 120.13 1700 0.86 80.83 3.77 −1.50 0.03 −18.16 0.62 114.98 1800 0.86 74.14 3.56 −8.13 0.03 −23.28 0.62 109.78 1900 0.86 67.39 3.37 −14.94 0.03 −28.54 0.62 104.46 2000 0.86 60.70 3.19 −21.68 0.03 −33.68 0.63 99.01 2100 0.86 53.97 3.02 −28.68 0.03 −39.37 0.63 93.58 2200 0.86 47.78 2.85 −35.14 0.03 −44.84 0.63 88.17 2300 0.86 41.57 2.69 −41.70 0.03 −50.27 0.64 83.06 2400 0.86 35.43 2.54 −48.11 0.03 −55.62 0.64 78.10 2500 0.86 29.74 2.39 −54.19 0.04 −60.71 0.65 73.31 2600 0.86 24.79 2.27 −60.06 0.04 −65.48 0.65 68.64 2700 0.85 19.58 2.15 −66.14 0.04 −70.66 0.66 64.16 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 10 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 10. Reliability information 11. Moisture sensitivity Table 11. Scattering parameters at 3.3 V, MMIC only VCC = 3.3 V; ICC = 130mA; Tcase = 25°C. f (MHz) s11 s21 s12 s22 Magnitude (ratio) Angle (degree) Magnitude (ratio) Angle (degree) Magnitude (ratio) Angle (degree) Magnitude (ratio) Angle (degree) 400 0.84 161.94 21.25 73.81 0.01 17.66 0.57 −154.41 500 0.91 159.25 11.56 79.01 0.01 28.15 0.65 178.05 600 0.90 151.98 9.67 70.71 0.01 24.80 0.66 171.32 700 0.90 145.57 8.29 63.37 0.01 21.89 0.66 165.59 800 0.89 139.18 7.26 56.54 0.02 19.04 0.66 160.37 900 0.88 132.87 6.48 49.74 0.02 15.35 0.66 155.28 1000 0.88 126.78 5.90 43.30 0.02 11.89 0.66 150.23 1100 0.87 120.46 5.39 36.53 0.02 8.33 0.66 144.88 1200 0.87 113.94 4.97 30.05 0.02 4.50 0.65 140.03 1300 0.87 107.48 4.62 23.62 0.02 0.35 0.65 135.35 1400 0.87 100.69 4.32 17.15 0.02 −3.92 0.64 130.48 1500 0.86 93.93 4.05 10.48 0.02 −8.62 0.64 125.46 1600 0.86 87.28 3.81 4.05 0.03 −13.28 0.64 120.31 1700 0.86 80.71 3.61 −2.66 0.03 −18.26 0.64 115.13 1800 0.86 74.00 3.40 −9.21 0.03 −23.51 0.64 109.99 1900 0.86 67.27 3.22 −15.97 0.03 −28.87 0.63 104.66 2000 0.86 60.64 3.05 −22.71 0.03 −34.22 0.64 99.36 2100 0.86 53.84 2.89 −29.68 0.03 −39.95 0.64 93.93 2200 0.86 47.60 2.72 −36.12 0.03 −45.44 0.64 88.55 2300 0.86 41.43 2.57 −42.66 0.03 −51.06 0.65 83.38 2400 0.86 35.35 2.42 −49.01 0.04 −56.53 0.65 78.44 2500 0.85 29.64 2.28 −55.12 0.04 −61.72 0.66 73.56 2600 0.85 24.72 2.16 −60.91 0.04 −66.76 0.66 68.80 2700 0.85 19.59 2.04 −66.91 0.04 −71.84 0.67 64.30 Table 12. Reliability Life test Conditions Intrinsic failure rate HTOL According JESD85; confidence level 60 %; Tj = 55 °C; activation energy = 0.7 eV; acceleration factor determined according Arrhenius 4 Table 13. Moisture sensitivity level Test methodology Class JESD-22-A113 1 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 11 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 12. Application information 12.1 5 V applications 12.1.1 920 MHz to 960 MHz See Table 14 for a list of components. PCB board specification: Rogers RO4003C; Height = 0.508 mm; εr = 3.38; Copper thickness = 35 μm. Fig 5. 5 V/130 mA application schematic; 920 MHz to 960 MHz C3 C10 C4 C6 C8 C9 C7 R1 R2 ICQ_ADJ SHDN enable L1 L2 C2 MSL1 C1 MSL2 MSL3 MSL5 MSL6 MSL7 MSL8 RF_IN J1 J3 J2 RF_OUT BGA7124 50 Ω 50 Ω VCC C5 014aab051 V MSL4 CC(RF) VCC(BIAS) (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 6. Output power at 1 dB gain compression as a function of frequency Fig 7. Power gain as a function of frequency f (GHz) 0.92 0.93 0.94 0.95 0.96 014aab052 24 26 22 28 30 PL(1dB) (dBm) 20 (1) (2) (3) f (GHz) 0.92 0.93 0.94 0.95 0.96 014aab053 22 24 20 26 28 Gp (dB) 18 (1) (2) (3) BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 12 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier Tcase = 25 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 8. Input return loss, output return loss and isolation as a function of frequency Fig 9. Output third-order intercept point as a function of frequency RLout RLin ISL f (GHz) 0.92 0.93 0.94 0.95 0.96 014aab054 −20 −10 0 RLin, RLout, ISL (dB) −30 f (GHz) 0.92 0.93 0.94 0.95 0.96 014aab055 38 40 42 IP3O (dBm) 36 (1) (3) (2) See Table 14 for a list of components. Fig 10. 5 V/130 mA application reference board; 920 MHz to 960 MHz J3 GND VCC GND n.c. enable GND C9 C10 C8 C6 C4 C5 L2 C1 C3 R2 C2 L1 C7 R1 MSL6 MSL7 MSL4 MSL5 MSL1 MSL3 MSL8 MSL2 J1 J I HG F E D C B A 1 2 3 4 5 6 7 8 910 11 12 13 RF in J2 RF out 014aab056 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 13 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier [1] MSL1 to MSL8 dimensions specified as Width (W), Spacing (S) and Length (L). Table 14. 5 V/130 mA application list of components; 920 MHz to 960 MHz See Figure 5 and Figure 10 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks C1, C6 capacitor 68 pF DC blocking Murata GRM1885C1H680JA01D C2, C3 capacitor 3.3 pF input match Murata GRM1885C1H3R3CZ01D C4 capacitor 3.9 pF output match Murata GRM1885C1H3R9CZ01D C5 capacitor 1.0 pF output match Murata GRM1885C1H1R0CZ01D C7 capacitor 68 pF RF decoupling Murata GRM1885C1H680JA01D C8 capacitor 100 nF DC decoupling AVX 0603YC104KAT2A C9 capacitor 10 μF DC decoupling AVX 1206ZG106ZAT2A C10 capacitor 12 pF noise decoupling Murata GRM1555C1H120JZ01D J1, J2 RF connector SMA Emerson Network Power 142-0701-841 J3 DC connector 6-pins MOLEX L1 inductor 2.2 nH output match Tyco electronics 36501J2N2JTDG L2 inductor 22 nH DC feed Tyco electronics 36501J022JTDG MSL1[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm input match MSL2[1] micro stripline 1.14 mm × 0.8 mm × 2.95 mm input match MSL3[1] micro stripline 1.14 mm × 0.8 mm × 7.75 mm input match MSL4[1] micro stripline 1.14 mm × 0.8 mm × 23.4 mm output match MSL5[1] micro stripline 1.14 mm × 0.8 mm × 2.2 mm output match MSL6[1] micro stripline 1.14 mm × 0.8 mm × 3.15 mm output match MSL7[1] micro stripline 1.14 mm × 0.8 mm × 2.3 mm output match MSL8[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm output match R1 resistor 0 Ω Multicomp MC 0.063W 0603 0R R2 resistor (trimmer) 2 kΩ bias adjustment Bourns 3214W-1-202E BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 14 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 12.1.2 1930 MHz to 1990 MHz See Table 15 for a list of components. PCB board specification: Rogers RO4003C; Height = 0.508 mm; εr = 3.38; Copper thickness = 35 μm. Fig 11. 5 V/130 mA application schematic; 1930 MHz to 1990 MHz C3 C4 C6 C7 C5 R1 R2 ICQ_ADJ SHDN enable L1 C2 MSL1 C1 MSL2 MSL4 MSL5 MSL6 RF_IN RF_OUT BGA7124 50 Ω 50 Ω VCC 014aab057 MSL3 VCC(BIAS) VCC(RF) J1 J3 J2 (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 12. Output power at 1 dB gain compression as a function of frequency Fig 13. Power gain as a function of frequency 014aab058 f (GHz) 1.93 1.95 1.97 1.99 24 26 22 28 30 PL(1dB) (dBm) 20 (1) (2) (3) 014aab059 f (GHz) 1.93 1.95 1.97 1.99 14 16 12 18 20 Gp (dB) 10 (1) (2) (3) BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 15 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier Tcase = 25 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 14. Input return loss, output return loss and isolation as a function of frequency Fig 15. Output third-order intercept point as a function of frequency RLout RLin ISL f (GHz) 1.93 1.95 1.97 1.99 014aab060 −20 −10 0 RLin, RLout, ISL (dB) −30 f (GHz) 1.93 1.95 1.97 1.99 014aab061 36 38 40 34 IP3O (dBm) (2) (1) (3) See Table 15 for a list of components. Fig 16. 5 V/130 mA application reference board; 1930 MHz to 1990 MHz J3 GND VCC GND n.c. enable GND C7 C6 C4 C2 C3 C1 R2 L1 C5 R1 MSL6 MSL4 MSL5 MSL1 MSL2 MSL3 J1 J I HG F E D C B A 1 2 3 4 5 6 7 8 910 11 12 13 RF in J2 RF out 014aab062 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 16 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier [1] MSL1 to MSL6 dimensions specified as Width (W), Spacing (S) and Length (L). 12.1.3 2110 MHz to 2170 MHz Table 15. 5 V/130 mA application list of components; 1930 MHz to 1990 MHz See Figure 11 and Figure 16 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks C1, C4 capacitor 15 pF DC blocking Murata GRM1885C1H150JA01D C2 capacitor 2.2 pF input match Murata GRM1885C1H2R2CZ01D C3 capacitor 1.2 pF output match Murata GRM1885C1H1R2CZ01D C5 capacitor 15 pF RF decoupling Murata GRM1885C1H150JA01D C6 capacitor 100 nF DC decoupling AVX 0603YC104KAT2A C7 capacitor 10 μF DC decoupling AVX 1206ZG106ZAT2A J1, J2 RF connector SMA Emerson Network Power 142-0701-841 J3 DC connector 6-pins MOLEX L1 inductor 22 nH DC feed Tyco electronics 36501J022JTDG MSL1[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm input match MSL2[1] micro stripline 1.14 mm × 0.8 mm × 10.8 mm input match MSL3[1] micro stripline 1.14 mm × 0.8 mm × 5.8 mm output match MSL4[1] micro stripline 1.14 mm × 0.8 mm × 2.2 mm output match MSL5[1] micro stripline 1.14 mm × 0.8 mm × 3.7 mm output match MSL6[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm output match R1 resistor 0 Ω Multicomp MC 0.063W 0603 0R R2 resistor (trimmer) 2 kΩ bias adjustment Bourns 3214W-1-202E See Table 16 for a list of components. PCB board specification: Rogers RO4003C; Height = 0.508 mm; εr = 3.38; Copper thickness = 35 μm. Fig 17. 5 V/130 mA application schematic; 2110 MHz to 2170 MHz RF_OUT C3 C4 C6 C7 C5 R1 R2 ICQ_ADJ SHDN enable L1 C2 MSL1 C1 MSL2 MSL4 MSL5 MSL6 RF_IN BGA7124 50 Ω 50 Ω VCC 014aab063 MSL3 VCC(BIAS) VCC(RF) J1 J3 J2 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 17 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 18. Output power at 1 dB gain compression as a function of frequency Fig 19. Power gain as a function of frequency 014aab064 f (GHz) 2.11 2.13 2.15 2.17 24 26 22 28 30 PL(1dB) (dBm) 20 (1) (2) (3) 014aab065 f (GHz) 2.11 2.13 2.15 2.17 14 16 12 18 20 Gp (dB) 10 (1) (2) (3) Tcase = 25 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 20. Input return loss, output return loss and isolation as a function of frequency Fig 21. Output third-order intercept point as a function of frequency RLout RLin ISL f (GHz) 2.11 2.13 2.15 2.17 014aab066 −20 −10 0 RLin, RLout, ISL (dB) −30 (3) (2) (1) f (GHz) 2.11 2.13 2.15 2.17 014aab067 36 38 40 IP3O (dBm) 34 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 18 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier See Table 16 for a list of components. Fig 22. 5 V/130 mA application reference board; 2110 MHz to 2170 MHz J3 GND VCC GND n.c. enable GND C7 C6 C4 C2 C3 C1 R2 L1 C5 R1 MSL6 MSL4 MSL5 MSL1 MSL2 MSL3 J1 J I HG F E D C B A 1 2 3 4 5 6 7 8 910 11 12 13 RF in J2 RF out 014aab068 Table 16. 5 V/130 mA application list of components; 2110 MHz to 2170 MHz See Figure 17 and Figure 22 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks C1, C4 capacitor 15 pF DC blocking Murata GRM1885C1H150JA01D C2 capacitor 2.7 pF input match Murata GRM1885C1H2R7CZ01D C3 capacitor 1.5 pF output match Murata GRM1885C1H1R5CZ01D C5 capacitor 15 pF RF decoupling Murata GRM1885C1H150JA01D C6 capacitor 100 nF DC decoupling AVX 0603YC104KAT2A C7 capacitor 10 μF DC decoupling AVX 1206ZG106ZAT2A J1, J2 RF connector SMA Emerson Network Power 142-0701-841 J3 DC connector 6-pins MOLEX L1 inductor 22 nH DC feed Tyco electronics 36501J022JTDG MSL1[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm input match MSL2[1] micro stripline 1.14 mm × 0.8 mm × 10.8 mm input match MSL3[1] micro stripline 1.14 mm × 0.8 mm × 5.8 mm output match MSL4[1] micro stripline 1.14 mm × 0.8 mm × 2.5 mm output match MSL5[1] micro stripline 1.14 mm × 0.8 mm × 3.5 mm output match BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 19 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier [1] MSL1 to MSL6 dimensions specified as Width (W), Spacing (S) and Length (L). 12.1.4 2405 MHz to 2485 MHz MSL6[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm output match R1 resistor 0 Ω Multicomp MC 0.063W 0603 0R R2 resistor (trimmer) 2 kΩ bias adjustment Bourns 3214W-1-202E Table 16. 5 V/130 mA application list of components; 2110 MHz to 2170 MHz …continued See Figure 17 and Figure 22 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks See Table 17 for a list of components. PCB board specification: Rogers RO4003C; Height = 0.508 mm; εr = 3.38; Copper thickness = 35 μm. Fig 23. 5 V/130 mA application schematic; 2405 MHz to 2485 MHz C3 C4 C5 C7 C8 C6 R1 R2 ICQ_ADJ SHDN enable L1 C2 MSL1 C1 MSL2 MSL3 MSL4 MSL5 RF_IN RF_OUT BGA7124 50 Ω 50 Ω VCC 014aab069 VCC(BIAS) VCC(RF) J1 J3 J2 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 20 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 24. Output power at 1 dB gain compression as a function of frequency Fig 25. Power gain as a function of frequency (3) (2) (1) f (GHz) 2.405 2.425 2.445 2.465 2.485 014aab070 20 22 18 24 26 PL(1dB) (dBm) 16 (3) (2) (1) f (GHz) 2.405 2.425 2.445 2.465 2.485 014aab071 14 16 12 18 20 Gp (dB) 10 Tcase = 25 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 26. Input return loss, output return loss and isolation as a function of frequency Fig 27. Output third-order intercept point as a function of frequency RLout RLin ISL f (GHz) 2.405 2.425 2.445 2.465 2.485 014aab072 −20 −10 0 RLin, RLout, ISL (dB) −30 f (GHz) 2.405 2.425 2.445 2.465 2.485 014aab073 34 36 38 32 IP3O (dBm) (1) (2) (3) BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 21 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier See Table 17 for a list of components. Fig 28. 5 V/130 mA application reference board; 2405 MHz to 2485 MHz J3 GND VCC GND n.c. enable GND C8 C7 C5 C2 C3 C4 C1 R2 L1 C6 R1 MSL1 MSL2 MSL3 MSL4 MSL5 J1 J I HG F E D C B A 1 2 3 4 5 6 7 8 910 11 12 13 RF in J2 RF out 014aab074 Table 17. 5 V/130 mA application list of components; 2405 MHz to 2485 MHz See Figure 23 and Figure 28 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks C1, C5 capacitor 12 pF DC blocking Murata GRM1885C1H120JA01D C2 capacitor 2.2 pF input match Murata GRM1885C1H2R2CZ01D C3 capacitor 0.82 pF output match Murata GRM1885C1HR82CZ01D C4 capacitor 0.68 pF output match Murata GRM1885C1HR68CZ01D C6 capacitor 12 pF RF decoupling Murata GRM1885C1H120JA01D C7 capacitor 100 nF DC decoupling AVX 0603YC104KAT2A C8 capacitor 10 μF DC decoupling AVX 1206ZG106ZAT2A J1, J2 RF connector SMA Emerson Network Power 142-0701-841 J3 DC connector 6-pins MOLEX L1 inductor 22 nH DC feed Tyco electronics 36501J022JTDG MSL1[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm input match MSL2[1] micro stripline 1.14 mm × 0.8 mm × 10.8 mm input match MSL3[1] micro stripline 1.14 mm × 0.8 mm × 7.3 mm output match MSL4[1] micro stripline 1.14 mm × 0.8 mm × 4.3 mm output match BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 22 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier [1] MSL1 to MSL5 dimensions specified as Width (W), Spacing (S) and Length (L). 12.2 3.3 V applications 12.2.1 920 MHz to 960 MHz MSL5[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm output match R1 resistor 2.2 Ω Multicomp MC 0.063W 0603 2R2 R2 resistor (trimmer) 2 kΩ bias adjustment Bourns 3214W-1-202E Table 17. 5 V/130 mA application list of components; 2405 MHz to 2485 MHz …continued See Figure 23 and Figure 28 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks See Table 18 for a list of components. PCB board specification: Rogers RO4003C; Height = 0.508 mm; εr = 3.38; Copper thickness = 35 μm. Fig 29. 3.3 V/130 mA application schematic; 920 MHz to 960 MHz C3 C4 C6 C8 C9 C7 R1 R2 ICQ_ADJ SHDN enable L1 L2 C2 MSL1 C1 MSL2 MSL3 RF_IN MSL4 MSL5 MSL6 MSL7 MSL8 RF_OUT BGA7124 50 Ω 50 Ω VCC C5 014aab075 VCC(BIAS) VCC(RF) J1 J3 J2 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 23 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 30. Output power at 1 dB gain compression as a function of frequency Fig 31. Power gain as a function of frequency f (GHz) 0.92 0.93 0.94 0.95 0.96 014aab076 24 26 22 28 30 PL(1dB) (dBm) 20 (1) (2) (3) f (GHz) 0.92 0.93 0.94 0.95 0.96 014aab077 22 24 20 26 28 Gp (dB) 18 (1) (2) (3) Tcase = 25 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 32. Input return loss, output return loss and isolation as a function of frequency Fig 33. Output third-order intercept point as a function of frequency RLout RLin ISL f (GHz) 0.92 0.93 0.94 0.95 0.96 014aab078 −20 −10 0 RLin, RLout, ISL (dB) −30 f (GHz) 0.92 0.93 0.94 0.95 0.96 014aab079 36 38 40 IP3O (dBm) 34 (1) (3) (2) BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 24 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier See Table 18 for a list of components. Fig 34. 3.3 V/130 mA application reference board; 920 MHz to 960 MHz J3 GND VCC GND n.c. enable GND C9 C8 C6 C2 C3 C4 C5 C1 R2 L2 L1 C7 R1 MSL1 MSL3 MSL8 MSL2 MSL4 MSL6 MSL7 MSL5 J1 J I HG F E D C B A 1 2 3 4 5 6 7 8 910 11 12 13 RF in J2 RF out 014aab080 Table 18. 3.3 V/130 mA application list of components; 920 MHz to 960 MHz See Figure 29 and Figure 34 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks C1, C6 capacitor 68 pF DC blocking Murata GRM1885C1H680JA01D C2, C3 capacitor 3.3 pF input match Murata GRM1885C1H3R3CZ01D C4 capacitor 3.9 pF output match Murata GRM1885C1H3R9CZ01D C5 capacitor 1.0 pF output match Murata GRM1885C1H1R0CZ01D C7 capacitor 68 pF RF decoupling Murata GRM1885C1H680JA01D C8 capacitor 100 nF DC decoupling AVX 0603YC104KAT2A C9 capacitor 10 μF DC decoupling AVX 1206ZG106ZAT2A J1, J2 RF connector SMA Emerson Network Power 142-0701-841 J3 DC connector 6-pins MOLEX L1 inductor 2.2 nH output match Tyco electronics 36501J2N2JTDG L2 inductor 22 nH DC feed Tyco electronics 36501J022JTDG MSL1[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm input match MSL2[1] micro stripline 1.14 mm × 0.8 mm × 2.95 mm input match MSL3[1] micro stripline 1.14 mm × 0.8 mm × 7.75 mm input match MSL4[1] micro stripline 1.14 mm × 0.8 mm × 23.4 mm output match MSL5[1] micro stripline 1.14 mm × 0.8 mm × 2.2 mm output match BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 25 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier [1] MSL1 to MSL8 dimensions specified as Width (W), Spacing (S) and Length (L). 12.2.2 2405 MHz to 2485 MHz MSL6[1] micro stripline 1.14 mm × 0.8 mm × 2.4 mm output match MSL7[1] micro stripline 1.14 mm × 0.8 mm × 2.3 mm output match MSL8[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm output match R1 resistor 0 Ω Multicomp MC 0.063W 0603 0R R2 resistor (trimmer) 2 kΩ bias adjustment Bourns 3214W-1-202E Table 18. 3.3 V/130 mA application list of components; 920 MHz to 960 MHz …continued See Figure 29 and Figure 34 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks See Table 19 for a list of components. PCB board specification: Rogers RO4003C; Height = 0.508 mm; εr = 3.38; Copper thickness = 35 μm Fig 35. 3.3 V/130 mA application schematic; 2405 MHz to 2485 MHz RF_OUT C3 C5 C7 C8 C6 R1 R2 ICQ_ADJ SHDN enable L1 C2 MSL1 C1 MSL2 MSL3 MSL4 MSL5 RF_IN BGA7124 50 Ω 50 Ω VCC C4 014aab081 VCC(BIAS) VCC(RF) J1 J3 J2 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 26 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 36. Output power at 1 dB gain compression as a function of frequency Fig 37. Power gain as a function of frequency f (GHz) 2.405 2.425 2.445 2.465 2.485 014aab082 20 22 18 24 26 PL(1dB) (dBm) 16 (3) (1) (2) f (GHz) 2.405 2.425 2.445 2.465 2.485 014aab083 14 16 12 18 20 Gp (dB) 10 (1) (2) (3) Tcase = 25 °C. (1) Tcase = −40 °C. (2) Tcase = 25 °C. (3) Tcase = 85 °C. Fig 38. Input return loss, output return loss and isolation as a function of frequency Fig 39. Output third-order intercept point as a function of frequency RLout RLin ISL f (GHz) 2.405 2.425 2.445 2.465 2.485 014aab084 −20 −10 0 RLin, RLout, ISL (dB) −30 (2) (1) (3) f (GHz) 2.405 2.425 2.445 2.465 2.485 014aab085 34 36 38 IP3O (dBm) 32 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 27 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier See Table 19 for a list of components. Fig 40. 3.3 V/130 mA application reference board; 2405 MHz to 2485 MHz J3 GND VCC GND n.c. enable GND C8 C7 C5 C2 C3 C4 C1 R2 L1 C6 R1 MSL1 MSL2 MSL3 MSL4 MSL5 J1 J I HG F E D C B A 1 2 3 4 5 6 7 8 910 11 12 13 RF in J2 RF out 014aab086 Table 19. 3.3 V/130 mA application list of components; 2405 MHz to 2485 MHz See Figure 35 and Figure 40 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks C1, C5 capacitor 12 pF DC blocking Murata GRM1885C1H120JA01D C2 capacitor 2.2 pF input match Murata GRM1885C1H2R2CZ01D C3 capacitor 0.82 pF output match Murata GRM1885C1HR82CZ01D C4 capacitor 0.68 pF output match Murata GRM1885C1HR68CZ01D C6 capacitor 12 pF RF decoupling Murata GRM1885C1H120JA01D C7 capacitor 100 nF DC decoupling AVX 0603YC104KAT2A C8 capacitor 10 μF DC decoupling AVX 1206ZG106ZAT2A J1, J2 RF connector SMA Emerson Network Power 142-0701-841 J3 DC connector 6-pins MOLEX L1 inductor 22 nH DC feed Tyco electronics 36501J022JTDG MSL1[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm input match MSL2[1] micro stripline 1.14 mm × 0.8 mm × 10.8 mm input match MSL3[1] micro stripline 1.14 mm × 0.8 mm × 7.3 mm output match MSL4[1] micro stripline 1.14 mm × 0.8 mm × 4.3 mm output match BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 28 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier [1] MSL1 to MSL5 dimensions specified as Width (W), Spacing (S) and Length (L). 12.3 PCB stack MSL5[1] micro stripline 1.14 mm × 0.8 mm × 10.95 mm output match R1 resistor 2.2 Ω Multicomp MC 0.063W 0603 2R2 R2 resistor (trimmer) 2 kΩ bias adjustment Bourns 3214W-1-202E Table 19. 3.3 V/130 mA application list of components; 2405 MHz to 2485 MHz …continued See Figure 35 and Figure 40 for component layout. Printed-Circuit Board (PCB): Rogers RO4003C stack; height = 0.508 mm; copper plating thickness = 35 μm. Component Description Value Function Remarks (1) Pre-pregnated RO4003Cdielectric constant εr = 3.38 Fig 41. PCB stack through via RF and analog ground RF and analog routing analog routing RF and analog ground 35 μm (1 oz.) copper + 0.3 μm gold plating RO4003C, 0.51 mm (20 mil) 35 μm (1 oz.) copper (1) 0.2 mm (8 mil) FR4, 0.15 mm (6 mil) 35 μm (1 oz.) copper 35 μm (1 oz.) copper 014aab087 BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 29 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 13. Package outline Fig 42. Package outline SOT908-1 (HVSON8) 1 0.2 0.5 0.05 0.00 UNIT A1 b D(1) Eh e y 1.5 e1 OUTLINE REFERENCES VERSION EUROPEAN PROJECTION ISSUE DATE IEC JEDEC JEITA mm 3.1 2.9 c Dh 1.65 1.35 y1 3.1 2.9 2.25 1.95 0.3 0.2 0.05 0.1 DIMENSIONS (mm are the original dimensions) SOT908-1 MO-229 E(1) 0.5 0.3 L 0.1 v 0.05 w SOT908-1 HVSON8: plastic thermal enhanced very thin small outline package; no leads; 8 terminals; body 3 x 3 x 0.85 mm A(1) max. 05-09-26 05-10-05 Note 1. Plastic or metal protrusions of 0.075 mm maximum per side are not included. X terminal 1 index area D B A E detail X A A1 c C y1 C y exposed tie bar (4×) exposed tie bar (4×) b terminal 1 index area e1 e v M C A B w M C Eh Dh L 1 4 8 5 0 1 2 mm scale BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 30 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 14. Abbreviations 15. Revision history Table 20. Abbreviations Acronym Description CPE Customer-Premises Equipment DC Direct Current ESD ElectroStatic Discharge HTOL High Temperature Operating Life ISM Industrial, Scientific and Medical MMIC Monolithic Microwave Integrated Circuit MoCA Multimedia over Coax Alliance RFID Radio Frequency IDentification SMA SubMiniature version A TX Transmit WLAN Wireless Local Area Network Table 21. Revision history Document ID Release date Data sheet status Change notice Supersedes BGA7124 v.3 20100909 Product data sheet - BGA7124 v.2 Modifications: • Figure 5 on page 11: MSL symbols have been corrected. • Figure 11 on page 14: MSL symbols have been corrected. • Figure 17 on page 16: MSL symbols have been corrected. • Figure 23 on page 19: MSL symbols have been corrected. • Figure 29 on page 22: MSL symbols have been corrected. • Figure 35 on page 25: MSL symbols have been corrected. BGA7124 v.2 20100623 Product data sheet - BGA7124 v.1 BGA7124 v.1 20100421 Product data sheet - - BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 31 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier 16. Legal information 16.1 Data sheet status [1] Please consult the most recently issued document before initiating or completing a design. [2] The term ‘short data sheet’ is explained in section “Definitions”. [3] The product status of device(s) described in this document may have changed since this document was published and may differ in case of multiple devices. The latest product status information is available on the Internet at URL http://www.nxp.com. 16.2 Definitions Draft — The document is a draft version only. The content is still under internal review and subject to formal approval, which may result in modifications or additions. NXP Semiconductors does not give any representations or warranties as to the accuracy or completeness of information included herein and shall have no liability for the consequences of use of such information. Short data sheet — A short data sheet is an extract from a full data sheet with the same product type number(s) and title. A short data sheet is intended for quick reference only and should not be relied upon to contain detailed and full information. For detailed and full information see the relevant full data sheet, which is available on request via the local NXP Semiconductors sales office. In case of any inconsistency or conflict with the short data sheet, the full data sheet shall prevail. Product specification — The information and data provided in a Product data sheet shall define the specification of the product as agreed between NXP Semiconductors and its customer, unless NXP Semiconductors and customer have explicitly agreed otherwise in writing. In no event however, shall an agreement be valid in which the NXP Semiconductors product is deemed to offer functions and qualities beyond those described in the Product data sheet. 16.3 Disclaimers Limited warranty and liability — Information in this document is believed to be accurate and reliable. However, NXP Semiconductors does not give any representations or warranties, expressed or implied, as to the accuracy or completeness of such information and shall have no liability for the consequences of use of such information. In no event shall NXP Semiconductors be liable for any indirect, incidental, punitive, special or consequential damages (including - without limitation - lost profits, lost savings, business interruption, costs related to the removal or replacement of any products or rework charges) whether or not such damages are based on tort (including negligence), warranty, breach of contract or any other legal theory. Notwithstanding any damages that customer might incur for any reason whatsoever, NXP Semiconductors’ aggregate and cumulative liability towards customer for the products described herein shall be limited in accordance with the Terms and conditions of commercial sale of NXP Semiconductors. Right to make changes — NXP Semiconductors reserves the right to make changes to information published in this document, including without limitation specifications and product descriptions, at any time and without notice. This document supersedes and replaces all information supplied prior to the publication hereof. Suitability for use — NXP Semiconductors products are not designed, authorized or warranted to be suitable for use in life support, life-critical or safety-critical systems or equipment, nor in applications where failure or malfunction of an NXP Semiconductors product can reasonably be expected to result in personal injury, death or severe property or environmental damage. NXP Semiconductors accepts no liability for inclusion and/or use of NXP Semiconductors products in such equipment or applications and therefore such inclusion and/or use is at the customer’s own risk. Applications — Applications that are described herein for any of these products are for illustrative purposes only. NXP Semiconductors makes no representation or warranty that such applications will be suitable for the specified use without further testing or modification. Customers are responsible for the design and operation of their applications and products using NXP Semiconductors products, and NXP Semiconductors accepts no liability for any assistance with applications or customer product design. It is customer’s sole responsibility to determine whether the NXP Semiconductors product is suitable and fit for the customer’s applications and products planned, as well as for the planned application and use of customer’s third party customer(s). Customers should provide appropriate design and operating safeguards to minimize the risks associated with their applications and products. NXP Semiconductors does not accept any liability related to any default, damage, costs or problem which is based on any weakness or default in the customer’s applications or products, or the application or use by customer’s third party customer(s). Customer is responsible for doing all necessary testing for the customer’s applications and products using NXP Semiconductors products in order to avoid a default of the applications and the products or of the application or use by customer’s third party customer(s). NXP does not accept any liability in this respect. Limiting values — Stress above one or more limiting values (as defined in the Absolute Maximum Ratings System of IEC 60134) will cause permanent damage to the device. Limiting values are stress ratings only and (proper) operation of the device at these or any other conditions above those given in the Recommended operating conditions section (if present) or the Characteristics sections of this document is not warranted. Constant or repeated exposure to limiting values will permanently and irreversibly affect the quality and reliability of the device. Terms and conditions of commercial sale — NXP Semiconductors products are sold subject to the general terms and conditions of commercial sale, as published at http://www.nxp.com/profile/terms, unless otherwise agreed in a valid written individual agreement. In case an individual agreement is concluded only the terms and conditions of the respective agreement shall apply. NXP Semiconductors hereby expressly objects to applying the customer’s general terms and conditions with regard to the purchase of NXP Semiconductors products by customer. No offer to sell or license — Nothing in this document may be interpreted or construed as an offer to sell products that is open for acceptance or the grant, conveyance or implication of any license under any copyrights, patents or other industrial or intellectual property rights. Export control — This document as well as the item(s) described herein may be subject to export control regulations. Export might require a prior authorization from national authorities. Document status[1][2] Product status[3] Definition Objective [short] data sheet Development This document contains data from the objective specification for product development. Preliminary [short] data sheet Qualification This document contains data from the preliminary specification. Product [short] data sheet Production This document contains the product specification. BGA7124 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2010. All rights reserved. Product data sheet Rev. 3 — 9 September 2010 32 of 33 NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier Non-automotive qualified products — Unless this data sheet expressly states that this specific NXP Semiconductors product is automotive qualified, the product is not suitable for automotive use. It is neither qualified nor tested in accordance with automotive testing or application requirements. NXP Semiconductors accepts no liability for inclusion and/or use of non-automotive qualified products in automotive equipment or applications. In the event that customer uses the product for design-in and use in automotive applications to automotive specifications and standards, customer (a) shall use the product without NXP Semiconductors’ warranty of the product for such automotive applications, use and specifications, and (b) whenever customer uses the product for automotive applications beyond NXP Semiconductors’ specifications such use shall be solely at customer’s own risk, and (c) customer fully indemnifies NXP Semiconductors for any liability, damages or failed product claims resulting from customer design and use of the product for automotive applications beyond NXP Semiconductors’ standard warranty and NXP Semiconductors’ product specifications. Export control — This document as well as the item(s) described herein may be subject to export control regulations. Export might require a prior authorization from national authorities. 16.4 Trademarks Notice: All referenced brands, product names, service names and trademarks are the property of their respective owners. 17. Contact information For more information, please visit: http://www.nxp.com For sales office addresses, please send an email to: salesaddresses@nxp.com NXP Semiconductors BGA7124 400 MHz to 2700 MHz 0.25 W high linearity silicon amplifier © NXP B.V. 2010. All rights reserved. For more information, please visit: http://www.nxp.com For sales office addresses, please send an email to: salesaddresses@nxp.com Date of release: 9 September 2010 Document identifier: BGA7124 Please be aware that important notices concerning this document and the product(s) described herein, have been included in section ‘Legal information’. 18. Contents 1 Product profile . . . . . . . . . . . . . . . . . . . . . . . . . . 1 1.1 General description . . . . . . . . . . . . . . . . . . . . . 1 1.2 Features and benefits. . . . . . . . . . . . . . . . . . . . 1 1.3 Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 1.4 Quick reference data . . . . . . . . . . . . . . . . . . . . 1 2 Pinning information. . . . . . . . . . . . . . . . . . . . . . 2 2.1 Pinning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 2.2 Pin description . . . . . . . . . . . . . . . . . . . . . . . . . 2 3 Ordering information. . . . . . . . . . . . . . . . . . . . . 2 4 Functional diagram . . . . . . . . . . . . . . . . . . . . . . 3 5 Shutdown control . . . . . . . . . . . . . . . . . . . . . . . 3 6 Limiting values. . . . . . . . . . . . . . . . . . . . . . . . . . 4 7 Thermal characteristics . . . . . . . . . . . . . . . . . . 5 8 Static characteristics. . . . . . . . . . . . . . . . . . . . . 5 8.1 Supply current adjustment . . . . . . . . . . . . . . . . 5 9 Dynamic characteristics . . . . . . . . . . . . . . . . . . 6 9.1 Scattering parameters . . . . . . . . . . . . . . . . . . . 9 10 Reliability information . . . . . . . . . . . . . . . . . . . 10 11 Moisture sensitivity . . . . . . . . . . . . . . . . . . . . . 10 12 Application information. . . . . . . . . . . . . . . . . . 11 12.1 5 V applications . . . . . . . . . . . . . . . . . . . . . . . 11 12.1.1 920 MHz to 960 MHz . . . . . . . . . . . . . . . . . . . 11 12.1.2 1930 MHz to 1990 MHz . . . . . . . . . . . . . . . . . 14 12.1.3 2110 MHz to 2170 MHz . . . . . . . . . . . . . . . . . 16 12.1.4 2405 MHz to 2485 MHz . . . . . . . . . . . . . . . . . 19 12.2 3.3 V applications . . . . . . . . . . . . . . . . . . . . . . 22 12.2.1 920 MHz to 960 MHz . . . . . . . . . . . . . . . . . . . 22 12.2.2 2405 MHz to 2485 MHz . . . . . . . . . . . . . . . . . 25 12.3 PCB stack. . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 13 Package outline . . . . . . . . . . . . . . . . . . . . . . . . 29 14 Abbreviations. . . . . . . . . . . . . . . . . . . . . . . . . . 30 15 Revision history. . . . . . . . . . . . . . . . . . . . . . . . 30 16 Legal information. . . . . . . . . . . . . . . . . . . . . . . 31 16.1 Data sheet status . . . . . . . . . . . . . . . . . . . . . . 31 16.2 Definitions. . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 16.3 Disclaimers . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 16.4 Trademarks. . . . . . . . . . . . . . . . . . . . . . . . . . . 32 17 Contact information. . . . . . . . . . . . . . . . . . . . . 32 18 Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 REV. C Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. a 12-Bit Serial Daisy-Chain CMOS D/A Converter DAC8143 FUNCTIONAL BLOCK DIAGRAM INPUT 12-BIT SHIFT REGISTER DAC REGISTER 12-BIT D/A CONVERTER DAC8143 LOAD IN OUT CLK VDD RFB IOUT1 IOUT2 AGND SRO DGND SRI STB2 STB3 STB4 STB1 LD2 LD1 VREF CLR ADDRESS BUS ADDRESS DECODER STROBE LOAD SRI SRO DAC8143 STROBE LOAD SRI SRO DAC8143 STROBE LOAD SRI SRO DAC8143 STROBE LOAD SRI SRO DAC8143 WR DBX mP Figure 1. Multiple DAC8143s with Three-Wire Interface FEATURES Fast, Flexible, Microprocessor Interfacing in Serially Controlled Systems Buffered Digital Output Pin for Daisy-Chaining Multiple DACs Minimizes Address-Decoding in Multiple DAC Systems—Three-Wire Interface for Any Number of DACs One Data Line One CLK Line One Load Line Improved Resistance to ESD –408C to +858C for the Extended Industrial Temperature Range APPLICATIONS Multiple-Channel Data Acquisition Systems Process Control and Industrial Automation Test Equipment Remote Microprocessor-Controlled Systems GENERAL INFORMATION The DAC8143 is a 12-bit serial-input daisy-chain CMOS D/A converter that features serial data input and buffered serial data output. It was designed for multiple serial DAC systems, where serially daisy-chaining one DAC after another is greatly simplified. The DAC8143 also minimizes address decoding lines enabling simpler logic interfacing. It allows three-wire interface for any number of DACs: one data line, one CLK line and one load line. Serial data in the input register (MSB first) is sequentially clocked out to the SRO pin as the new data word (MSB first) is simultaneously clocked in from the SRI pin. The strobe inputs are used to clock in/out data on the rising or falling (user selected) strobe edges (STB1, STB2, STB3, STB4). When the shift register’s data has been updated, the new data word is transferred to the DAC register with use of LD1 and LD2 inputs. Separate LOAD control inputs allow simultaneous output updating of multiple DACs. An asynchronous CLEAR input resets the DAC register without altering data in the input register. Improved linearity and gain error performance permits reduced circuit parts count through the elimination of trimming components. Fast interface timing reduces timing design considerations while minimizing microprocessor wait states. The DAC8143 is available in plastic packages that are compatible with autoinsertion equipment. Plastic packaged devices come in the extended industrial temperature range of –40°C to +85°C. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 1999 ELECTRICAL CHARACTERISTICS Parameter Symbol Conditions Min Typ Max Units STATIC ACCURACY Resolution N 12 Bits Nonlinearity INL ±1 LSB Differential Nonlinearity1 DNL ±1 LSB Gain Error2 GFSE ±2 LSB Gain Tempco (DGain/DTemp)3 TCGFS ±5 ppm/°C Power Supply Rejection Ratio (DGain/DVDD) PSRR DVDD = ±5% ±0.0006 ±0.002 %/% Output Leakage Current4 ILKG TA = +25°C ±5 nA TA = Full Temperature Range ±25 nA Zero Scale Error5, 6 IZSE TA = +25°C ±0.002 ±0.03 LSB TA = Full Temperature Range ±0.01 ±0.15 LSB Input Resistance7 RIN VREF Pin 7 11 15 kW AC PERFORMANCE Output Current Settling Time3, 8 tS 0.380 1 ms AC Feedthrough Error (VREF to IOUT1)3, 9 FT VREF = 20 V p-p @ f = 10 kHz, TA = +25°C 2.0 mV p-p Digital-to-Analog Glitch Energy3, 10 Q VREF = 0 V, IOUT Load = 100 W, CEXT = 13 pF 20 nVs Total Harmonic Distortion3 THD VREF = 6 V rms @ 1 kHz DAC Register Loaded with All 1s –92 dB Output Noise Voltage Density3, 11 en 10 Hz to 100 kHz Between RFB and IOUT 13 nV/ÖHz DIGITAL INPUTS/OUTPUT Digital Input HIGH VIH 2.4 V Digital Input LOW VIL 0.8 V Input Leakage Current12 IIN VIN = 0 V to +5 V ±1 mA Input Capacitance CIN VIN = 0 V 8 pF Digital Output High VOH IOH = –200 mA 4 V Digital Output Low VOL IOL = 1.6 mA 0.4 V ANALOG OUTPUTS Output Capacitance3 COUT1 Digital Inputs = All 1s 90 pF COUT2 Digital Inputs = All 0s 90 pF Output Capacitance3 COUT1 Digital Inputs = All 0s 60 pF COUT2 Digital Inputs = All 1s 60 pF TIMING CHARACTERISTICS3 Serial Input to Strobe Setup Times tDS1 STB1 Used as the Strobe 50 ns (tSTB = 80 ns) tDS2 STB2 Used as the Strobe 20 ns tDS3 STB3 Used as the Strobe TA = +25°C 10 ns TA = Full Temperature Range 20 ns tDS4 STB4 Used as the Strobe 20 ns tDH1 STB1 Used as the Strobe TA = +25°C 40 ns TA = Full Temperature Range 50 ns tDH2 STB2 Used as the Strobe TA = +25°C 50 ns TA = Full Temperature Range 60 ns Serial Input to Strobe Hold Times (tSTB = 80 ns) tDH3 STB3 Used as the Strobe 80 ns tDH4 STB4 Used as the Strobe 80 ns –2– REV. C (@ VDD = +5 V; VREF = +10 V; VOUT1 = VOUT2 = VAGND = VDGND = 0 V; TA = Full Temperature Range specified under Absolute Maximum Ratings, unless otherwise noted.) DAC8143–SPECIFICATIONS ELECTRICAL CHARACTERISTICS DAC8143 Parameter Symbol Conditions Min Typ Max Units STB to SRO Propagation Delay13 tPD TA = +25°C 220 ns TA = Full Temperature Range 300 ns SRI Data Pulsewidth tSRI 100 ns STB1 Pulsewidth (STB1 = 80 ns)14 tSTB1 80 ns STB2 Pulsewidth (STB2 = 100 ns)14 tSTB2 80 ns STB3 Pulsewidth (STB3 = 80 ns)14 tSTB3 80 ns STB4 Pulsewidth (STB4 = 80 ns)14 tSTB4 80 ns Load Pulsewidth tLD1, tLD2 TA = +25°C 140 ns TA = Full Temperature Range 180 ns LSB Strobe into Input Register to Load DAC Register Time tASB 0 ns CLR Pulsewidth tCLR 80 ns POWER SUPPLY Supply Voltage VDD 4.75 5 5.25 V Supply Current IDD All Digital Inputs = VIH or VIL 2 mA All Digital Inputs = 0 V or VDD 0.1 mA Power Dissipation PD Digital Inputs = 0 V or VDD 0.5 mW 5 V ´ 0.1 mA Digital Inputs = VIH or VIL 10 mW 5 V ´ 2 mA NOTES 11All grades are monotonic to 12 bits over temperature. 12Using internal feedback resistor. 13Guaranteed by design and not tested. 14Applies to IOUT1; all digital inputs = VIL, VREF = +10 V; specification also applies for IOUT2 when all digital inputs = VIH. 15VREF = +10 V, all digital inputs = 0 V. 16Calculated from worst case RREF: IZSE (in LSBs) = (RREF ´ ILKG ´ 4096) /VREF. 17Absolute temperature coefficient is less than +300 ppm/°C. 18IOUT, Load = 100 W. CEXT = 13 pF, digital input = 0 V to VDD or VDD to 0 V. Extrapolated to 1/2 LSB: tS = propagation delay (tPD) +9 t, where t equals measured time constant of the final RC decay. 19All digital inputs = 0 V. 10VREF = 0 V, all digital inputs = 0 V to VDD or VDD to 0 V. 11Calculations from en = Ö4K TRB where: K = Boltzmann constant, J/KR = resistance W T = resistor temperature, K B = bandwidth, Hz 12Digital inputs are CMOS gates; IIN typically 1 nA at +25°C. 13Measured from active strobe edge (STB) to new data output at SRO; CL = 50 pF. 14Minimum low time pulsewidth for STB1, STB2, and STB4, and minimum high time pulsewidth for STB3. Specifications subject to change without notice. (@ VDD = +5 V; VREF = +10 V; VOUT1 = V0UT2 = VAGND = VDGND = 0 V; TA = Full Temperature Range specified under Absolute Maximum Ratings, unless otherwise noted.) DAC8143 REV. C –3– DAC8143 –4– REV. C PIN CONNECTIONS 16-Lead Epoxy Plastic DIP 16-Lead SOIC TOP VIEW (Not to Scale) 16 15 14 13 12 11 10 9 1 2 3 4 5 6 7 8 IOUT1 RFB DAC8143 IOUT2 VREF AGND VDD STB1 CLR LD1 DGND SRO STB4 SRI STB3 STB2 LD2 ABSOLUTE MAXIMUM RATINGS (TA = +25°C, unless otherwise noted.) VDD to DGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +17 V VREF to DGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±25 V VRFB to DGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±25 V AGND to DGND . . . . . . . . . . . . . . . . . . . . . . . . VDD + 0.3 V DGND to AGND . . . . . . . . . . . . . . . . . . . . . . . . VDD + 0.3 V Digital Input Voltage Range . . . . . . . . . . . . . . . –0.3 V to VDD Output Voltage (Pin 1, Pin 2) . . . . . . . . . . . . . . –0.3 V to VDD Operating Temperature Range FP/FS Versions . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . .+150°C Storage Temperature . . . . . . . . . . . . . . . . . . –65°C to +150°C Lead Temperature (Soldering, 60 sec) . . . . . . . . . . . .+300°C Package Type uJA* uJC Units 16-Lead Plastic DIP 76 33 °C/W 16-Lead SOIC 92 27 °C/W *qJA is specified for worst case mounting conditions, i.e., qJA is specified for device in socket for P-DIP package; qJA is specified for device soldered to printed circuit board for SOIC package. CAUTION 1. Do not apply voltage higher than VDD or less than DGND potential on any terminal except VREF (Pin 15) and RFB (Pin 16). 2. The digital control inputs are Zener-protected; however, permanent damage may occur on unprotected units from high energy electrostatic fields. Keep units in conductive foam at all times until ready to use. 3. Use proper antistatic handling procedures. 4. Absolute Maximum Ratings apply to packaged devices. Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. ORDERING GUIDE Gain Temperature Package Package Model Nonlinearity Error Range Descriptions Options DAC8143FP ±1 LSB ±2 LSB –40°C to +85°C 16-Lead Plastic DIP N-16 DAC8143FS ±1 LSB ±2 LSB –40°C to +85°C 16-Lead SOIC R-16W Die Size: 99 ´ 107 mil, 10,543 sq. mils. CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the DAC8143 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. WARNING! ESD SENSITIVE DEVICE DAC8143 REV. C –5– 10 FREQUENCY – Hz THD – dB –90 0.032 THD – % 0.010 –85 –80 –75 –70 –95 0.018 0.0056 0.0032 0.0018 100 1k 10k 100k VIN = 5V rms OUTPUT OP AMP: OP-42 Figure 3. Multiplying Mode Total Harmonic Distortion vs. Frequency Typical Performance Characteristics– ALL BITS ON 100 FREQUENCY – Hz B10 0 ATTENUATION – dB (MSB) B11 B9 B8 B7 B6 B5 B4 B3 B2 B1 (LSB) B0 DATA BITS "ON" (ALL OTHER DATA BITS "OFF") 1k 10k 100k 1M 10M 12 24 36 48 60 72 84 96 108 Figure 2. Multiplying Mode Frequency Response vs. Digital Code 3 0 VIN – Volts IDD – mA 2 1 0 1 2 3 4 5 Figure 4. Supply Current vs. Logic Input Voltage 4 1 VDD – Volts THRESHOLD VOLTAGE – Volts 3 2 0 1 2.4 –0.8 3 5 7 9 11 13 15 17 Figure 7. Logic Threshold Voltage vs. Supply Voltage 0.5 0 DIGITAL INPUT CODE – Decimal LINEARITY ERROR – LSB 0.4 0.3 0.2 0.1 0.0 –0.1 –0.2 –0.3 –0.4 –0.5 512 1024 1536 2048 2560 3072 3584 4095 Figure 5. Linearity Error vs. Digital Code 0.5 2 VREF – Volts DNL – LSB 4 6 8 10 0.25 0 –0.25 –0.5 Figure 8. DNL Error vs. Reference Voltage 0.5 2 VREF – Volts INL – LSB 4 6 8 10 0.25 0 –0.25 –0.5 Figure 6. Linearity Error vs. Reference Voltage 40 0 SRO – VOLTAGE OUT – Volts SINK 30 20 10 0 –10 –20 –30 –40 1 2 3 4 5 SOURCE OUTPUT CURRENT – mA TA = +258C LOGIC 1 LOGIC 0 Figure 9. Digital Output Voltage vs. Output Current DAC8143 –6– REV. C DEFINITION OF SPECIFICATIONS RESOLUTION The resolution of a DAC is the number of states (2n) into which the full-scale range (FSR) is divided (or resolved), where “n” is equal to the number of bits. SETTLING TIME Time required for the analog output of the DAC to settle to within 1/2 LSB of its final value for a given digital input stimulus; i.e., zero to full-scale. GAIN Ratio of the DAC’s external operational amplifier output voltage to the VREF input voltage when all digital inputs are HIGH. FEEDTHROUGH ERROR Error caused by capacitive coupling from VREF to output. Feedthrough error limits are specified with all switches off. OUTPUT CAPACITANCE Capacitance from IOUT1 to ground. OUTPUT LEAKAGE CURRENT Current appearing at IOUT1 when all digital inputs are LOW, or at IOUT2 terminal when all inputs are HIGH. GENERAL CIRCUIT INFORMATION The DAC8143 is a 12-bit serial-input, buffered serial-output, multiplying CMOS D/A converter. It has an R-2R resistor ladder network, a 12-bit input shift register, 12-bit DAC register, control logic circuitry, and a buffered digital output stage. The control logic forms an interface in which serial data is loaded, under microprocessor control, into the input shift register and then transferred, in parallel, to the DAC register. In addition, buffered serial output data is present at the SRO pin when input data is loaded into the input register. This buffered data follows the digital input data (SRI) by 12 clock cycles and is available for daisy-chaining additional DACs. An asynchronous CLEAR function allows resetting the DAC register to a zero code (0000 0000 0000) without altering data stored in the registers. A simplified circuit of the DAC8143 is shown in Figure 10. An inversed R-2R ladder network consisting of silicon-chrome, thin-film resistors, and twelve pairs of NMOS current-steering switches. These switches steer binarily weighted currents into either IOUT1 or IOUT2. Switching current to IOUT1 or IOUT2 yields a constant current in each ladder leg, regardless of digital input code. This constant current results in a constant input resistance at VREF equal to R (typically 11 kW). The VREF input may be driven by any reference voltage or current, ac or dc, that is within the limits stated in the Absolute Maximum Ratings chart. The twelve output current-steering switches are in series with the R-2R resistor ladder, and therefore, can introduce bit errors. It was essential to design these switches such that the switch “ON” resistance be binarily scaled so that the voltage drop across each switch remains constant. If, for example, Switch 1 of Figure 10 was designed with an “ON” resistance of 10 W, Switch 2 for 20 W, etc., a constant 5 mV drop would then be maintained across each switch. To further ensure accuracy across the full temperature range, permanently “ON” MOS switches were included in series with the feedback resistor and the R-2R ladder’s terminating resistor. The Simplified DAC Circuit, Figure 10, shows the location of these switches. These series switches are equivalently scaled to two times Switch 1 (MSB) and top Switch 12 (LSB) to maintain constant relative voltage drops with varying temperature. During any testing of the resistor ladder or RFEEDBACK (such as incoming inspection), VDD must be present to turn “ON” these series switches. VREF RFEEDBACK IOUT2 IOUT1 10kV 10kV 10kV 20kV 20kV 20kV 20kV 20kV S1 S2 S3 S12 10kV BIT 1 (MSB) BIT 2 BIT 3 BIT 12 (LSB) DIGITAL INPUTS (SWITCHES SHOWN FOR DIGITAL INPUTS "HIGH") * * *THESE SWITCHES PERMANENTLY "ON" Figure 10. Simplified DAC Circuit DAC8143 REV. C –7– ESD PROTECTION The DAC8143 digital inputs have been designed with ESD resistance incorporated through careful layout and the inclusion of input protection circuitry. Figure 11 shows the input protection diodes. High voltage static charges applied to the digital inputs are shunted to the supply and ground rails through forward biased diodes. These protection diodes were designed to clamp the inputs well below dangerous levels during static discharge conditions. VDD DTL/TTL/CMOS INPUTS Figure 11. Digital Input Protection EQUIVALENT CIRCUIT ANALYSIS Figures 12 and 13 show equivalent circuits for the DAC8143’s internal DAC with all bits LOW and HIGH, respectively. The reference current is switched to IOUT2 when all data bits are LOW, and to IOUT1 when all bits are HIGH. The ILEAKAGE current source is the combination of surface and junction leakages to the substrate. The 1/4096 current source represents the constant 1-bit current drain through the ladder’s terminating resistor. Output capacitance is dependent upon the digital input code. This is because the capacitance of a MOS transistor changes with applied gate voltage. This output capacitance varies between the low and high values. RFEEDBACK IOUT1 IOUT2 R = 10kV ILEAKAGE 60pF 1/4096 ILEAKAGE 90pF R = 10kV IREF VREF Figure 12. Equivalent Circuit (All Inputs LOW) IOUT2 ILEAKAGE 60pF RFEEDBACK IOUT1 R = 10kV 1/4096 ILEAKAGE 90pF R = 10kV IREF VREF Figure 13. Equivalent Circuit (All Inputs HIGH) DYNAMIC PERFORMANCE ANALOG OUTPUT IMPEDANCE The output resistance, as in the case of the output capacitance, varies with the digital input code. This resistance, looking back into the IOUT1 terminal, varies between 11 kW (the feedback resistor alone when all digital input are LOW) and 7.5 kW (the feedback resistor in parallel with approximately 30 kW of the R-2R ladder network resistance when any single bit logic is HIGH). Static accuracy and dynamic performance will be affected by these variations. The gain and phase stability of the output amplifier, board layout, and power supply decoupling will all affect the dynamic performance of the DAC8143. The use of a small compensation capacitor may be required when high speed operational amplifiers are used. It may be connected across the amplifier’s feedback resistor to provide the necessary phase compensation to critically damp the output. The considerations when using high speed amplifiers are: 1. Phase compensation (see Figures 16 and 17). 2. Power supply decoupling at the device socket and use of proper grounding techniques. OUTPUT AMPLIFIER CONSIDERATIONS When using high speed op amps, a small feedback capacitor (typically 5 pF–30 pF) should be used across the amplifiers to minimize overshoot and ringing. For low speed or static applications, ac specifications of the amplifier are not very critical. In high speed applications, slew rate, settling time, openloop gain and gain/phase margin specifications of the amplifier should be selected for the desired performance. It has already been noted that an offset can be caused by including the usual bias current compensation resistor in the amplifier’s noninverting input terminal. This resistor should not be used. Instead, the amplifier should have a bias current that is low over the temperature range of interest. Static accuracy is affected by the variation in the DAC’s output resistance. This variation is best illustrated by using the circuit of Figure 14 and the equation: VERROR = VOS 1+RFB RO æ è ç ö ø ÷ VOS VREF R R R ETC RFB R2 R2 R2 OP-77 Figure 14. Simplified Circuit DAC8143 –8– REV. C Where RO is a function of the digital code, and: RO = 10 kW for more than four bits of Logic 1, RO = 30 kW for any single bit of Logic 1. Therefore, the offset gain varies as follows: at code 0011 1111 1111, VERROR1 = VOS 1+10 kW 10 kW æ è ç ö ø ÷ = 2 VOS at code 0100 0000 0000, VERROR2 = VOS 1+10 kW 30 kW æ è ç ö ø ÷ = 4/3 VOS The error difference is 2/3 VOS. Since one LSB has a weight (for VREF = +10 V) of 2.4 mV for the DAC8143, it is clearly important that VOS be minimized, using either the amplifier’s pulling pins, an external pulling network, or by selection of an amplifier with inherently low VOS. Amplifiers with sufficiently low VOS include OP77, OP97, OP07, OP27, and OP42. INTERFACE LOGIC OPERATION The microprocessor interface of the DAC8143 has been designed with multiple STROBE and LOAD inputs to maximize interfacing options. Control signals decoding may be done on chip or with the use of external decoding circuitry (see Figure 21). Serial data is clocked into the input register and buffered output stage with STB1, STB2, or STB4. The strobe inputs are active on the rising edge. STB3 may be used with a falling edge clock data. WORD N –1 WORD N WORD N –2 WORD N –1 WORD N BIT 2 BIT 11 BIT 12 LSB BIT 1 MSB BIT 12 LSB BIT 1 BIT 2 SRI MSB BIT 1 BIT 2 MSB BIT 1 MSB BIT 2 BIT 12 LSB BIT 1 LSB tDS1, tDS2, tDS3, tDS4 SRO tDH1, tDH2, tDH3, tDH4 tPD tSTB1 tSTB2 tSTB3 tSTB4 tSTB1 tSTB2 tSTB3 tSTB4 * STROBE (STB1, STB2, STB4) 1 2 12 1 2 tLD1 tLD2 tSR1 11 12 tASB LD1 AND LD2 LOAD NEW 12-BIT WORD INTO INPUT REGISTER AND SHIFT OUT PREVIOUS WORD LOAD INPUT REGISTER'S DATA INTO DAC REGISTER NOTES: * STROBE WAVEFORM IS INVERTED IF STB3 IS USED TO STROBE SERIAL DATA BITS INTO INPUT REGISTER. ** DATA IS STROBED INTO AND OUT OF THE INPUT SHIFT REGISTER MSB FIRST. Figure 15. Timing Diagram Serial data output (SRO) follows the serial data input (SRI) by 12 clocked bits. Holding any STROBE input at its selected state (i.e., STB1, STB2 or STB4 at logic HIGH or STB3 at logic LOW) will act to prevent any further data input. When a new data word has been entered into the input register, it is transferred to the DAC register by asserting both LOAD inputs. The CLR input allows asynchronous resetting of the DAC register to 0000 0000 0000. This reset does not affect data held in the input registers. While in unipolar mode, a CLEAR will result in the analog output going to 0 V. In bipolar mode, the output will go to –VREF. INTERFACE INPUT DESCRIPTION STB1 (Pin 4), STB2 (Pin 8), STB4 (Pin 11)—Input Register and Buffered Output Strobe. Inputs Active on Rising Edge. Selected to load serial data into input register and buffered output stage. See Table I for details. STB3 (Pin 10)—Input Register and Buffered Output Strobe Input. Active on Falling Edge. Selected to load serial data into input register and buffered output stage. See Table I for details. LD1 (Pin 5), LD2 (Pin 9)—Load DAC Register Inputs. Active Low. Selected together to load contents of input register into DAC register. CLR (Pin 13)—Clear Input. Active Low. Asynchronous. When LOW, 12-bit DAC register is forced to a zero code (0000 0000 0000) regardless of other interface inputs. DAC8143 REV. C –9– Table I. Truth Table DAC8143 Logic Inputs Input Register/ Digital Output Control Inputs DAC Register Control Inputs STB4 STB3 STB2 STB1 CLR LD2 LD1 DAC8143 Operation Notes 0 1 0 g X X X 0 1 g 0 X X X Serial Data Bit Loaded from SRI 0 f 0 0 X X X into Input Register and Digital Output 2, 3 g 1 0 0 X X X (SRO Pin) after 12 Clocked Bits. 1 X X X X 0 X X No Operation (Input Register and SRO) 3 X X 1 X X X X 1 Reset DAC Register to Zero Code 0 X X (Code: 0000 0000 0000) 1, 3 (Asynchronous Operation) 1 1 X No Operation (DAC Register and SRO) 3 1 X 1 1 0 0 Load DAC Register with the Contents 3 of Input Register NOTES 1CLR = 0 asynchronously resets DAC Register to 0000 0000 0000, but has no effect on Input Register. 2Serial data is loaded into Input Register MSB first, on edges shown. g is positive edge, f is negative edge. 30 = Logic LOW, 1 = Logic HIGH, X = Don’t Care. APPLICATIONS INFORMATION UNIPOLAR OPERATION (2-QUADRANT) The circuit shown in Figures 16 and 17 may be used with an ac or dc reference voltage. The circuit’s output will range between 0 V and +10(4095/4096) V depending upon the digital input code. The relationship between the digital input and the analog output is shown in Table II. The VREF voltage range is the maximum input voltage range of the op amp or ±25 V, whichever is lowest. Table II. Unipolar Code Table Digital Input Nominal Analog Output (VOUT as Shown MSB LSB in Figures 16 and 17) 1 1 1 1 1 1 1 1 1 1 1 1 –VREF 4095 4096 æ è ç ö ø ÷ 1 0 0 0 0 0 0 0 0 0 0 1 –VREF 2049 4096 æèöø 1 0 0 0 0 0 0 0 0 0 0 0 –VREF 2048 4096 æè öø = – VREF 2 0 1 1 1 1 1 1 1 1 1 1 1 –VREF 2047 4096 æè öø 0 0 0 0 0 0 0 0 0 0 0 1 –VREF 1 4096 æè öø 0 0 0 0 0 0 0 0 0 0 0 0 –VREF 0 4096 æè öø = 0 NOTES 1Nominal full scale for the circuits of Figures 16 and 17 is given by FS = –VREF 4095 4096 æ è ç ö ø ÷ . 2Nominal LSB magnitude for the circuits of Figures 16 and 17 is given by LSB = VREF 1 4096 æ è ç ö ø ÷ or VREF(2–n). OP-77 +5V VREF VDD RFEEDBACK IOUT1 IOUT2 AGND DGND SRO (BUFFERED DIGITAL DATA OUT) 15pF +15V –15V VOUT 7 6 4 3 2 15 14 13 4, 5 8–11 7 1 2 3 6 12 CONTROL DAC8143 INPUTS SRI (SERIAL DATA IN) VREF –10V CLR Figure 16. Unipolar Operation with High Accuracy Op Amp (2-Quadrant) OP-42 +5V VREF VDD RFEEDBACK IOUT1 IOUT2 AGND DGND SRO (BUFFERED DIGITAL DATA OUT) 15pF +15V –15V VOUT 7 6 4 3 2 15 14 13 4, 5 8–11 7 1 2 3 6 12 CONTROL DAC8143 INPUTS SRI (SERIAL DATA IN) VREF –10V R2 50V R1 100V CLR Figure 17. Unipolar Operation with Fast Op Amp and Gain Error Trimming (2-Quadrant) DAC8143 –10– REV. C In many applications, the DAC8143’s zero scale error and low gain error, permit the elimination of external trimming components without adverse effects on circuit performance. For applications requiring a tighter gain error than 0.024% at 25°C for the top grade part, or 0.048% for the lower grade part, the circuit in Figure 17 may be used. Gain error may be trimmed by adjusting R1. The DAC register must first be loaded with all 1s. R1 is then adjusted until VOUT = –VREF (4095/4096). In the case of an adjustable VREF, R1 and RFEEDBACK may be omitted, with VREF adjusted to yield the desired full-scale output. BIPOLAR OPERATION (4-QUADRANT) Figure 18 details a suggested circuit for bipolar, or offset binary, operation. Table III shows the digital input-to-analog output relationship. The circuit uses offset binary coding. Twos complement code can be converted to offset binary by software inversion of the MSB or by the addition of an external inverter to the MSB input. Resistor R3, R4 and R5 must be selected to match within 0.01% and must all be of the same (preferably metal foil) type to assure temperature coefficient match. Mismatching between R3 and R4 causes offset and full-scale error. Calibration is performed by loading the DAC register with 1000 0000 0000 and adjusting R1 until VOUT = 0 V. R1 and R2 may be omitted by adjusting the ratio of R3 to R4 to yield VOUT = 0 V. Full scale can be adjusted by loading the DAC register with 1111 1111 1111 and adjusting either the amplitude of VREF or the value of R5 until the desired VOUT is achieved. Table III. Bipolar (Offset Binary) Code Table Digital Input Nominal Analog Output MSB LSB (VOUT as Shown in Figure 18) 1 1 1 1 1 1 1 1 1 1 1 1 +VREF 2047 2048 æè öø 1 0 0 0 0 0 0 0 0 0 0 1 +VREF 1 2048 æè öø 1 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 –VREF 1 2048 æè öø 0 0 0 0 0 0 0 0 0 0 0 1 –VREF 2047 2048 æè öø 0 0 0 0 0 0 0 0 0 0 0 0 –VREF 2048 2048 æè öø NOTES 1Nominal full scale for the circuits of Figure 18 is given by FS = VREF 2047 2048 æè öø . 2Nominal LSB magnitude for the circuits of Figure 18 is given by LSB = VREF 1 2048 æè öø . DAISY-CHAINING DAC8143s Many applications use multiple serial input DACs that use numerous interconnecting lines for address decoding and data lines. In addition, they use some type of buffering to reduce loading on the bus. The DAC8143 is ideal for just such an application. It not only reduces the number of interconnecting lines, but also reduces bus loading. The DAC8143 can be daisychained with only three lines: one data line, one CLK line and one load line, see Figure 19. VOUT 1/2 OP200 +5V R2 50V 12 15 7 R1 100V SERIAL DATA INPUT VIN 14 15 1 2 3 8-11 4, 5 13 6 DGND VREF SRI CONTROL BITS SRO CONTROL INPUTS FROM SYSTEM RESET BUFFERED SERIAL DATA OUT VDD RFB AGND IOUT2 IOUT1 DAC8143 C1 10-33pF COMMON GROUND R3 A1 10kV R4 20kV R5 20kV 1/2 OP200 A2 CLR Figure 18. Bipolar Operation (4-Quadrant, Offset Binary) DAC8143 REV. C –11– ANALOG/DIGITAL DIVISION The transfer function for the DAC8143 connect in the multiplying mode as shown in Figures 16 and 17 is: VO = –VIN A1 21 + A2 22 + A3 23 + ... A12 212 æ è ç ö ø ÷ where AX assumes a value of 1 for an “ON” bit and 0 for an “OFF” bit. The transfer function is modified when the DAC is connected in the feedback of an operational amplifier as shown in Figure 20 and is: VO = –VIN A1 21 + A2 22 + A3 23 + ... A12 212 æ è ççç ö ø ÷÷÷ The above transfer function is the division of an analog voltage (VREF) by a digital word. The amplifier goes to the rails with all bits “OFF” since division by zero is infinity. With all bits “ON” the gain is 1 (±1 LSB). The gain becomes 4096 with the LSB, Bit 12, “ON”. BUFFERED DIGITAL DATA OUT +5V SRO VREF RFB VDD IOUT1 DAC8143 AGND 3 2 12 DGND 15 6 16 14 1 3 2 6 VIN VOUT 4 13 DIGITAL INPUTS OP-42 + – Figure 20. Analog/Digital Divider APPLICATION TIPS In most applications, linearity depends on the potential of IOUT1, IOUT2, and AGND (Pins 1, 2 and 3) being exactly equal to each other. In most applications, the DAC is connected to an external op amp with its noninverting input tied to ground (see Figures 16 and 17). The amplifier selected should have a low input bias current and low drift over temperature. The amplifier’s input offset voltage should be nulled to less than ±200 mV (less than 10% of 1 LSB). The operational amplifier’s noninverting input should have a minimum resistance connection to ground; the usual bias current compensation resistor should not be used. This resistor can cause a variable offset voltage appearing as a varying output error. All grounded pins should tie to a single common ground point, avoiding ground loops. The VDD power supply should have a low noise level with no transients greater than +17 V. It is recommended that the digital inputs be taken to ground or VDD via a high value (1 MW) resistor; this will prevent the accumulation of static charge if the PC card is disconnected from the system. Peak supply current flows as the digital input pass through the transition region (see Figure 4). The supply current decreases as the input voltage approaches the supply rails (VDD or DGND), i.e., rapidly slewing logic signals that settle very near the supply rails will minimize supply current. INTERFACING TO THE MC6800 As shown in Figure 21, the DAC8143 may be interfaced to the 6800 by successively executing memory WRITE instruction while manipulating the data between WRITEs, so that each WRITE presents the next bit. In this example, the most significant bits are found in memory locations 0000 and 0001. The four MSBs are found in the lower half of 0000, the eight LSBs in 0001. The data is taken from the DB7 line. The serial data loading is triggered by STB4 which is asserted by a decoded memory WRITE to a memory location, R/W, and F2. A WRITE to another address location transfers data from input register to DAC register. STB1 DAC8143* SRI SRO LD2 LD1 STB3 STB2 STB4 CLR 74LS138 ADDRESS DECODER E1 A0 A2 E3 E2 A0 A15 R/W DB0 DB7 MC6800 16-BIT ADDRESS BUS 8-BIT DATA BUS +5V FROM SYSTEM RESET *ANALOG CIRCUITRY OMITTED FOR SIMPLICITY f2 Figure 21. DAC8143—MC6800 Interface ADDRESS DECODER STROBE LOAD DAC8143 SRI SRO ADDRESS BUS STROBE LOAD DAC8143 SRI SRO STROBE LOAD DAC8143 SRI SRO STROBE LOAD DAC8143 SRI SRO DBX mP WR Figure 19. Multiple DAC8143s with Three-Wire Interface DAC8143 –12– REV. C DAC8143 INTERFACE TO THE 8085 The DAC8143’s interface to the 8085 microprocessor is shown in Figure 22. Note that the microprocessor’s SOD line is used to present data serially to the DAC. Data is strobed into the DAC8143 by executing memory write instructions. The strobe 2 input is generated by decoding an address location and WR. Data is loaded into the DAC register with a memory write instruction to another address location. Serial data supplied to the DAC8143 must be present in the right-justified format in registers H and L of the microprocessor. STB1 DAC8143* SRI SRO LD2 LD1 STB3 STB2 STB4 CLR 74LS138 ADDRESS DECODER E1 A0 A2 E3 E2 WR ALE SOD 8085 ADDRESS BUS (16) DATA +5V FROM SYSTEM RESET *ANALOG CIRCUITRY OMITTED FOR SIMPLICITY +5V A0–A15 8212 (8) (8)AD0–7 Figure 22. DAC8143—8085 Interface DAC8143 INTERFACE TO THE 68000 Figure 23 shows the DAC8143 configured to the 68000 microprocessor. Serial data input is similar to that of the 6800 in Figure 21. STB1 DAC8143 SRI LD2 LD1 STB3 STB4 CLR ADDRESS DECODER A1 A23 AS DB15 DB0 68000mP ADDRESS BUS DATA BUS FROM SYSTEM RESET CS VMA VPA UDS +5V 1/4 74HC125 + STB2 Figure 23. DAC8143 to 68000 mP Interface OUTLINE DIMENSIONS Dimensions are shown in inches and (mm). 16-Lead Plastic DIP (N-16) 16 1 8 9 PIN 1 0.840 (21.34) 0.745 (18.92) 0.280 (7.11) 0.240 (6.10) SEATING PLANE 0.060 (1.52) 0.015 (0.38) 0.210 (5.33) MAX 0.022 (0.558) 0.014 (0.356) 0.160 (4.06) 0.115 (2.93) 0.100 (2.54) BSC 0.070 (1.77) 0.045 (1.15) 0.130 (3.30) MIN 0.195 (4.95) 0.115 (2.93) 0.015 (0.381) 0.008 (0.204) 0.325 (8.25) 0.300 (7.62) 16-Lead SOIC (R-16W) SEATING PLANE 0.0118 (0.30) 0.0040 (0.10) 0.0192 (0.49) 0.0138 (0.35) 0.1043 (2.65) 0.0926 (2.35) 0.050 (1.27) BSC 16 9 1 8 0.4193 (10.65) 0.3937 (10.00) 0.2992 (7.60) 0.2914 (7.40) PIN 1 0.4133 (10.50) 0.3977 (10.00) 0.0125 (0.32) 0.0091 (0.23) 88 08 0.0291 (0.74) 0.0098 (0.25)3 458 0.0500 (1.27) 0.0157 (0.40) PRINTED IN U.S.A. C3114c–2–3/99 Low Voltage, Low Power, Factory-Calibrated 16-/24-Bit Dual - ADC FEATURES HIGH RESOLUTION - ADCs 2 Independent ADCs (16- and 24-Bit Resolution) Factory-Calibrated (Field Calibration Not Required) Output Settles in 1 Conversion Cycle (Single Conversion Mode) Programmable Gain Front End Simultaneous Sampling and Conversion of 2 Signal Sources Separate Reference Inputs for Each Channel Simultaneous 50 Hz and 60 Hz Rejection at 20 Hz Update Rate ISOURCE Select™ 24-Bit No Missing Codes—Main ADC 13-Bit p-p Resolution @ 20 Hz, 20 mV Range 18-Bit p-p Resolution @ 20 Hz, 2.56 V Range INTERFACE 3-Wire Serial SPI®, QSPI™, MICROWIRE™, and DSP Compatible Schmitt Trigger on SCLK POWER Specified for Single 3 V and 5 V Operation Normal: 1.5 mA Typ @ 3 V Power-Down: 10 A (32 kHz Crystal Running) ON-CHIP FUNCTIONS Rail-Rail Input Buffer and PGA 4-Bit Digital I/O Port On-Chip Temperature Sensor Dual Switchable Excitation Current Sources Low-Side Power Switches Reference Detect Circuit APPLICATIONS Sensor Measurement Temperature Measurement Pressure Measurement Weigh Scales Portable Instrumentation 4 to 20 mA Transmitters GENERAL DESCRIPTION The AD7719 is a complete analog front end for low frequency measurement applications. It contains two high resolution Σ-Δ ADCs, switchable matched excitation current sources, low-side power switches, digital I/O port, and temperature sensor. The 24-bit main channel with PGA accepts fully differential, unipolar, and bipolar input signal ranges from 1.024 × REFIN1/128 to 1.024 × REFIN1. Signals can be converted directly from a transducer without the need for signal conditioning. The 16-bit auxiliary channel has an input signal range of REFIN2 or REFIN2/2. The device operates from a 32 kHz crystal with an on-chip PLL generating the required internal operating frequency. The output data rate from the part is software programmable. The peak-to-peak resolution from the part varies with the programmed gain and output data rate. The part operates from a single 3 V or 5 V supply. When operating from 3 V supplies, the power dissipation for the part is 4.5 mW with both ADCs enabled and 2.85 mW with only the main ADC enabled in unbuffered mode. The AD7719 is housed in 28-lead SOIC and TSSOP packages. FUNCTIONAL BLOCK DIAGRAM MUX AVDD IEXC1 200A IEXC2 200A AVDD AGND MUX1 BUF PGA TEMP SENSOR I/O PORT AVDD MAIN CHANNEL 24-BIT - ADC AUXILIARY CHANNEL 16-BIT - ADC REFERENCE DETECT OSC. AND PLL SERIAL INTERFACE AND CONTROL LOGIC DVDD DGND IOUT1 IOUT2 AIN1 AIN2 AIN3 AIN4 AIN5 AIN6 AVDD AGND REFIN2 PWRGND P1/SW1 P2/SW2 P3 P4 DOUT DIN SCLK CS RDY RESET REFIN1(+) REFIN1(–) XTAL1 XTAL2 AD7719 MUX2 REV. A AD7719 –2– FEATURES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . 1 SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . 6 ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 TIMING CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . 7 DIGITAL INTERFACE . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 PIN CONFIGURATION . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 PIN FUNCTION DESCRIPTIONS . . . . . . . . . . . . . . . . . . 9 TYPICAL PERFORMANCE CHARACTERISTICS . . . . 11 DUAL-CHANNEL ADC CIRCUIT INFORMATION . . . 12 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 Main Channel . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 Auxiliary Channel . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 Both Channels . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 MAIN AND AUXILIARY ADC NOISE PERFORMANCE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 ON-CHIP REGISTERS . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 Communications Register (A3, A2, A1, A0 = 0, 0, 0, 0) . . . . . . . . . . . . . . . . . . . . 19 Status Register (A3, A2, A1, A0 = 0, 0, 0, 0; Power-On Reset = 0x00) . . . . . . . . . . . . . . . . . . . . . . . 20 Mode Register (A3, A2, A1, A0 = 0, 0, 0, 1; Power-On Reset = 0x00) . . . . . . . . . . . . . . . . . . . . . . . 21 Operating Characteristics when Addressing the Mode and Control Registers . . . . . . . . . . . . . . . . . . . . . 22 Main ADC Control Register (AD0CON): (A3, A2, A1, A0 = 0, 0, 1, 0; Power-On Reset = 0x07) . . . . . . . . . . . . . . . . . . . . . . . 22 Aux ADC Control Registers (AD1CON): (A3, A2, A1, A0 = 0, 0, 1, 1; Power-On Reset = 0x01) . . . . . . . . . . . . . . . . . . . . . . . 23 Filter Register (A3, A2, A1, A0 = 0, 1, 0, 0; Power-On Reset = 0x45) . . . . . . . . . . . . . . . . . . . . . . . 24 I/O and Current Source Control Register (IOCON): (A3, A2, A1, A0 = 0, 1, 1, 1; Power-On Reset = 0x0000) . . . . . . . . . . . . . . . . . . . . . 24 Main ADC Data Result Registers (DATA0): (A3, A2, A1, A0 = 0, 1, 0, 1; Power-On Reset = 0x00 0000) . . . . . . . . . . . . . . . . . . . 26 Aux ADC Data Result Registers (DATA1): (A3, A2, A1, A0 = 0, 1, 1, 0; Power-On Reset = 0x0000) . . . . . . . . . . . . . . . . . . . . . 26 Main ADC Offset Calibration Coefficient Registers (OF0): (A3, A2, A1, A0 = 1, 0, 0, 0; Power-On Reset = 0x80 0000) . . . . . . . . . . . . . . . . . . . 26 Aux ADC Offset Calibration Coefficient Registers (OF1): (A3, A2, A1, A0 = 1, 0, 0, 1; Power-On Reset = 0x8000) . . . . . . . . . . . . . . . . . . . . . 26 Main ADC Gain Calibration Coefficient Registers (GNO): (A3, A2, A1, A0 = 1, 0, 1, 0; Power-On Reset = 0x5X XXX5) . . . . . . . . . . . . . . . . . 26 Aux ADC Gain Calibration Coefficient Registers (GN1): (A3, A2, A1, A0 = 1, 0, 1, 1; Power-On Reset = 0x59XX) . . . . . . . . . . . . . . . . . . . . . 26 ID Register (ID): (A3, A2, A1, A0 = 1, 1, 1, 1; Power-On Reset = 0x0X) . . . . . . . . . . . . . . . . . . . . . . . 26 User Nonprogrammable Test Registers . . . . . . . . . . . . . . 26 CONFIGURING THE AD7719 . . . . . . . . . . . . . . . . . . . . . 27 MICROCOMPUTER/MICROPROCESSOR INTERFACING . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 AD7719-to-68HC11 Interface . . . . . . . . . . . . . . . . . . . . . 29 AD7719-to-8051 Interface . . . . . . . . . . . . . . . . . . . . . . . . 29 AD7719-to-ADSP-2103/ADSP-2105 Interface . . . . . . . . 30 CIRCUIT DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . 30 Analog Input Channels . . . . . . . . . . . . . . . . . . . . . . . . . . 31 Programmable Gain Amplifier . . . . . . . . . . . . . . . . . . . . . 32 Bipolar/Unipolar Configuration . . . . . . . . . . . . . . . . . . . . 32 Data Output Coding . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 Burnout Currents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 Excitation Currents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 Crystal Oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 Reference Input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 Reference Detect . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 Reset Input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 Power-Down Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 Idle Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 ADC Disable Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 Calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 Grounding and Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . 35 APPLICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35 Pressure Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 Temperature Measurement . . . . . . . . . . . . . . . . . . . . . . . 36 3-Wire RTD Configurations . . . . . . . . . . . . . . . . . . . . . . 37 Smart Transmitters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38 OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . 39 REVISION HISTORY . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40 TABLE OF CONTENTS REV. A –3– AD7719 (AVDD = 2.7 V to 3.6 V or 4.75 V to 5.25 V, DVDD = 2.7 V to 3.6 V or 4.75 V to 5.25 V, REFIN(+) = 2.5 V; REFIN(–) = AGND; AGND = DGND = 0 V; XTAL1/XTAL2 = 32.768 kHz Crystal; all specifications TMIN to TMAX, unless otherwise noted.) Parameter AD7719B Unit Test Conditions ADC CHANNEL SPECIFICATION Output Update Rate 5.4 Hz min Both Channels Synchronized 105 Hz max 0.732 ms Increments MAIN CHANNEL No Missing Codes2 24 Bits min 20 Hz Update Rate Resolution 13 Bits p-p ±20 mV Range, 20 Hz Update Rate 18 Bits p-p ±2.56 V Range, 20 Hz Update Rate Output Noise and Update Rates See Tables II to V Integral Nonlinearity ±10 ppm of FSR max Typically 2 ppm. Offset Error3 ±3 μV typ Offset Error Drift vs. Temperature4 ±10 nV/°C typ Full-Scale Error5, 6, 7 ±10 μV typ At the Calibrated Conditions Gain Drift vs. Temperature4 ±0.5 ppm/°C typ Power Supply Rejection (PSR) 80 dB min Input Range = ±2.56 V, 100 dB typ. 110 dB typ on ±20 mV Range ANALOG INPUTS Differential Input Voltage Ranges ±1.024 × REFIN1/GAIN V nom REFIN1 = REFIN1(+) – REFIN1(–) GAIN = 1 to 128. ADC Range Matching ±2 μV typ Input Voltage = 19 mV on All Ranges Absolute AIN Voltage Limits AGND + 100 mV V min BUF = 0; Buffered Mode of Operation AVDD – 100 mV V max Analog Input Current2 BUF = 0 DC Input Current ±1 nA max DC Input Current Drift ±5 pA /°C typ Absolute AIN Voltage Limits AGND – 30 mV V min BUF = 1; Unbuffered mode of operation. AVDD + 30 mV V max Analog Input Current BUF = 1. Unbuffered Mode of Operation. DC Input Current ±125 nA/V typ Input Current Varies with Input Voltage DC Input Current Drift ±2 pA/V/°C typ Normal-Mode Rejection2, 8 @ 50 Hz 100 dB min 50 Hz ± 1 Hz, 16.65 Hz Update Rate, SF = 82 @ 60 Hz 100 dB min 60 Hz ± 1 Hz, 20 Hz Update Rate, SF = 68 Common-Mode Rejection @ DC 90 dB min Input Range = ±2.56 V, AIN = 1 V. 100 dB typ. 110 dB typ on ±20 mV Range @ 50 Hz2 100 dB min 50 Hz ± 1 Hz, Range = ±2.56 V, AIN = 1 V @ 60 Hz2 100 dB min 60 Hz ± 1 Hz, Range = ±2.56 V, AIN = 1 V REFERENCE INPUT (REFIN1) REFIN1 Voltage 2.5 V nom REFIN1 = REFIN1(+) – REFIN1(–) REFIN1 Voltage Range2 1 V min AVDD V max REFIN1 Common-Mode Range AGND – 30 mV V min AVDD + 30 mV V max Reference DC Input Current 0.5 μA/V typ Reference DC Input Current Drift ±0.01 nA/V/°C typ Normal-Mode Rejection2, 8 @ 50 Hz 100 dB min 50 Hz ± 1 Hz, SF = 82 @ 60 Hz 100 dB min 60 Hz ± 1 Hz, SF = 68 Common-Mode Rejection @ DC 110 dB typ Input Range = ±2.56 V, AIN = 1 V @ 50 Hz 110 dB typ 50 Hz ± 1 Hz, Range = 2.56 V, AIN = 1 V @ 60 Hz 110 dB typ 60 Hz ± 1 Hz, Range = 2.56 V, AIN = 1 V Reference Detect Levels 0.3 V min NOXREF Bit Active if VREF < 0.3 V 0.65 V max NOXREF Bit Inactive if VREF > 0.65 V AUXILIARY CHANNEL No Missing Codes2 16 Bits min Resolution 16 Bits p-p ±2.5 V Range, 20 Hz Update Rate Output Noise and Update Rates See Tables VI and VIII Integral Nonlinearity ±15 ppm of FSR max FSR REFIN Gain = 2 × 1.024 1 –SPECIFICATIONS1 REV. A AD7719 –4– Parameter AD7719B Unit Test Conditions AUXILIARY CHANNEL (continued) Offset Error3 ±3 μV typ Selected Channel = AIN5/AIN6 Offset Error Drift vs. Temperature4 ±10 nV/°C typ Full-Scale Error6, 7 ±0.75 LSB typ Gain Drift vs. Temperature4 0.5 ppm/°C typ Negative Full-Scale Error ±1 LSB typ Power Supply Rejection (PSR) 70 dB min AIN = 1 V Input Range = ±2.5 V, Typically 80 dB ANALOG INPUTS Differential Input Voltage Ranges ±REFIN2 V nom ARN = 1 ±REFIN2/2 V nom ARN = 0 Absolute AIN Voltage Limits AGND – 30 mV V min Unbuffered Input AVDD + 30 mV V max Analog Input Current DC Input Current ±125 nA/V typ Input Current Varies with Input Voltage DC Input Current Drift ±2 pA/V/°C typ Normal-Mode Rejection2, 8 @ 50 Hz 100 dB min 50 Hz ±1 Hz, SF = 82 @ 60 Hz 100 dB min 60 Hz ±1 Hz, SF = 68 Common-Mode Rejection @ DC 85 dB min Input Range = ±2.5 V, AIN = 1 V @ 50 Hz2 90 dB min 50 Hz ±1 Hz, Range = 2.5 V, AIN = 1 V @ 60 Hz2 90 dB min 60 Hz ±1 Hz, Range = 2.5 V, AIN = 1 V REFERENCE INPUT (REFIN2) With Respect to AGND REFIN2 Voltage 2.5 V nom REFIN2 Range2 1 V min AVDD V max Reference DC Input Current2 0.2 μA/V typ Reference DC Input Current Drift 0.003 nA/V/°C typ EXCITATION CURRENT SOURCES (IEXC1 and IEXC2) Output Current 200 μA nom Initial Tolerance at 25°C ±10 % typ Drift 200 ppm/°C typ Initial Current Matching at 25°C ±1 % typ Matching between IEXC1 and IEXC2 No Load Drift Matching 20 ppm/°C typ Line Regulation (AVDD) 2.1 μA/V max AVDD = 5 V ± 5%. Typically 1.25 μA/V Load Regulation 300 nA/V typ Output Compliance AVDD – 0.6 V max AGND – 30 mV V min LOW-SIDE POWER SWITCHES (SW1, SW2) RON 5 Ω max AVDD = 5 V. Typically 3 Ω 7 Ω max AVDD = 3 V. Typically 4.5 Ω Allowable Current2 20 mA max Continuous Current per Switch TEMPERATURE SENSOR Accuracy See TPC 5 °C typ TRANSDUCER BURNOUT AIN(+) Current –100 nA typ AIN(–) Current 100 nA typ Initial Tolerance @ 25°C ±15 % typ Drift 0.03 %/°C typ SYSTEM CALIBRATION2, 9 Full-Scale Calibration Limit 1.05 × FS10 V max Zero-Scale Calibration Limit –1.05 × FS V min Input Span 0.8 × FS V min 2.1 × FS V max REV. A –5– AD7719 Parameter AD7719B Unit Test Conditions LOGIC INPUTS All Inputs Except SCLK and XTAL12 VINL, Input Low Voltage 0.8 V max DVDD = 5 V 0.4 V max DVDD = 3 V VINH, Input High Voltage 2.0 V min DVDD = 3 V or 5 V SCLK Only (Schmitt-Triggered Input)2 VT(+) 1.4/2 V min/V max DVDD = 5 V VT(–) 0.8/1.4 V min/V max DVDD = 5 V VT(+) – VT(–) 0.3/0.85 V min/V max DVDD = 5 V VT(+) 0.95/2 V min/V max DVDD = 3 V VT(–) 0.4/1.1 V min/V max DVDD = 3 V VT(+) – VT(–) 0.3/0.85 V min/V max DVDD = 3 V XTAL1 Only2 VINL, Input Low Voltage 0.8 V max DVDD = 5 V VINH, Input High Voltage 3.5 V min DVDD = 5 V VINL, Input Low Voltage 0.4 V max DVDD = 3 V VINH, Input High Voltage 2.5 V min DVDD = 3 V Input Currents ±10 μA max VIN = DVDD –70 μA max VIN = DGND, Typically –40 μA at 5 V and –20 μA at 3 V Input Capacitance2 10 pF typ All Digital Inputs LOGIC OUTPUTS (Excluding XTAL2) VOH, Output High Voltage2 DVDD – 0.6 V min DVDD = 3 V, ISOURCE = 100 μA VOL, Output Low Voltage2 0.4 V max DVDD = 3 V, ISINK = 100 μA VOH, Output High Voltage2 4 V min DVDD = 5 V, ISOURCE = 200 μA VOL, Output Low Voltage 2 0.4 V max DVDD = 5 V, ISINK = 1.6 mA Floating-State Leakage Current ±10 μA max Floating-State Output Capacitance ±10 pF typ Data Output Coding Binary Unipolar Mode Offset Binary Bipolar Mode I/O PORT11 I/O Port Voltages Are with Respect to AVDD and AGND VINL, Input Low Voltage2 0.8 V max AVDD = 5 V 0.4 V max AVDD = 3 V VINH, Input High Voltage2 2.0 V min AVDD = 3 V or 5 V Input Currents ±10 μA max VIN = AVDD –70 μA max VIN = AGND, Typically –40 μA at AVDD = 5 V and –20 μA at AVDD = 3 V Input Capacitance 10 pF typ All Digital Inputs VOH, Output High Voltage2 AVDD – 0.6 V min AVDD = 3 V, ISOURCE = 100 μA VOL, Output Low Voltage2 0.4 V max AVDD = 3 V, ISINK = 100 μA VOH, Output High Voltage2 4 V min AVDD = 5 V, ISOURCE = 200 μA VOL, Output Low Voltage2 0.4 V max AVDD = 5 V, ISINK = 1.6 mA Floating-State Output Leakage Current ±10 μA max Floating-State Output Capacitance ±10 pF typ START-UP TIME From Power-On 300 ms typ From Idle Mode 1 ms typ From Power-Down Mode 1 ms typ Osc. Active in Power-Down 300 ms typ Osc. Powered Down POWER REQUIREMENTS Power Supply Voltages AVDD – AGND 2.7/3.6 V min/max AVDD = 3 V nom 4.75/5.25 V min/max AVDD = 5 V nom DVDD – DGND 2.7/3.6 V min/max DVDD = 3 V nom 4.75/5.25 V min DVDD = 5 V nom Power Supply Currents DIDD Current (Normal Mode)12 0.6 mA max DVDD = 3 V, 0.5 mA typ 0.75 mA max DVDD = 5 V, 0.6 mA typ REV. A AD7719 –6– Parameter AD7719B Unit Test Conditions Power Supply Currents (Continued) AIDD Current (Main ADC) 1.1 mA max AVDD = 3 V or 5 V, Buffered Mode, 0.85 mA typ 0.55 mA max AVDD = 3 V or 5 V, Unbuffered Mode, 0.45 mA typ AIDD Current (Aux ADC) 0.3 mA max AVDD = 3 V or 5 V, 0.25 mA typ AIDD Current (Main and Aux ADC) 1.25 mA max AVDD = 3 V or 5 V, Main ADC Buffered, 1 mA typ DIDD (ADC Disable Mode)13 0.35 mA max DVDD = 3 V, 0.25 mA typ 0.4 mA max DVDD = 5 V, 0.3 mA typ AIDD (ADC Disable Mode) 0.15 mA max AVDD = 3 V or 5 V DIDD (Power-Down Mode) 10 μA max DVDD = 3 V, 32.768 kHz Osc. Running 2 μA max DVDD = 3 V, Oscillator Powered Down 30 μA max DVDD = 5 V, 32.768 kHz Osc. Running 8 μA max DVDD = 5 V, Oscillator Powered Down AIDD (Power-Down Mode) 1 μA max AVDD = 3 V or 5 V NOTES 1Temperature range –40°C to +85°C. 2Guaranteed by design and/or characterization data on production release. 3System zero calibration will remove this error. 4A calibration at any temperature will remove this drift error. 5The main ADC is factory-calibrated with AVDD = DVDD = 4 V, TA = 25°C, REFIN1(+) – REFIN1(–) = 2.5 V. If the user power supplies or temperature conditions are significantly different from these, internal full-scale calibration will restore this error to the published specification. System calibration can be used to reduce this error to the order of the noise. Full-scale error applies to both positive and negative full scale. 6A system full-scale calibration will remove this error. 7A typical gain error of ±10 μV results following a user self-calibration. 8Simultaneous 50 Hz and 60 Hz rejection is achieved using 19.8 Hz (SF = 69) update rate. Normal mode rejection in this case is 60 dB min. 9After a calibration if the analog input exceeds positive full scale, the converter will output all 1s. If the analog input is less than negative full scale, the device will output all 0s. 10FS = Full-Scale Input. FS = 1.024 × REFIN1/Gain on the main ADC, where REFIN1 = REFIN1(+) – REFIN1(–). FS = REFIN2 on the aux ADC when ARN = 1 in the aux ADC control register (AD1CON) and REFIN2/2 on the aux ADC when ARN = 0. 11 Input and output levels on the I/O Port are with respect to AVDD and AGND. 12Normal mode refers to the case where both main and aux ADCs are running. 13ADC disable is entered by setting both the AD0EN and AD1EN bits in the main and aux ADC control registers to a 0 and setting the mode bits (MD2, MD1, MD0) in the mode register to non-0. Specifications subject to change without notice. ABSOLUTE MAXIMUM RATINGS1 (TA = 25°C, unless otherwise noted.) AVDD to AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V AVDD to DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V DVDD to AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V DVDD to DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V AGND to DGND2. . . . . . . . . . . . . . . . . . . –20 mV to +20 mV PWRGND to AGND . . . . . . . . . . . . . . . . –20 mV to +20 mV AVDD to DVDD . . . . . . . . . . . . . . . . . . . . . . . . . . –5 V to +5 V Analog Input Voltage to AGND . . . . –0.3 V to AVDD +0.3 V Reference Input Voltage to AGND . . –0.3 V to AVDD +0.3 V Total AIN/REFIN Current (Indefinite) . . . . . . . . . . . . 30 mA Digital Input Voltage to DGND . . . . –0.3 V to DVDD +0.3 V Digital Output Voltage to DGND . . . –0.3 V to DVDD +0.3 V Operating Temperature Range . . . . . . . . . . . –40°C to +85°C Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C SOIC Package θJA Thermal Impedance . . . . . . . . . . . . . . . . . . . . 71.4°C/W θJC Thermal Impedance . . . . . . . . . . . . . . . . . . . . . . 23°C/W TSSOP Package θJA Thermal Impedance . . . . . . . . . . . . . . . . . . . . 97.9°C/W θJC Thermal Impedance . . . . . . . . . . . . . . . . . . . . . . 14°C/W Lead Temperature, Soldering Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . . . . 215°C Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220°C NOTES 1 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sections of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 AGND and DGND are connected internally within the AD7719. ORDERING GUIDE Model Temperature Range Package Description Package Option AD7719BR –40°C to +85°C SOIC R-28 AD7719BRU –40°C to +85°C TSSOP RU-28 EVAL-AD7719EB Evaluation Board REV. A –7– AD7719 TIMING CHARACTERISTICS1, 2 Limit at TMIN, TMAX Parameter (B Version) Unit Conditions/Comments t1 32.768 kHz typ Crystal Oscillator Frequency t2 50 ns min RESET Pulsewidth Read Operation t3 0 ns min RDY to CS Setup Time t4 0 ns min CS Falling Edge to SCLK Active Edge Setup Time3 t5 4 0 ns min SCLK Active Edge to Data Valid Delay3 60 ns max DVDD = 4.75 V to 5.25 V 80 ns max DVDD = 2.7 V to 3.6 V t5A 4, 5 0 ns min CS Falling Edge to Data Valid Delay3 60 ns max DVDD = 4.75 V to 5.25 V 80 ns max DVDD = 2.7 V to 3.6 V t6 100 ns min SCLK High Pulsewidth t7 100 ns min SCLK Low Pulsewidth t8 0 ns min CS Rising Edge to SCLK Inactive Edge Hold Time3 t9 6 10 ns min Bus Relinquish Time after SCLK Inactive Edge3 80 ns max t10 100 ns max SCLK Active Edge to RDY High3, 7 Write Operation t11 0 ns min CS Falling Edge to SCLK Active Edge Setup Time3 t12 30 ns min Data Valid to SCLK Edge Setup Time t13 25 ns min Data Valid to SCLK Edge Hold Time t14 100 ns min SCLK High Pulsewidth t15 100 ns min SCLK Low Pulsewidth t16 0 ns min CS Rising Edge to SCLK Edge Hold Time NOTES 1Sample tested during initial release to ensure compliance. All input signals are specified with t R = tF = 5 ns (10% to 90% of DVDD) and timed from a voltage level of 1.6 V. 2See Figures 2 and 3. 3SCLK active edge is falling edge of SCLK. 4These numbers are measured with the load circuit of Figure 1 and defined as the time required for the output to cross the VOL or VOH limits. 5This specification only comes into play if CS goes low while SCLK is low. It is required primarily for interfacing to DSP machines. 6These numbers are derived from the measured time taken by the data output to change 0.5 V when loaded with the circuit of Figure 1. The measured number is then extrapolated back to remove effects of charging or discharging the 50 pF capacitor. This means that the times quoted in the Timing Characteristics are the true bus relinquish times of the part and as such are independent of external bus loading capacitances. 7RDY returns high after a read of both ADCs. The same data can be read again, if required, while RDY is high, although care should be taken that subsequent reads do not occur close to the next output update. Specifications subject to change without notice. (AVDD = 2.7 V to 3.6 V or AVDD = 4.75 V to 5.25 V; DVDD = 2.7 V to 3.6 V or DVDD = 4.75 V to 5.25 V; AGND = DGND = 0 V; XTAL = 32.768 kHz; Input Logic 0 = 0 V, Logic 1 = DVDD, unless otherwise noted.) ISINK (1.6mA WITH DVDD = 5V 100A WITH DVDD = 3V) 1.6V ISOURCE (200A WITH DVDD = 5V 100A WITH DVDD = 3V) TO OUTPUT PIN 50pF Figure 1. Load Circuit for Timing Characterization CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD7719 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. WARNING! ESD SENSITIVE DEVICE REV. A AD7719 –8– DIGITAL INTERFACE As previously outlined, the AD7719’s programmable functions are controlled using a set of on-chip registers. Data is written to these registers via the part’s serial interface; read access to the on-chip registers is also provided by this interface. All communications to the part must start with a write operation to the Communications register. After power-on or RESET, the device expects a write to its Communications register. The data written to this register determines whether the next operation to the part is a read or a write operation and also determines to which register this read or write operation occurs. Therefore, write access to any of the other registers on the part starts with a write operation to the Communications register followed by a write to the selected register. A read operation from any other register on the part (including the output data register) starts with a write operation to the Communications register followed by a read operation from the selected register. The AD7719’s serial interface consists of five signals: CS, SCLK, DIN, DOUT, and RDY. The DIN line is used for transferring data into the on-chip registers while the DOUT line is used for accessing data from the on-chip registers. SCLK is the serial clock input for the device, and all data transfers (either on DIN or DOUT) take place with respect to this SCLK signal. The RDY line is used as a status signal to indicate when data is ready to be read from the AD7719’s data register. RDY goes low when a new data-word is available in the output register of either the main or aux ADCs. It is reset high when a read operation from the data register is complete. It also goes high prior to the updating of the output register to indicate when not to read from the device to ensure that a data read is not attempted while the register is being updated. CS is used to select the device. It can be used to decode the AD7719 in systems where a number of parts are connected to the serial bus. Figures 2 and 3 show timing diagrams for interfacing to the AD7719 with CS used to decode the part. Figure 3 is for a read operation from the AD7719’s output shift register while Figure 2 shows a write operation to the input shift register. It is possible to read the same data twice from the output register even though the RDY line returns high after the first read operation. Care must be taken, however, to ensure that the read operations have been completed before the next output update is about to take place. The AD7719 serial interface can operate in 3-wire mode by tying the CS input low. In this case, the SCLK, DIN, and DOUT lines are used to communicate with the AD7719, and the status of RDY bits (RDY0 and RDY1) can be obtained by interrogating the STATUS register. This scheme is suitable for interfacing to microcontrollers. If CS is required as a decoding signal, it can be generated from a port bit. For microcontroller interfaces, it is recommended that the SCLK idles high between data transfers. The AD7719 can also be operated with CS used as a frame synchronization signal. This scheme is suitable for DSP interfaces. In this case, the first bit (MSB) is effectively clocked out by CS since CS would normally occur after the falling edge of SCLK in DSPs. The SCLK can continue to run between data transfers provided the timing numbers are obeyed. t12 t13 t14 t15 t11 t16 MSB LSB CS SCLK DIN Figure 2. Write Cycle Timing Diagram t5 t5A t4 t6 t3 t9 MSB LSB CS SCLK t8 t10 t7 t6 DOUT RDY Figure 3. Read Cycle Timing Diagram REV. A AD7719 –9– PIN CONFIGURATION 14 13 12 11 10 17 16 15 19 18 20 28 27 26 25 24 23 22 21 9 8 1 2 3 4 7 6 5 TOP VIEW (Not to Scale) AD7719 IOUT1 DGND DVDD XTAL2 XTAL1 IOUT2 AVDD AGND RDY DOUT REFIN1(–) DIN REFIN1(+) AIN1 AIN2 AIN3 AIN4 RESET SCLK CS AIN5 AIN6 REFIN2 P4 P1/SW1 P3 P2/SW2 PWRGND PIN FUNCTION DESCRIPTIONS Pin No. Mnemonic Function 1 IOUT1 Output for Internal 200 μA Excitation Current Source. Current source IEXC1 and/or IEXC2 can be switched to this output. 2 IOUT2 Output for Internal 200 μA Excitation Current Source. Current source IEXC1 and/or IEXC2 can be switched to this output. 3 AVDD Analog Supply Voltage. 4 AGND Analog Ground. 5 REFIN1(–) Negative Reference Input for Main ADC Channel. This reference input can lie anywhere between AGND and AVDD – 1 V. 6 REFIN1(+) Positive Reference Input for Main ADC Channel. REFIN1(+) can lie anywhere between AVDD and AGND + 1 V. The nominal reference voltage (REFIN1(+) – REFIN1(–)) is 2.5 V, but the part is functional with a reference range from 1 V to AVDD. 7 AIN1 Analog Input. AIN1 is dedicated to the main channel. 8 AIN2 Analog Input. AIN2 is dedicated to the main channel. 9 AIN3 Analog Input. AIN3 can be multiplexed to either the main or auxiliary channel. 10 AIN4 Analog Input. AIN4 can be multiplexed to either the main or auxiliary channel. 11 AIN5 Analog Input. AIN5 is dedicated to the auxiliary channel and is referenced to AIN6 or AGND. 12 AIN6 Analog Input. AIN6 is dedicated to the auxiliary channel. It forms a differential input pair with AIN5 in fully differential input mode or is referenced to AGND in pseudodifferential mode. 13 REFIN2 Single-Ended Reference Input for Auxiliary Channel. The nominal input reference is 2.5 V. The auxiliary channel will function with an input reference range from 1 V to AVDD. 14 P4 General-Purpose I/O Bit. The input and output voltage levels are referenced to AVDD and AGND. 15 P3 General-Purpose I/O Bit. The input and output voltage levels are referenced to AVDD and AGND. The serial interface can be reset by exercising the RESET input on the part. It can also be reset by writing a series of 1s on the DIN input. If a logic 1 is written to the AD7719 DIN line for at least 32 serial clock cycles, the serial interface is reset. This ensures that in 3-wire systems, if the interface gets lost, either via a software error or by some glitch in the system, it can be reset back to a known state. This state returns the interface to where the AD7719 is expecting a write operation to its Communications register. This operation resets the contents of all registers to their power-on reset values. Some microprocessor or microcontroller serial interfaces have a single serial data line. In this case, it is possible to connect the AD7719’s DATA OUT and DATA IN lines together and connect them to the single data line of the processor. A 10 kΩ pull-up resistor should be used on this single data line. In this case, if the interface gets lost, because the read and write operations share the same line, the procedure to reset it to a known state is somewhat different than previously described. It requires a read operation of 24 serial clocks followed by a write operation where a logic 1 is written for at least 32 serial clock cycles to ensure that the serial interface is back in a known state. REV. A AD7719 –10– PIN FUNCTION DESCRIPTIONS (continued) Pin No. Mnemonic Function 16 P2/SW2 Dual-Purpose Pin. It can act as a general-purpose output (P2) bit referenced between AVDD and AGND or as a low-side power switch (SW2) to PWRGND. 17 PWRGND Ground Point for the Low-Side Power Switches SW2 and SW1. PWRGND must be tied to AGND. 18 P1/SW1 Dual-Purpose Pin. It can act as a general-purpose output (P1) bit referenced between AVDD and AGND or as a low-side power switch (SW1) to PWRGND. 19 RESET Digital Input Used to Reset the ADC to Its Power-On Reset Status. This pin has a weak pull-up internally to DVDD. 20 SCLK Serial Clock Input for Data Transfers to and from the ADC. The SCLK has a Schmitt-triggered input, making the interface suitable for opto-isolated applications. The serial clock can be continuous with all data transmitted in a continuous train of pulses. Alternatively, it can be a noncontinuous clock with the information being transmitted to or from the AD7719 in smaller batches of data. A weak pull-up to DVDD is provided on the SCLK input. 21 CS Chip Select Input. This is an active low logic input used to select the AD7719. CS can be used to select the AD7719 in systems with more than one device on the serial bus or as a frame synchronization signal in communicating with the device. CS can be hardwired low, allowing the AD7719 to be operated in 3-wire mode with SCLK, DIN, and DOUT used to interface with the device. A weak pull-up to DVDD is provided on the CS input. 22 RDY RDY is a logic low status output from the AD7719. RDY is low if either the main ADC or auxiliary ADC channel has valid data in its data register. This output returns high on completion of a read operation from the data register. If data is not read, RDY will return high prior to the next update, indicating to the user that a read operation should not be initiated. The RDY pin also returns low following the completion of a calibration cycle. The RDY pin is effectively the digital NOR function of the RDY0 and RDY1 bits in the Status register. If one of the ADCs is disabled, the RDY pin reflects the active ADC. RDY does not return high after a calibration until the mode bits are written to, enabling a new conversion or calibration. Since the RDY pin provides information on both the main and aux ADCs, when either the main or aux ADC is disabled, it is recommended to immediately read its data register to ensure that its RDY bit goes inactive and releases the RDY pin to indicate output data updates on the remaining active ADC. 23 DOUT Serial Data Output Accessing the Output Shift Register of the AD7719. The output shift register can contain data from any of the on-chip data, calibration, or control registers. 24 DIN Serial Data Input Accessing the Input Shift Register on the AD7719. Data in this shift register is transferred to the calibration or control registers within the ADC depending on the selection bits of the Communications register. A weak pull-up to DVDD is provided on the DIN input. 25 DGND Ground Reference Point for the Digital Circuitry. 26 DVDD Digital Supply Voltage, 3 V or 5 V Nominal. 27 XTAL2 Output from the 32 kHz Crystal Oscillator Inverter. 28 XTAL1 Input to the 32 kHz Crystal Oscillator Inverter. REV. A AD7719 –11– Typical Performance Characteristics– READING NO. 8389600 8389400 8388000 0 100 1000 CODE READ 200 300 8389200 400 500 600 700 800 900 8389000 8388800 8388600 8388400 8388200 AVDD = DVDD = 5V INPUT RANGE = 20mV REFIN1(+)–REFIN1(–) = 2.5V UPDATE RATE = 19.79Hz MAIN ADC IN BUFFERED MODE RMS NOISE = 0.58V rms TA = 25C VREF = 2.5V TPC 1. Typical Noise Plot on ±20 mV Input Range with 19.79 Hz Update Rate 8 7 0 8388039 8388721 8388687 8388657 8388615 8388579 8388547 8388499 8388449 8388382 8388754 8389110 8389033 8388985 8388941 8388906 8388874 8388841 8388805 8388779 6 5 4 3 2 1 9 TPC 2. Noise Distribution Histogram 2.5 0 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 2.0 1.5 1.0 0.5 3.0 VREF (V) RMS NOISE (V) 20mV RANGE 2.56V RANGE AVDD = DVDD = 5V VREF = 2.5V INPUT RANGE = 2.56V UPDATE RATE = 19.79Hz TA = 25C TPC 3. RMS Noise vs. Reference Input 16 0 10 20 30 40 50 60 70 80 90 100 24 22 20 18 26 UPDATE RATE (Hz) NO MISSING CODES (Min) 110 TPC 4. No-Missing-Codes Performance TEMPERATURE SENSOR ( C) 1400 1200 0 10 20 30 40 50 HITS 800 600 400 200 1000 THE AMBIENT TEMPERATURE VARIES FROM 25C TO 30C WHILE RECORDING THE DATA FROM THE DEVICES. TPC 5. Temperature Sensor Accuracy MAIN CAL ACC. @ 4V (V) 1200 0 –20 –10 0 10 20 HITS 800 600 400 200 1000 –30 TPC 6. Full-Scale Error Distribution REV. A AD7719 –12– DUAL-CHANNEL ADC CIRCUIT INFORMATION Overview The AD7719 incorporates two independent Σ-Δ ADC channels (main and auxiliary) with on-chip digital filtering intended for the measurement of wide dynamic range, low frequency signals such as those in weigh-scale, strain gage, pressure transducer, or temperature measurement applications. Main Channel This channel is intended to convert the primary sensor input. This channel can be operated in buffered or unbuffered mode, and can be programmed to have one of eight input voltage ranges from ±20 mV to ±2.56 V. This channel can be configured as either two fully differential inputs (AIN1/AIN2 and AIN3/AIN4) or three pseudodifferential input channels (AIN1/AIN4, AIN2/ AIN4, and AIN3/AIN4). Buffering the input channel means that the part can accommodate significant source impedances on the analog input and that R, C filtering (for noise rejection or RFI reduction) can be placed on the analog inputs if required. Operating in unbuffered mode leads to lower power consumption in low power applications, but care must be exercised in unbuffered mode because source impedances can introduce gain errors. The main ADC also features sensor burnout currents that can be switched on and off. These currents can be used to check that a transducer is still operational before attempting to take measurements. The ADC employs a Σ-Δ conversion technique to realize up to 24 bits of no-missing-codes performance. The Σ-Δ modulator converts the sampled input signal into a digital pulse train whose duty cycle contains the digital information. A Sinc3 programmable low-pass filter is then employed to decimate the modulator output data stream to give a valid data conversion result at programmable output rates from 5.35 Hz (186.77 ms) to 105.03 Hz (9.52 ms). A chopping scheme is also employed to minimize ADC channel offset errors. A block diagram of the main ADC input channel is shown in Figure 4. The sampling frequency of the modulator loop is many times higher than the bandwidth of the input signal. The integrator in the modulator shapes the quantization noise (which results from the analog-to-digital conversion) so that the noise is pushed toward one-half of the modulator frequency. The output of the Σ-Δ modulator feeds directly into the digital filter. The digital filter then band-limits the response to a frequency significantly lower than one-half of the modulator frequency. In this manner, the 1-bit output of the comparator is translated into a band-limited, low noise output from the AD7719 ADC. The AD7719 filter is a low-pass, Sinc3, or (SIN(x)/x)3 filter whose primary function is to remove the quantization noise introduced at the modulator. The cutoff frequency and decimated output data rate of the filter are programmable via the SF word loaded to the filter register. A chopping scheme is employed where the complete signal chain is chopped, resulting in excellent dc offset and offset drift specifications, and is extremely beneficial in applications where drift, noise rejection, and optimum EMI rejection are important factors. With chopping, the ADC repeatedly reverses its inputs. The decimated digital output words from the Sinc3 filters therefore have a positive offset and negative offset term included. As a result, a final summing stage is included so that each output word from the filter is summed and averaged with the previous filter output to produce a new valid output result to be written to the ADC data register. Auxiliary Channel The Auxiliary (Aux) channel is intended to convert supplementary inputs such as those from a cold junction diode or thermistor. This channel is unbuffered and has an input range of ±REFIN2 or ±REFIN2/2, determined by the ARN bit in the auxiliary ADC control register (AD1CON). AIN3 and AIN4 can be multiplexed into the auxiliary channel as single-ended inputs with respect to AGND, while AIN5 and AIN6 can operate as a differential input pair. With AIN6 tied to AGND, AIN5 can be operated as an additional single-ended input. A block diagram of the auxiliary ADC channel is shown in Figure 5. SINC3 FILTER MUX BUF PGA MOD0 ANALOG XOR INPUT DIGITAL OUTPUT 1 8 SF 3 ( ) 3 (8 SF ) 12 AIN + VOS AIN – VOS fCHOP fIN fMOD fCHOP fADC - Figure 4. Main ADC Channel Block Diagram OSCILLATOR AVDD = DVDD = 5V TA = 25C TIME BASE = 100ms/DIV TRACE 1 = TRACE 2 = 2V/DIV VDD TPC 7. Typical Oscillator Power-Up REV. A AD7719 –13– Both Channels The operation of the aux channel is identical to the main channel with the exception that there is no PGA on the aux channel. The input chopping is incorporated into the input multiplexer while the output chopping is accomplished by an XOR gate at the output of the modulator. The chopped modulator bit stream is applied to a Sinc3 filter. The programming of the Sinc3 decimation factor is restricted to an 8-bit register SF; the actual decimation factor is the register value times 8. The decimated output rate from the Sinc3 filter (and the ADC conversion rate) will therefore be f SF ADC = × fMOD × × 1 3 1 8 where: fADC is the ADC update rate. SF is the decimal equivalent of the word loaded to the filter register. fMOD is the modulator sampling rate of 32.768 kHz. Programming the filter register determines the update rate for both the main and aux ADC. Both ADCs operate with the same update rate. The chop rate of the channel is half the output data rate. The frequency response of the filter H (f) is as follows: 1 8 8 1 2 2 3 SF SF f f f f f f f f MOD MOD OUT × OUT × × × × × × × × × × sin ( / ) sin ( / ) sin ( / ) sin ( / ) π π π π where: fMOD = 32,768 Hz SF = value programmed into SF SFR. fOUT = fMOD/(SF × 8 × 3) The following shows plots of the filter frequency response for the SF words shown in Table I. The overall frequency response is the product of a Sinc3 and a sinc response. There are Sinc3 notches at integer multiples of 3 × fADC and there are sinc notches SINC3 FILTER MUX - MOD1 ANALOG XOR INPUT DIGITAL OUTPUT 3 (8 SF ) 12 AIN + VOS AIN – VOS fCHOP fMOD fCHOP fADC 1 8 SF 3 ( ) Figure 5. Auxiliary ADC Channel Block Diagram at odd integer multiples of fADC/2. The 3 dB frequency for all values of SF obeys the following equation: f (3 dB) = 0.24 × fADC The signal chain is chopped as shown in Figures 4 and 5. The chop frequency is f f CHOP = ADC 2 As shown in the block diagram, the Sinc3 filter outputs alternately contain +VOS and –VOS, where VOS is the respective channel offset. This offset is removed by performing a running average of 2. This average by 2 means that the settling time to any change in programming of the ADC will be twice the normal conversion time, while an asynchronous step change on the analog input will not be fully reflected until the third subsequent output. t f SETTLE t ADC = ADC = × 2 2 The allowable range for SF is 13 to 255, with a default of 69 (0x45). The corresponding conversion rates, conversion times, and settling times are tabulated in Table I. Note that the conversion time increases by 0.732 ms for each increment in SF. Table I. ADC Conversion and Settling Times for Various SF Words SF Data Update Rate Settling Time Word fADC (Hz) tSETTLE (ms) 13 105.3 19.04 69 (Default) 19.79 101.07 255 5.35 373.54 Normal mode rejection is the major function of the digital filter on the AD7719. The normal mode 50 Hz ± 1 Hz rejection with an SF word of 82 is typically –100 dB. The 60 Hz ± 1 Hz rejection with SF = 68 is typically –100 dB. Simultaneous 50 Hz and 60 Hz rejection of better than 60 dB is achieved with an SF of 69. Choosing an SF word of 69 places notches at both 50 Hz and 60 Hz. Figures 6 to 9 show the filter rejection for a selection of SF words. REV. A AD7719 –14– MAIN AND AUXILIARY ADC NOISE PERFORMANCE Tables II to VII show the output rms noise and output peak-topeak resolution in bits (rounded to the nearest 0.5 LSB) for a selection of output update rates on both the main and auxiliary ADCs. The numbers are typical and are generated at a differential input voltage of 0 V. The output update rate is selected via the SF7 to SF0 bits in the Filter register. It is important to note that the peak-to-peak resolution figures represent the resolution for which there will be no code flicker within a six-sigma limit. The output noise comes from two sources. The first is the electrical noise in the semiconductor devices (device noise) used in the implementation of the modulator. Secondly, when the analog input is converted into the digital domain, quantization noise is added. The device noise is at a low level and is independent of frequency. The quantization noise starts at an even lower level but rises rapidly with increasing frequency to become the dominant noise source. The numbers in the tables are given for the bipolar input ranges. For the unipolar ranges, the rms noise numbers will be the same as the bipolar range, but the peak-to-peak resolution is now based on half the signal range, which effectively means losing one bit of resolution. FREQUENCY (Hz) 0 –140 –200 0 50 100 150 200 250 300 350 400 450 500 550 600 650 700 ATTENUATION (dB) –20 –120 –160 –180 –60 –100 –40 –80 SF = 13 OUTPUT DATA RATE = 105Hz INPUT BANDWIDTH = 25.2Hz FIRST NOTCH = 52.5Hz 50Hz REJECTION = –23.6dB, 50Hz 1Hz REJECTION = –20.5dB 60Hz REJECTION = –14.6dB, 60Hz 1Hz REJECTION = –13.6dB Figure 6. Filter Profile with SF = 13 FREQUENCY (Hz) 0 –80 –160 0 10 100 ATTENUATION (dB) 20 30 40 50 60 70 80 90 –20 –40 –120 –140 –60 –100 SF = 82 OUTPUT DATA RATE = 16.65Hz INPUT BANDWIDTH = 4Hz 50Hz REJECTION = –171dB, 50Hz 1Hz REJECTION = –100dB 60Hz REJECTION = –58dB, 60Hz 1Hz REJECTION = –53dB Figure 7. Filter Profile with SF = 82 FREQUENCY (Hz) 0 –80 –160 0 10 100 ATTENUATION (dB) 20 30 40 50 60 70 80 90 –20 –40 –120 –140 –60 –100 SF = 69 OUTPUT DATA RATE = 19.8Hz INPUT BANDWIDTH = 4.74Hz FIRST NOTCH = 9.9Hz 50Hz REJECTION = –66dB, 50Hz 1Hz REJECTION = –60dB 60Hz REJECTION = –117dB, 60Hz 1Hz REJECTION = –94dB Figure 8. Filter Profile with Default SF = 69 Giving Filter Notches at Both 50 Hz and 60 Hz FREQUENCY (Hz) 0 –80 –160 0 10 100 ATTENUATION (dB) 20 30 40 50 60 70 80 90 –20 –40 –120 –140 –60 –100 SF = 255 OUTPUT DATA RATE = 5.35Hz INPUT BANDWIDTH = 1.28Hz 50Hz REJECTION = –93dB, 50Hz 1Hz REJECTION = –93dB 60Hz REJECTION = –74dB, 60Hz 1Hz REJECTION = –68dB Figure 9. Filter Profile with SF = 255 REV. A AD7719 –15– Table II. Typical Output RMS Noise vs. Input Range and Update Rate for Main ADC (Buffered Mode) Output RMS Noise in V SF Data Update Input Range Word Rate (Hz) 20 mV 40 mV 80 mV 160 mV 320 mV 640 mV 1.28 V 2.56 V 13 105.3 1.50 1.50 1.60 1.75 3.50 4.50 6.70 11.75 69 19.79 0.60 0.65 0.65 0.65 0.65 0.95 1.40 2.30 255 5.35 0.35 0.35 0.37 0.37 0.37 0.51 0.82 1.25 Table III. Peak-to-Peak Resolution vs. Input Range and Update Rate for Main ADC (Buffered Mode) Peak-to-Peak Resolution in Bits SF Data Update Input Range Word Rate (Hz) 20 mV 40 mV 80 mV 160 mV 320 mV 640 mV 1.28 V 2.56 V 13 105.3 12 13 14 15 15 15.5 16 16 69 19.79 13 14 15 16 17 17.5 18 18.5 255 5.35 14 15 16 17 18 18.5 18.8 19.2 Table IV. Typical Output RMS Noise vs. Input Range and Update Rate for Main ADC (Unbuffered Mode) Output RMS Noise in V SF Data Update Input Range Word Rate (Hz) 20 mV 40 mV 80 mV 160 mV 320 mV 640 mV 1.28 V 2.56 V 13 105.3 1.27 1.27 1.35 1.48 2.95 3.82 5.69 10.2 69 19.79 0.52 0.56 0.56 0.56 0.56 0.82 1.21 2.00 255 5.35 0.30 0.30 0.32 0.32 0.32 0.44 0.71 1.10 Table V. Peak-to-Peak Resolution vs. Input Range and Update Rate for Main ADC (Unbuffered Mode) Peak-to-Peak Resolution in Bits SF Data Update Input Range Word Rate (Hz) 20 mV 40 mV 80 mV 160 mV 320 mV 640 mV 1.28 V 2.56 V 13 105.3 12 13 14 15 15 15.5 16 16 69 19.79 13 14 15 16 17 17.5 18 18.5 255 5.35 14 15 16 17 18 18.5 19 19.5 Table VI. Typical Output RMS Noise vs. Update Rate for Auxiliary ADC (Unbuffered Mode) SF Data Update Input Range Word Rate (Hz) 2.5 V 13 105.3 10.75 μV 69 19.79 2.00 μV 255 5.35 1.15 μV Table VII. Peak-to-Peak Resolution vs. Update Rate for Auxiliary ADC (Unbuffered Mode) SF Data Update Input Range Word Rate (Hz) 2.5 V 13 105.3 16 Bits 69 19.79 16 Bits 255 5.35 16 Bits REV. A AD7719 –16– DOUT FILTER REGISTER DIN DOUT TEST REGISTER DIN DOUT ID REGISTER DOUT AUX ADC GAIN REGISTER DIN ADC STATUS REGISTER DOUT DOUT MODE REGISTER DIN DOUT MAIN ADC CONTROL REGISTER DIN DOUT AUX ADC CONTROL REGISTER DIN DOUT I/O CONTROL REGISTER DIN DOUT MAIN ADC GAIN REGISTER DIN DOUT AUX ADC OFFSET REGISTER DIN DOUT MAIN ADC OFFSET REGISTER DIN DOUT AUX ADC DATA REGISTER DOUT MAIN ADC DATA REGISTER WEN R/W 0 0 A3 A2 A1 A0 COMMUNICATIONS REGISTER DOUT DIN REGISTER SELECT DECODER Figure 10. On-Chip Registers ON-CHIP REGISTERS Both the main and auxiliary ADC channels are controlled and configured via a number of on-chip registers as shown in Figure 10 and described in more detail in the following pages. In the following descriptions, SET implies a logic 1 state and CLEARED implies a logic 0 state, unless otherwise stated. REV. A AD7719 –17– Table VIII. Registers—Quick Reference Guide Power-On/Reset Register Name Type Size Default Value Function Communications Write Only 8 Bits Not Applicable All operations to other registers are initiated through the Communications register. This controls whether subsequent operations are read or write operations and also selects the register for that subsequent operation. Status Register Read Only 8 Bits 0x00 Provides status information on conversions, calibrations, error conditions, and the validity of the reference voltage. Mode Register Read/Write 8 Bits 0x00 Controls functions such as mode of operation, channel configuration, and oscillator operation in power-down. Main ADC (AD0CON) Control Register Read/Write 8 Bits 0x07 This register is used to enable the main ADC and to configure the main ADC for range, channel selection, 16-/24-bit operation, and unipolar or bipolar operation. Aux ADC (AD1CON) Control Register Read/Write 8 Bits 0x01 This register is used to enable the aux ADC and to configure the Aux ADC for range, channel selection, unipolar or bipolar operation, and input range. I/O (IOCON) Control Register Read/Write 16 Bits 0x0000 This register is used to control and configure the various excitation and burnout current source options available on-chip along with controlling the I/O port. Filter Register Read/Write 8 Bits 0x45 This register determines the amount of averaging performed by the sinc filter and consequently determines the data update rate of the AD7719. The filter register determines the update rate for both the main and aux ADCs. MSB LSB RDY0 RDY1 CAL NOREF ERR0 ERR1 0 LOCK MSB LSB SF7 SF6 SF5 SF4 SF3 SF2 SF1 SF0 MSB PSW2 PSW1 0 BO I2PIN I1PIN I2EN I1EN LSB P4DIR P3DIR P2EN P1EN P4DAT P3DAT P2DAT P1DAT MSB LSB WEN R/W 0 0 A3 A2 A1 A0 MSB LSB 0 BUF 0 CHCON OSCPD MD2 MD1 MD0 MSB LSB AD0EN WL CH1 CH0 U/B RN2 RN1 RN0 MSB LSB AD1EN ACH2 ACH1 ACH0 U/B 0 0 ARN REV. A AD7719 –18– Power-On/Reset Register Name Type Size Default Value Function Main ADC (DATA0) Data Register Read Only 16 Bits or 24 Bits 0x00 0000 Provides the most up-to-date conversion result from the main ADC. Main ADC data register length can be programmed to be 16-bit or 24-bit. Aux ADC (DATA1) Data Register Read Only 16 Bits 0x0000 Provides the most up-to-date conversion result from the auxiliary ADC. Aux ADC data register length is 16 bits. Main ADC Offset Register Read/Write 24 Bits 0x80 0000 Contains a 24-bit word that is the offset calibration coefficient for the part. The contents of this register are used to provide offset correction on the output from the digital filter. There are three offset registers on the part and these are associated with input channel pairs as outlined in the AD0CON register. Main ADC Gain Register Read/Write 24 Bits 0x5X XXX5 Contains a 24-bit word that is the gain calibration coefficient for the part. The contents of this register are used to provide gain correction on the output from the digital filter. There are three Gain registers on the part, which are associated with input channel pairs as outlined in the AD0CON register. Aux ADC Offset Register Read/Write 16 Bits 0x8000 Contains a 16-bit word that is the offset calibration coefficient for the part. The contents of this register are used to provide offset correction on the output from the digital filter. Aux ADC Gain Register Read/Write 24 Bits 0x59XX Contains a 16-bit word that is the gain calibration coefficient for the part. The contents of this register are used to provide gain correction on the output from the digital filter. ID Register Read 8 Bits 0x0X Contains an 8-bit byte that is the identifier for the part. Test Registers Read/Write 16 Bits 0x0000 Controls the test modes of the part, which are used when testing the part. The user is advised not to change the contents of these registers. REV. A AD7719 –19– CR7 CR6 CR5 CR4 CR3 CR2 CR1 CR0 WEN (0) R/W (0) 0 (0) 0 (0) A3 (0) A2 (0) A1 (0) A0 (0) Table IX. Communications Register Bit Designations Bit Bit Location Name Description CR7 WEN Write Enable Bit. A 0 must be written to this bit so the write operation to the Communications register actually takes place. If a 1 is written to this bit, the part will not clock on to subsequent bits in the register. It will stay at this bit location until a 0 is written to this bit. Once a 0 is written to the WEN bit, the next seven bits will be loaded to the Communications register. CR6 R/W A 0 in this bit location indicates that the next operation will be a write to a specified register. A 1 in this position indicates that the next operation will be a read from the designated register. CR5 Zero A 0 must be written to this bit position to ensure correct operation of the AD7719. CR4 Zero A 0 must be written to this bit position to ensure correct operation of the AD7719. CR3–CR0 A3–A0 Register Address Bits. These address bits are used to select which of the AD7719’s registers is being accessed during this serial interface communication. A3 is the MSB of the three selection bits. Communications Register (A3, A2, A1, A0 = 0, 0, 2, 0) The Communications register is an 8-bit write-only register. All communications to the part must start with a write operation to the Communications register. The data written to the Communications register determines whether the next operation is a read or write operation, and to which register this operation takes place. For read or write operations, once the subsequent read or write operation to the selected register is complete, the interface returns to where it expects a write operation to the Communications register. This is the default state of the interface, and on power-up or after a RESET, the AD7719 is in this default state waiting for a write operation to the Communications register. In situations where the interface sequence is lost, a write operation of at least 32 serial clock cycles with DIN high returns the AD7719 to this default state by resetting the part. Table IX outlines the bit designations for the Communications register. CR0 through CR7 indicate the bit location, with CR denoting that the bits are in the Communications register. CR7 denotes the first bit of the data stream. Table X. Register Selection Table A3 A2 A1 A0 Register 0 0 0 0 Communications Register during a Write Operation 0 0 0 0 Status Register during a Read Operation 0 0 0 1 Mode Register 0 0 1 0 Main ADC Control Register (AD0CON) 0 0 1 1 Aux ADC Control Register (AD1CON) 0 1 0 0 Filter Register 0 1 0 1 Main ADC Data Register 0 1 1 0 Aux ADC Data Register 0 1 1 1 I/O Control Register 1 0 0 0 Main ADC Offset Calibration Register 1 0 0 1 Aux ADC Offset Calibration Register 1 0 1 0 Main ADC Gain Calibration Register 1 0 1 1 Aux ADC Gain Calibration Register 1 1 0 0 Test 1 Register 1 1 0 1 Test 2 Register 1 1 1 0 Undefined 1 1 1 1 ID Register REV. A AD7719 –20– Table XI. Status Register Bit Designations Bit Bit Location Name Description SR7 RDY0 Ready Bit for Main ADC. Set when data is written to main ADC data registers or on completion of calibration cycle. The RDY0 bit is cleared automatically after the main ADC data register has been read or after a period of time before the data register is updated with a new conversion result. This bit is also cleared by a write to the mode bits to indicate a conversion or calibration. SR6 RDY1 Ready Bit for Aux ADC. Set when data is written to aux ADC data registers or on completion of calibration cycle. The RDY1 bit is cleared automatically after the aux ADC data register has been read or a period of time before the data register is updated with a new conversion result. This bit is also cleared by a write to the mode bits to indicate a conversion or calibration. SR5 CAL Calibration Status Bit. Set to indicate completion of calibration. It is set at the same time that the RDY0 and/or RDY1 bits are set high. Cleared by a write to the mode bits to start another ADC conversion or calibration. SR4 NOXREF No External Reference Bit. (Only active if main ADC is active and applies to REFIN1 only.) Set to indicate that one or both of the REFIN1 pins is floating or the applied voltage is below a specified threshold. When Set, conversion results are clamped to all 1s. Cleared to indicate valid reference applied between REFIN1(+) and REFIN1(–). SR3 ERR0 Main ADC Error Bit. Set to indicate that the result written to the main ADC data registers has been clamped to all 0s or all 1s. After a calibration, this bit also flags error conditions that caused the calibration registers not to be written. Error sources include Overrange, Underrange, and NOXREF. Cleared by a write to the mode bits to initiate a conversion or calibration. SR2 ERR1 Aux ADC Error Bit. Set to indicate that the result written to the Aux ADC data registers has been clamped to all 0s or all 1s. After a calibration, this bit also flags error conditions that caused the calibration registers not to be written. Error sources include Overrange, Underrange, and NOXREF. Cleared by a write to the mode bits to initiate a conversion or calibration. SR1 0 Reserved for Future Use. SR0 LOCK PLL Lock Status Bit. Set if the PLL has locked onto the 32 kHz crystal oscillator clock. If the user is worried about exact sampling frequencies, for example, the LOCK bit should be interrogated and the result discarded if the LOCK bit is 0. Status Register (A3, A2, A1, A0 = 0, 0, 0, 0; Power-On Reset = 0x00) The ADC Status register is an 8-bit read-only register. To access the ADC Status register, the user must write to the Communications register selecting the next operation to be a read and loading bits A3 to A0 with 0, 0, 0, 0. Table XI outlines the bit designations for the Status register. SR0 through SR7 indicate the bit location, with SR denoting that the bits are in the Status register. SR7 denotes the first bit of the data stream. The number in parentheses indicates the power-on/reset default status of that bit. SR7 SR6 SR5 SR4 SR3 SR2 SR1 SR0 RDY0 (0) RDY1 (0) CAL (0) NOXREF (0) ERR0 (0) ERR1 (0) (0) LOCK (0) REV. A AD7719 –21– Mode Register (A3, A2, A1, A0 = 0, 0, 0, 1; Power-On Reset = 0x00) The Mode register is an 8-bit register from which data can be read or to which data can be written. This register configures the operating modes of the AD7719. Table XII outlines the bit designations for the Mode register. MR7 through MR0 indicate the bit location, with MR denoting the bits are in the Mode register. MR7 denotes the first bit of the data stream. The number in parentheses indicates the power-on/reset default status of that bit. Table XII. MODE Register Bit Designations Bit Bit Location Name Description MR7 0 Reserved for Future Use. MR6 BUF Configures the main ADC for buffered or unbuffered mode of operation. If set, the main ADC operates in unbuffered mode, lowering the power consumption of the AD7719. If cleared, the Main ADC operates in buffered mode, allowing the user to place source impedances on the front end without contributing gain errors to the system. MR5 0 Reserved for Future Use. MR4 CHCON Channel Configure Bit. If this bit is set, the main ADC operates with three pseudodifferential input channels and the aux ADC does not have AIN3/AIN4 as an input option. If cleared, the main ADC operates with two fully differential input channels and the aux channel operates as one fully differential input and two single-ended inputs or as three single-ended inputs. MR3 OSCPD Oscillator Power-Down Bit. If this bit is set, placing the AD7719 in standby mode will stop the crystal oscillator, reducing the power drawn by the AD7719 to a minimum. The oscillator will require 300 ms to begin oscillating when the ADC is taken out of standby mode. If this bit is cleared, the oscillator is not shut off when the ADC is put into standby mode and will not require the 300 ms start-up time when the ADC is taken out of standby. MR2–MR0 MD2–MD0 Main and Aux ADC Mode Bits. These bits select the operational mode of the enabled ADC as follows: MD2 MD1 MD0 Description 0 0 0 Power-Down Mode (Power-On Default). The current sources, power switches, and PLL are shut off in Power-Down mode. 0 0 1 Idle Mode. In Idle mode, the ADC filter and modulator are held in a reset state although the modulator clocks are still provided. 0 1 0 Single Conversion Mode. In Single Conversion mode, a single conversion is performed on the enabled channels. On completion of the conversion, the ADC data registers are updated, the relevant flags in the STATUS register are written, and idle mode is entered with the MD2–MD0 being written accordingly to 001. 0 1 1 Continuous Conversion. In continuous conversion mode, the ADC data registers are regularly updated at the selected update rate (see Filter register). 1 0 0 Internal Zero-Scale Calibration. Internal short automatically connected to the enabled channel(s). Returns to Idle mode (001) when complete. 1 0 1 Internal Full-Scale Calibration. External VREF is connected automatically to the ADC input for this calibration. Returns to idle mode when complete. 1 1 0 System Zero-Scale Calibration. User should connect system zero-scale input to the channel input pins as selected by the CH1/CH0 and ACH1/ACH0 bits in the control registers. 1 1 1 System Full-Scale Calibration. User should connect system full-scale input to the channel input pins as selected by the CH1/CH0 and ACH1/ACH0 bits in the control registers. MR7 MR6 MR5 MR4 MR3 MR2 MR1 MR0 0 (0) BUF (0) 0 (0) CHCON (0) OSCPD (0) MD2 (0) MD1 (0) MD0 (0) REV. A AD7719 –22– Operating Characteristics when Addressing the Mode and Control Registers 1. Any change to the MD bits will immediately reset both ADCs. A write to the MD2–0 bits with no change is also treated as a reset. (See exception to this in Note 3.) 2. If AD0CON is written when AD0EN = 1, or if AD0EN is changed from 0 to 1, both ADCs are also immediately reset. In other words, the main ADC is given priority over the aux ADC and any change requested on main is immediately responded to. 3. On the other hand, if AD1CON is written to, only the aux ADC is reset. For example, if the main ADC is continuously converting when the aux ADC change or enable occurs, the main ADC continues undisturbed. Rather than allow the aux ADC to operate with a phase difference from the main ADC, the aux ADC will fall into step with the outputs of the main ADC. The result is that the first conversion time for the aux channel will be delayed up to three outputs while the aux ADC update rate is synchronized to the main ADC. 4. Once the MODE has been written with a calibration mode, the RDY0/1 bits (STATUS) are immediately reset and the calibration commences. On completion, the appropriate calibration registers are written, the relevant bits in STATUS are written, and the MD2–0 bits are reset to 001 to indicate the ADC is back in Idle mode. 5. Any calibration request of the aux ADC while the temperature sensor is selected will fail to complete. 6. Calibrations are performed with the maximum allowable SF value. SF register is reset to user configuration after calibration. Main ADC Control Register (AD0CON): (A3, A2, A1, A0 = 0, 0, 1, 0; Power-On Reset = 0x07) The main ADC control register is an 8-bit register from which data can be read or to which data can be written. This register is used to configure the main ADC for range, channel selection, 16-/24-bit operation, and unipolar or bipolar coding. Table XIII outlines the bit designations for the main ADC control register. AD0CON7 through AD0CON0 indicate the bit location, AD0CON denoting the bits are in the main ADC control register. AD0CON7 denotes the first bit of the data stream. The number in parentheses indicates the power-on/reset default status of that bit. Table XIII. Main ADC Control Register (AD0CON) Bit Designations Bit Location Bit Name Description AD0CON7 AD0EN Main ADC Enable Bit. Set by user to enable the main ADC. When set, the main ADC operates according to the MD bits in the mode register. Cleared by the user to power down the Main ADC. AD0CON6 WL 16-/24-Bit Operating Mode. Set by user to enable 16-bit mode. The conversion results from the main ADC will be rounded to 16 bits and the main ADC data register will be 16 bits wide. Cleared by user to enable 24-bit mode. The conversion results from the main ADC will be rounded to 24 bits and the main ADC data register will be 24 bits wide. AD0CON5 CH1 Main ADC Channel Selection Bits. AD0CON4 CH0 Written by the user to select the differential input pairs used by the main ADC as follows: (Note: The CHCON bit resides in the Mode register.) CHCON CH1 CH0 Positive Input Negative Input Calibration Register Pair 0 0 0 AIN1 AIN2 0 0 0 1 AIN3 AIN4 1 0 1 0 AIN2 AIN2 0 0 1 1 AIN3 AIN2 1 1 0 0 AIN1 AIN4 0 1 0 1 AIN3 AIN4 1 1 1 0 AIN4 AIN4 0 1 1 1 AIN2 AIN4 2 AD0CON3 U/B Main ADC Unipolar/Bipolar Bit. Set by user to enable unipolar coding, i.e., zero differential input will result in 0x00 0000 output and a full-scale differential input will result in 0xFF FFFF output when operated in 24-bit mode. Cleared by user to enable bipolar coding, Negative full-scale differential input will result in an output code of 0x00 0000, zero differential input will result in an output code of 0x80 0000, and a Positive full-scale differential input will result in an output code of 0xFF FFFF. AD0CON7 AD0CON6 AD0CON5 AD0CON4 AD0CON3 AD0CON2 AD0CON1 AD0CON0 AD0EN (0) WL (0) CH1 (0) CH0 (0) U/B (0) RN2 (1) RN1 (1) RN0 (1) REV. A AD7719 –23– Table XIII. Main ADC Control Register (AD0CON) Bit Designations (continued) Bit Location Bit Name Description AD0CON2 RN2 Main ADC Range Bits. AD0CON1 RN1 Written by the user to select the main ADC input range as follows. AD0CON0 RN0 RN2 RN1 RN0 Selected Main ADC Input Range (VREF = 2.5 V) 0 0 0 ±20 mV 0 0 1 ±40 mV 0 1 0 ±80 mV 0 1 1 ±160 mV 1 0 0 ±320 mV 1 0 1 ±640 mV 1 1 0 ±1.28 V 1 1 1 ±2.56 V Aux ADC Control Registers (AD1CON): (A3, A2, A1, A0 = 0, 0, 1, 1; Power-On Reset = 0x00) The aux ADC control register is an 8-bit register from which data can be read or to which data can be written. This register is used to configure the aux ADC for range, channel selection, and unipolar or bipolar coding. Table XIV outlines the bit designations for the aux ADC control register. AD1CON7 through AD1CON0 indicate the bit location, with AD1CON denoting that the bits are in the aux ADC control register. AD1CON7 denotes the first bit of the data stream. The number in parentheses indicates the power-on/reset default status of that bit. Table XIV. Aux ADC Control Register (AD1CON) Bit Designations Bit Location Bit Name Description AD1CON7 AD1EN Aux ADC Enable Bit. Set by user to enable the Aux ADC. When set, the aux ADC operates according to the MD bits in the mode register. Cleared by the user to power down the aux ADC. AD1CON6 ACH2 Aux ADC Channel Selection Bits. AD1CON5 ACH1 Written by the user to select the active input channels used by the aux ADC as follows: AD1CON4 ACH0 CHCON ACH2 ACH1 ACH0 Positive Input Negative Input 0 0 0 0 AIN3 AGND 0 0 0 1 AIN4 AGND 0 0 1 0 AIN5 AIN6 0 0 1 1 Temp Sensor (Temp Sensor Routed to the ADC Inputs) 0 1 0 0 AGND AGND (Internal Short) 1 0 0 0 AIN5 AGND 1 0 0 1 AIN6 AGND 1 0 1 0 AIN5 AIN6 1 0 1 1 Temp Sensor (Temp Sensor Routed to the ADC Inputs) 1 1 0 0 AGND AGND (Internal Short) X 1 0 1 Not Defined X 1 1 0 Not Defined X 1 1 1 Not Defined AD1CON3 U/B Aux ADC Unipolar/Bipolar Selection Bit. Set by user to enable unipolar coding, i.e., zero differential input will result in 0x0000 output. Cleared by user to enable bipolar coding, zero differential input will result in 0x8000 output. AD1CON2 0 Must be zero for specified operation. AD1CON1 0 Must be zero for specified operation. AD1CON0 ARN Auxiliary Channel Input Range Bit. When set by the user, the input range is ±REFIN2. When cleared by the user, the input range is ±REFIN2/2. NOTES 1. When the temperature sensor is selected, the AD7719 automatically selects its internal reference. The temperature sensor is not factory calibrated. Temp sensor is suitable for relative temperature measurements. The temperature sensor yields conversion results where a conversion result of 0x8000 equates to typically 0°C. 2. A 1°C change in temperature will normally result in a 256 LSB change in the AD1 data register (ADC conversion result). AD1CON7 AD1CON6 AD1CON5 AD1CON4 AD1CON3 AD1CON2 AD1CON1 AD1CON0 AD1EN (0) ACH2 (0) ACH1 (0) ACH0 (0) U/B (0) 0 (0) 0 (0) ARN (1) REV. A AD7719 –24– FR7 FR6 FR5 FR4 FR3 FR2 FR1 FR0 SF7 (0) SF6 (1) SF5 (0) SF4 (0) SF3 (0) SF2 (1) SF1 (0) SF0 (1) IOCON15 IOCON14 IOCON13 IOCON12 IOCON11 IOCON10 IOCON9 IOCON8 PSW2 (0) PSW1 (0) 0 (0) BO (0) I2PIN (0) I1PIN (1) I2EN (0) I1EN (0) IOCON7 IOCON6 IOCON5 IOCON4 IOCON3 IOCON2 IOCON1 IOCON0 P4DIR (0) P3DIR (0) P2EN (0) P1EN (0) P4DAT (0) P3DAT (0) P2DAT (0) P1DAT (0) Filter Register (A3, A2, A1, A0 = 0, 1, 0, 0; Power-On Reset = 0x45) The Filter register is an 8-bit register from which data can be read or to which data can be written. This register determines the amount of averaging performed by the sinc filter. Table XV outlines the bit designations for the Filter register. FR7 through FR0 indicate the bit location, with FR denoting that the bits are in the Filter register. FR7 denotes the first bit of the data stream. The number in parentheses indicates the power-on/reset default status of that bit. The number in this register is used to set the decimation factor and thus the output update rate for the main and aux ADCs. The filter register cannot be written to by the user while either ADC is active. The update rate is used for both main and aux ADCs and is calculated as follows: f SF ADC = × fMOD × × 1 3 1 8 where: fADC = ADC output update rate fMOD = Modulator clock frequency = 32.768 kHz (main and aux ADC) SF = Decimal value written to SF register The allowable range for SF is 13dec to 255dec. Examples of SF values and corresponding conversion rate (fADC) and time (tADC) are shown in Table XV. It should also be noted that both ADC input channels are chopped to minimize offset errors. This means that the time for a single conversion or the time to the first conversion result is 2 × tADC. Table XV. Update Rate vs. SF WORD SF (dec) SF (Hex) fADC (Hz) tADC (ms) 13 0D 105.3 9.52 69 45 19.79 50.34 255 FF 5.35 186.77 I/O and Current Source Control Register (IOCON): (A3, A2, A1, A0 = 0, 1, 1, 1; Power-On Reset = 0x0000) The IOCON register is a 16-bit register from which data can be read or to which data can be written. This register is used to control and configure the various excitation and burnout current source options available on-chip along with controlling the I/O port. Table XVI outlines the bit designations for this register. IOCON15 through IOCON0 indicate the bit location, with IOCON denoting that the bits are in the I/O and Current Source control register. IOCON15 denotes the first bit of the data stream. The number in parentheses indicates the power-on/reset default status of that bit. A write to the IOCON register has immediate effect and does not reset the ADCs. Thus if a current source is switched while the ADC is converting, the user will have to wait for the full settling time of the filter before getting a fully settled output. Since the ADC is chopped, this equates to three outputs. REV. A AD7719 –25– Table XVI. IOCON (I/O and Current Source Control Register) Bit Designations Bit Bit Location Name Description IOCON15 PSW2 Power Switch 2 Control Bit. Set by user to enable power switch P2 to PWRGND. Cleared by user to enable use as a standard I/O pin. When the ADC is in standby mode, the power switches are open. IOCON14 PSW1 Power Switch 1 Control Bit. Set by user to enable power switch P1 to PWRGND. Cleared by user to enable use as a standard I/O pin. When ADC is in standby mode, the power switches are open. IOCON13 0 This bit must be zero for correct operation. IOCON12 BO Burnout Current Enable Bit. Set by user to enable the 100 nA current sources in the main ADC signal path. A 100 nA current source is applied to the positive input leg while a 100 nA sink is applied to the negative input. Cleared by user to disable both transducer burnout current sources. IOCON11 I2PIN IEXE2, 200 μA Current Source Direction Bit. Set by user to enable IEXC2 current source to IOUT1. Cleared by user to enable IEXC2 current source to IOUT2. IOCON10 I1PIN IEXE1, 200 μA Current Source Direction Bit. Set by user to enable IEXC1 current source to IOUT2. Cleared by user to enable IEXC1 current source to IOUT1. IOCON9 I2EN IEXC2 Current Source Enable Bit. Set by user to turn on the IEXC2 excitation current source. Cleared by user to turn off the IEXC2 excitation current source. IOCON8 I1EN IEXC1 Current Source Enable Bit. Set by user to turn on the IEXC1 excitation current source. Cleared by user to turn off the IEXC1 excitation current source. IOCON7 P4DIR P4, I/O Direction Control Bit. Set by user to enable P4 as an output. Cleared by user to enable P4 as an input. There are weak active pull-ups internally when enabled as an input. IOCON6 P3DIR P3, I/O Direction Control Bit. Set by user to enable P3 as an output. Cleared by user to enable P3 as an input. There are weak active pull-ups internally when enabled as an input. IOCON5 P2EN P2 Digital Output Enable Bit. Set by user to enable P2 as a regular digital output pin. Cleared by user to three-state P2 output. PSW2 takes precedence over P2EN. IOCON4 P1EN P1 Digital Output Enable Bit. Set by user to enable P1 as a regular digital output pin. Cleared by user to three-state P1 output. PSW1 takes precedence over P1EN. IOCON3 P4DAT Digital I/O Port Data Bits. IOCON2 P3DAT The readback values of these bits indicate the status of their respective pin when the I/O port is active as IOCON1 P2DAT an input. IOCON0 P1DAT The values written to these data bits appear at the output port when the I/O bits are enabled as outputs. P2 and P1 are outputs only, so reading P2DAT and P1DAT will return what was last written to these bits. REV. A AD7719 –26– Main ADC Data Result Registers (DATA0): (A3, A2, A1, A0 = 0, 1, 0, 1; Power-On Reset = 0x00 0000) The conversion results for the main ADC channel are stored in the main ADC data register (DATA0). This register is either 16 or 24 bits wide, depending on the status of the WL bit in the main ADC control register (AD0CON). This is a read-only register. On completion of a read from this register, the RDY0 bit in the status register is cleared. Aux ADC Data Result Registers (DATA1): (A3, A2, A1, A0 = 0, 1, 1, 0; Power-On Reset = 0x0000) The conversion results for the aux ADC channel are stored in the aux ADC data register (DATA1). This register is 16 bits wide and is a read-only register. On completion of a read from this register, the RDY1 bit in the status register is cleared. Main ADC Offset Calibration Coefficient Registers (OF0): (A3, A2, A1, A0 = 1, 0, 0, 0; Power-On Reset = 0x80 0000) The offset calibration registers hold the 24-bit data offset calibration coefficient for the main ADC. There are three registers associated with the main ADC channel. In fully differential operating mode, there are two input channels and a register is dedicated to each input. When operating in pseudodifferential mode, the main ADC can be configured for three input channels and there is a dedicated register for each pseudodifferential input. These registers have a power-on reset value of 0x80 0000. The channel bits, in association with the communication register address for the OF0 register, allow access to these registers. These registers are read/write registers. The calibration registers can only be written to if the ADC is inactive (MD bits in the mode register = 000 or 001 or both AD0EN and AD1EN bits in the control registers are cleared). Reading of the calibration registers does not clear the RDY0 bit. Aux ADC Offset Calibration Coefficient Registers (OF1): (A3, A2, A1, A0 = 1, 0, 0, 1; Power-On Reset = 0x8000) The offset calibration register OF1 holds the 16-bit data offset calibration coefficient for the aux ADC. This register has a poweron- reset value of 0x8000. The channel bits, in association with the communication register address for the OF1 register, allow access to these registers. These registers are read/write registers. The calibration registers can only be written to if the ADC is inactive (MD bits in the mode register = 000 or 001 or both AD0EN and AD1EN bits in the control registers are cleared). Reading of the calibration registers does not clear the RDY1 bit. Main ADC Gain Calibration Coefficient Registers (GNO): (A3, A2, A1, A0 = 1, 0, 1, 0; Power-On Reset = 0x5X XXX5) The gain calibration registers hold the 24-bit data gain calibration coefficient for the main ADC. These registers are configured at power-on with factory calculated internal full-scale calibration coefficients. Every device will have different coefficients. However, these bytes will be automatically overwritten if an internal or system full-scale calibration is initiated by the user via MD2–0 bits in the Mode register. There are three gain calibration registers associated with the main ADC channel. In fully differential operating mode, there are two input channels and a register is dedicated to each input. When operating in pseudodifferential mode, the main ADC can be configured for three input channels and there is a dedicated register for each pseudodifferential input. These registers are read/write registers. The calibration registers can only be written to if the ADC is inactive (MD bits in the mode register = 000 or 001 or both AD0EN and AD1EN bits in the control registers are cleared). Reading of the calibration registers does not clear the RDY1 bit. Aux ADC Gain Calibration Coefficient Registers (GN1): (A3, A2, A1, A0 = 1, 0, 1, 1; Power-On Reset = 0x59XX) The gain calibration register GN1 holds the 16-bit data gain calibration coefficient for the aux ADC. This register is configured at power-on with factory calculated internal zero-scale calibration coefficients. Every device will have different coefficients. However, these coefficients will be automatically overwritten if an internal or system zero-scale calibration is initiated by the user via the MD2–0 bits in the Mode register. These registers are read/write registers. The calibration registers can only be written to if the ADC is inactive (MD bits in the mode register = 000 or 001 or both AD0EN and AD1EN bits in the control registers are cleared). Reading of the calibration registers does not clear the RDY1 bit. ID Register (ID): (A3, A2, A1, A0 = 1, 1, 1, 1; Power-On Reset = 0x0X) This register is a read-only 8-bit register. The contents are used to determine the die revision of the AD7719. Table XVII indicates the bit locations. User Nonprogrammable Test Registers The AD7719 contains two test registers. The bits in this test register control the test modes of the AD7719, which are used for the testing of the device. The user is advised not to change the contents of these registers. Table XVII. ID Register Bit Designations ID7 ID6 ID5 ID4 ID3 ID2 ID1 ID0 0 0 0 0 X X X X REV. A AD7719 –27– CONFIGURING THE AD7719 All user-accessible registers on the AD7719 are accessed via the serial interface. Communication with any of these registers is initiated by first writing to the Communications register. Figure 11 outlines a flow diagram of the sequence used to configure all registers after a power-up or reset on the AD7719. The flowchart shows two methods of determining when it is valid to read the data register or determine when a calibration cycle is complete. The first method is hardware polling of the RDY pin and the second method involves software interrogation of bits in the status and mode registers. The flowchart details all the necessary programming steps required to initialize the ADC and read data from the main and aux channel following a power-on or reset. The steps can be broken down as follows: 1. Configure and initialize the microcontroller or microprocessor serial port. 2. Initialize the AD7719 by configuring the following registers: a) IOCON to configure the current sources and digital I/O port. b) FILTER to configure the update rate for both channels. c) AD1CON to enable the aux channel, select the analog input, select unipolar or bipolar operation and input range. d) AD0CON to enable the main ADC channel and select 16-/24-bit mode, analog input range, and either unipolar or bipolar operation. e) MODE to configure the operating mode. Operating mode consists of calibration or conversion. All of these operations consist of a write to the communications register to specify the next operation as a write to a specified register. Data is then written to this register. When each sequence is complete, the ADC defaults to waiting for another write to the Communications register to specify the next operation. 3. When the operating mode is selected, the user needs to determine when it is valid to read the data in conversion mode or when the calibration is complete in calibration mode. This is accomplished either by polling the RDY pin (hardware polling) or by interrogating the bits in either the Status or Mode registers (software polling). Both are shown in Figure 11. It is assumed that both the main and aux ADCs are being used and calibration is required. If the AD7719 is operated at the factory-calibrated conditions, a field calibration will not be required and these steps can be bypassed. MICROCOMPUTER/MICROPROCESSOR INTERFACING The AD7719’s flexible serial interface allows for easy interface to most microcomputers and microprocessors. The flowchart of Figure 11 outlines the sequence that should be followed when interfacing a microcontroller or microprocessor to the AD7719. Figures 12, 13, and 14 show some typical interface circuits. The serial interface on the AD7719 is capable of operating from just three wires and is compatible with SPI interface protocols. The 3-wire operation makes the part ideal for isolated systems where minimizing the number of interface lines minimizes the number of opto-isolators required in the system. The serial clock input is a Schmitt-triggered input to accommodate slow edges from optocouplers. The rise and fall times of other digital inputs to the AD7719 should be no longer than 1 μs. Most of the registers on the AD7719 are 8-bit registers, which facilitates easy interfacing to the 8-bit serial ports of microcontrollers. The main channel data register (AD0) on the AD7719 can be either 16 or 24 bits, the aux ADC data register (AD1) is 16 bits wide and the offset and gain registers are 24-bit registers, but data transfers to these registers can consist of multiple 8-bit transfers to the serial port of the microcontroller. DSP processors and microprocessors generally transfer 16 bits of data in a serial data operation. Some of these processors, such as the ADSP-2105, have the facility to program the amount of cycles in a serial transfer. This allows the user to tailor the number of bits in any transfer to match the register length of the required register in the AD7719. Even though some of the registers on the AD7719 are only eight bits in length, communicating with two of these registers in successive write operations can be handled as a single 16-bit data transfer, if required. For example, if the Filter register is to be updated, the processor must first write to the Communications register (saying that the next operation is a write to the Filter register) and then write eight bits to the Setup register. If required, this can all be done in a single 16-bit transfer because once the eight serial clocks of the write operation to the Communications register have been completed, the part immediately sets itself up for a write operation to the Setup register. REV. A AD7719 –28– START POWER-ON/RESET FOR AD7719 CONFIGURE AND INITIALIZE C/P SERIAL PORT WRITE TO THE COMMUNICATIONS REGISTER SELECTING NEXT OPERATION TO BE A WRITE TO THE IOCON REGISTER WRITE TO THE IOCON REGISTER TO CONFIGURE THE CURRENT SOURCES, DIGITAL I/O PORT, AND POWER SWITCHES WRITE TO COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A WRITE TO THE FILTER REGISTER WRITE TO FILTER REGISTER CONFIRMING THE REQUIRED UPDATE RATE WRITE TO COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A WRITE TO THE AUX CHANNEL ADC CONTROL REGISTER (AD1CON) WRITE TO AD1CON REGISTER ENABLING THE AUX ADC, SELECT THE INPUT CHANNEL BIPOLAR/ UNIPOLAR OPERATION AND INPUT RANGE WRITE TO COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A WRITE TO THE MAIN CHANNEL ADC CONTROL REGISTER (AD0CON) WRITE TO AD1CON REGISTER ENABLING THE MAIN ADC, SELECT THE INPUT CHANNEL, WORD LENGTH, BIPOLAR/UNIPOLAR OPERATION, AND INPUT RANGE WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A WRITE TO THE MODE REGISTER WRITE TO MODE REGISTER SELECTING FULL-SCALE CALIBRATION HARDWARE POLLING SOFTWARE POLLING HARDWARE POLLING POLL RDY PIN RDY LOW? NO YES WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A WRITE TO THE MODE REGISTER WRITE TO MODE REGISTER SELECTING ZERO-SCALE CALIBRATION POLL RDY PIN RDY LOW? NO YES WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A WRITE TO THE MODE REGISTER WRITE TO MODE REGISTER SELECTING CONTINUOUS CONVERSION MODE WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A READ FROM THE MAIN ADC DATA REGISTER (AD0) POLL RDY PIN RDY LOW? NO YES READ FROM DATA REGISTER (AD0) POLL RDY PIN RDY LOW? NO YES WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A READ FROM THE AUX ADC DATA REGISTER (AD1) READ FROM DATA REGISTER (AD0) SOFTWARE POLLING WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A READ FROM THE MODE REGISTER READ FROM MODE REGISTER MD BITS = 001? NO YES WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A WRITE TO THE MODE REGISTER WRITE TO MODE REGISTER SELECTING FULL-SCALE CALIBRATION WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A READ FROM THE MODE REGISTER READ FROM MODE REGISTER MD BITS = 001? NO YES WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A WRITE TO THE MODE REGISTER WRITE TO MODE REGISTER SELECTING CONTINUOUS CONVERSION MODE WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A WRITE TO THE MODE REGISTER WRITE TO MODE REGISTER SELECTING CONTINUOUS CONVERSION MODE WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A READ OF STATUS REGISTER READ STATUS REGISTER RDY0 = 1? NO YES WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A READ OF THE MAIN ADC DATA REGISTER (AD0) READ AD0 WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A READ OF STATUS REGISTER READ STATUS REGISTER RDY1 = 1? NO YES WRITE TO THE COMMUNICATIONS REGISTER SETTING UP NEXT OPERATION TO BE A READ OF THE AUX ADC DATA REGISTER (AD1) READ AD1 Figure 11. Flowchart for Initializing, Calibrating, and Reading Data from the AD7719 Main and Aux Channels REV. A AD7719 –29– AD7719-to-68HC11 Interface Figure 12 shows an interface between the AD7719 and the 68HC11 microcontroller. The diagram shows the minimum (3-wire) interface with CS on the AD7719 hardwired low. In this scheme, the RDY bits of the Status register are monitored to determine when the Data register is updated. RDY0 indicates the status of the main ADC channel while RDY1 indicates the status of the aux channel. An alternative scheme, which increases the number of interface lines to four, is to monitor the RDY output line from the AD7719. The monitoring of the RDY line can be done in two ways. First, RDY can be connected to one of the 68HC11’s port bits (such as PC0), which is configured as an input. This port bit is then polled to determine the status of RDY. The second scheme is to use an interrupt driven system, in which case the RDY output is connected to the IRQ input of the 68HC11. For interfaces that require control of the CS input on the AD7719, one of the port bits of the 68HC11 (such as PC1) that is configured as an output, can be used to drive the CS input. The 68HC11 is configured in the master mode with its CPOL bit set to a logic 1 and its CPHA bit set to a logic 1. When the 68HC11 is configured like this, its SCLK line idles high between data transfers. The AD7719 is not capable of full duplex operation. If the AD7719 is configured for a write operation, no data appears on the DOUT lines even when the SCLK input is active. Similarly, if the AD7719 is configured for a read operation, data presented to the part on the DIN line is ignored even when SCLK is active. 68HC11 AD7719 VDD SS SCK MISO MOSI RESET SCLK CS DIN DOUT VDD Figure 12. AD7719-to-68HC11 Interface AD7719-to-8xC51 Interface An interface circuit between the AD7719 and the 8xC51 microcontroller is shown in Figure 13. The diagram shows the minimum number of interface connections with CS on the AD7719 hardwired low. In the case of the 8xC51 interface, the minimum number of interconnects is just two. In this scheme, the RDY bits of the Status register are monitored to determine when the Data register is updated. The alternative scheme, which increases the number of interface lines to three, is to monitor the RDY output line from the AD7719. The monitoring of the RDY line can be done in two ways. First, RDY can be connected to one of the 8xC51’s port bits (such as P1.0) that is configured as an input. This port bit is then polled to determine the status of RDY. DVDD 8xC51 AD7719 P3.0 P3.1 RESET SCLK CS DVDD 10k DIN DOUT Figure 13. AD7719-to-8XC51 Interface The second scheme is to use an interrupt-driven system, in which case the RDY output is connected to the INT1 input of the 8xC51. For interfaces that require control of the CS input on the AD7719, one of the port bits of the 8xC51 (such as P1.1) that is configured as an output can be used to drive the CS input. The 8xC51 is configured in its Mode 0 serial interface mode. Its serial interface contains a single data line. As a result, the DOUT and DIN pins of the AD7719 should be connected together with a 10 kΩ pull-up resistor. The serial clock on the 8xC51 idles high between data transfers. The 8xC51 outputs the LSB first in a write operation, while the AD7719 expects the MSB first so the data to be transmitted has to be rearranged before being written to the output serial register. Similarly, the AD7719 outputs the MSB first during a read operation while the 8xC51 expects the LSB first. Therefore, the data read into the serial buffer needs to be rearranged before the correct data word from the AD7719 is available in the accumulator. REV. A AD7719 –30– AD7719-to-ADSP-2103/ADSP-2105 Interface Figure 14 shows an interface between the AD7719 and the ADSP-2103/ADSP-2105 DSP processor. In the interface shown, the RDY bits of the Status register are again monitored to determine when the Data register is updated. The alternative scheme is to use an interrupt-driven system, in which case the RDY output is connected to the IRQ2 input of the ADSP-2103/ ADSP-2105. The serial interface of the ADSP-2103/ADSP-2105 is set up for alternate framing mode. The RFS and TFS pins of the ADSP-2103/ADSP-2105 are configured as active low outputs and the ADSP-2103/ADSP-2105 serial clock line, SCLK, is also configured as an output. The CS for the AD7719 is active when either the RFS or TFS outputs from the ADSP-2103/ ADSP-2105 are active. The serial clock rate on the ADSP-2103/ ADSP-2105 should be limited to 3 MHz to ensure correct operation with the AD7719. RESET DVDD CS AD7719 SCLK SCLK DT DR RFS TFS ADSP-2103 / ADSP-2105 DIN DOUT Figure 14. AD7719-to-ADSP-2103/ADSP-2105 Interface CIRCUIT DESCRIPTION The AD7719 is a Σ-Δ A/D converter incorporating two independent Σ-Δ A/D converters with on-chip digital filtering, intended for the measurement of wide dynamic range, low frequency signals such as those in weigh scale, pressure, temperature, industrial control, or process control applications. The main ADC is intended to convert the primary sensor input. The main ADC employs a Σ-Δ conversion technique to realize up to 24 bits of no-missing-codes performance. The Σ-Δ modulator converts the sampled input signal into a digital pulse train whose duty cycle contains the digital information. A Sinc3 programmable low-pass filter is then employed to decimate the modulator output data stream to give a valid data conversion result at programmable output rates from 5.35 Hz (186.77 ms) to 105.03 Hz (9.52 ms). A Chopping scheme is also employed to minimize ADC offset and offset and gain drift errors. The analog input to the main ADC can be operated in buffered or unbuffered mode and can be programmed for one of eight input ranges from ±20 mV to ±2.56 V. The input channels can be configured for either fully differential inputs or pseudodifferential input channels via the CH1 and CH0 bits in the main ADC control register (AD0CON) and the CHCON bit in the mode register. When configured for buffered mode (BUF = 0), the input channels are internally buffered, allowing the part to handle significant source impedances on the analog input, allowing R/C filtering (for noise rejection or RFI reduction) to be placed on the analog inputs if required. When operating in unbuffered mode, care has to be exercised when selecting front end source impedances so as not to introduce gain errors. On-chip burnout currents are available and can be used to check that a transducer on the selected channel is still operational before attempting to take measurements. The second or auxiliary ADC is intended to convert secondary inputs such as those from a cold junction diode or thermistor. This ADC is unbuffered and has a fixed input range of 0 V to REFIN2 (ARN bit = 1) or 0 to REFIN2/2 (ARN bit = 0). Again, this ADC can be configured for differential or pseudodifferential inputs via the ACH2, ACH1, and ACH0 bits in the auxiliary ADC control register (AD1CON). The auxiliary ADC is specified for 16-bit performance and, since its analog inputs are unbuffered, care must be exercised when placing filtering on the front end to avoid introducing gain errors into the measurement system. The basic connection diagram for the AD7719 is shown in Figure 15. This shows both the AVDD and DVDD pins of the AD7719 being driven from the analog 5 V supply. Some applications will have AVDD and DVDD driven from separate supplies. AVDD and DVDD can be operated independently of each other, allowing the device to be operated with 5 V analog supply and 3 V digital supply or vice versa. An AD780/REF195 precision 2.5 V reference provides the reference source for the part. A quartz crystal or ceramic resonator provides the 32 kHz master clock source for the part. In some cases, it will be necessary to connect capacitors on the crystal or resonator to ensure that it does not oscillate at overtones of its fundamental operating frequency. The values of capacitors will vary depending on the manufacturer’s specifications. AD780/ REF195 XTAL1 XTAL2 RECEIVE (READ) P1/SW1 P2/SW2 P3 P4 0.1F 10F ANALOG 5V SUPPLY GND VIN VOUT PWRGND AGND DGND 10F 0.1F 0.1F AVDD DVDD IOUT1 IOUT2 AIN1 AIN2 AIN3 AIN4 AIN5 REFIN2 REFIN1(+) REFIN1(–) AIN6 RESET CS DOUT DIN SCLK 5V CHIP SELECT SERIAL DATA (WRITE) SERIAL CLOCK 32kHz CRYSTAL ANALOG 5V SUPPLY AD7719 Figure 15. Basic Connection Diagram REV. A AD7719 –31– Analog Input Channels The main ADC has four associated analog input pins (labeled AIN1 to AIN4) that can be configured as two fully differential input channels or three pseudodifferential input channels. Channel selection bits CH1 and CH0 in the ADC0CON register, along with the CHCON bit of the mode register, detail the different configurations. The auxiliary ADC has four external input pins (labeled AIN3 to AIN6) as well as an internal connection to the internal on-chip temperature sensor. Channel selection bits ACH2, ACH1, and ACH0 in the ADC1CON register, along with the CHCON bit in the mode register, detail the various configurations on these input channels. Two input multiplexers (MUX1 and MUX2) switch the selected input channel to the on-chip buffer amplifier in the case of the main ADC when operated in buffered mode, and directly to the Σ-Δ modulator input in the case of the auxiliary ADC and when the main ADC is operated in unbuffered mode. When the analog input channel is switched, the settling time of the part must elapse before a new valid word is available from the ADC. Figure 16 shows the analog input channel configurations available to the user when the CHCON bit in the mode register is set to a zero. In this case, the main ADC can be configured as one or two fully differential input channels (AIN1/AIN2 and AIN3/AIN4) and the aux can be configured as two single-ended inputs with respect to AGND (AIN3/AGND and AIN4/AGND) and one fully differential input AIN5/AIN6). The aux can also be configured as three single-ended inputs with respect to AGND (AIN3/AGND, AIN4/AGND, and AIN5/AGND) by tying AIN6 externally to AGND. The temp sensor is available as an internal connection. SINGLEENDED INPUT SINGLEENDED INPUT FULLY DIFFERENTIAL FULLY DIFFERENTIAL FULLY DIFFERENTIAL AIN(+) AIN(–) AIN(+) AIN(–) AIN1 AIN2 AIN3 AIN4 AIN1 AIN2 AIN3 AIN4 AIN5 AIN6 AIN3 AIN4 AGND MUX1 (MAIN ADC) (AUX ADC) AIN5 AIN6 MAIN CHANNEL AUX CHANNEL MUX2 Figure 16. Input Channel Configurations with CHCON = 0 Figure 17 shows the analog input channel configurations available to the user when the CHCON bit in the mode register is set to 1. In this case, the main ADC is configured as three pseudodifferential input channels (AIN1/AIN4, AIN2/AIN4, and AIN3/AIN4) and the aux can be configured as two single-ended inputs with respect to AGND (AIN5/AGND and AIN6/AGND) and one fully differential input (AIN5/AIN6). The temp sensor is available as an internal connection. SINGLEENDED INPUT FULLY DIFFERENTIAL AIN(+) AIN(–) AIN(+) AIN(–) AIN1 AIN2 AIN3 AIN4 AIN1 AIN2 AIN3 AIN4 AIN5 AIN6 AIN3 AIN4 AGND MUX1 (MAIN ADC) (AUX ADC) AIN5 AIN6 SINGLEENDED INPUT AIN3/AIN4 AIN2/AIN4 AIN1/AIN4 MUX2 MAIN CHANNEL AUX CHANNEL PSEUDODIFFERENTIAL INPUT PSEUDODIFFERENTIAL INPUT Figure 17. Input Channel Configurations with CHCON = 1 In buffered mode (BUF = 0), the output of the main ADC multiplexer feeds into a high impedance input stage of the buffer amplifier. As a result, the main ADC inputs can handle significant source impedances and are tailored for direct connection to external resistive-type sensors like strain gages or resistance temperature detectors (RTDs). The auxiliary ADC and the main ADC when operated with BUF = 1, however, are unbuffered, resulting in higher analog input current. It should be noted that these unbuffered input paths provide a dynamic load to the driving source. Therefore, resistor/capacitor combinations on the input pins can cause dc gain errors, depending on the output impedance of the source that is driving the ADC inputs. Table XVIII and XIX show the allowable external resistance/capacitance values for unbuffered mode such that no gain error at the 16- and 20-bit level, respectively, is introduced. The absolute input voltage range on the main ADC when operated in buffered mode is restricted to a range between AGND + 100 mV and AVDD – 100 mV. Care must be taken in setting up the common-mode voltage and input voltage range so that these limits are not exceeded; otherwise there will be a degradation in linearity and noise performance. REV. A AD7719 –32– The absolute input voltage range on the auxiliary ADC and the main ADC in unbuffered mode includes the range between AGND – 30 mV to AVDD + 30 mV as a result of being unbuffered. The negative absolute input voltage limit does allow the possibility of monitoring small true bipolar signals with respect to AGND. Programmable Gain Amplifier The output from the buffer on the main ADC is applied to the input of the on-chip programmable gain amplifier (PGA). The PGA can be programmed through eight different unipolar and bipolar ranges. The PGA gain range is programmed via the range bits in the ADC0CON register. With an external 2.5 V reference applied, the unipolar ranges are 0 mV to 20 mV, 0 mV to 40 mV, 0 mV to 80 mV, 0 mV to 160 mV, 0 mV to 320 mV, 0 mV to 640 mV, 0 V to 1.28 V and 0 V to 2.56 V while bipolar ranges are ±20 mV, ±40 mV, ±80 mV, ±160 mV, ±320 mV, ±640 mV, ±1.28 V, and ±2.56 V. These are the ranges that should appear at the input to the on-chip PGA. The ADC range matching specification of 2 μV (typ) across all ranges means that calibration need only be carried out on a single range and does not have to be repeated when the PGA range is changed. This is a significant advantage when compared with similar ADCs available on the market. Typical matching across ranges is shown in Figure 18. Here, the primary ADC is configured in fully differential, bipolar mode with an external 2.5 V reference, while an analog input voltage of just greater than 19 mV is forced on its analog inputs. The ADC continuously converts the dc voltage at an update rate of 5.35 Hz, i.e., SF = 0xFF. In total, 800 conversion results are gathered. The first 100 results are gathered with the primary ADC operating in the ±20 mV range. The ADC range is then switched to ±40 mV and 100 more results are gathered; this continues until the last 100 samples are gathered with the ADC configured in the ±2.5 V range. From Figure 18, the variation in the sample mean through each range, i.e., the range matching, is seen to be on the order of 2 μV. The auxiliary ADC does not incorporate an eight range PGA. The aux ADC operates at a gain of 1 or a gain of 2 as determined by the ARN bit in the AD1CON register. 0 100 200 300 400 500 600 700 800 SAMPLE COUNT ADC INPUT VOLTAGE ( mV) 19.372 19.371 19.370 19.369 19.368 19.367 19.366 19.365 19.364 ADC RANGE 20mV 40mV 80mV 160mV 320mV 640mV 1.28V 2.56V Figure 18. Main ADC Range Matching Bipolar/Unipolar Configuration The analog inputs on the AD7719 can accept either unipolar or bipolar input voltage ranges. Bipolar input ranges do not imply that the part can handle negative voltages with respect to system AGND. Unipolar and bipolar signals on the AIN(+) input on the main ADC are referenced to the voltage on the respective AIN(–) input. AIN(+) and AIN(–) refer to the signals seen by the modulator that come from the output of the multiplexer, as shown in Figures 16 and 17. For example, if AIN(–) is 2.5 V and the main ADC is configured for an analog input range of 0 mV to 20 mV, the input voltage range on the AIN(+) input is 2.5 V to 2.52 V. If AIN(–) is 2.5 V and the AD7719 is configured for an analog input range of ±1.28 V, the analog input range on the AIN(+) input is 1.22 V to 3.78 V (i.e., 2.5 V ± 1.28 V). Bipolar or unipolar options are chosen by programming the main and auxiliary U/B bit in the ADC0CON and ADC1CON registers, respectively. This programs the relevant ADC for either unipolar or bipolar operation. Programming for either unipolar or bipolar operation does not change any of the input signal conditioning; it simply changes the data output coding and the points on the transfer function where calibrations occur. Table XVIII. Max Resistance for No 16-Bit Gain Error (Unbuffered Mode) External Capacitance Gain 0 pF 50 pF 100 pF 500 pF 1000 pF 5000 pF 1 111.3K 27.8K 16.7K 4.5K 2.58K 700 2 53.7K 13.5K 8.1K 2.2K 1.26K 360 4 25.4K 6.4K 3.9K 1.0K 600 170 8–128 10.7K 2.9K 1.7K 480 270 75 Table XIX. Max Resistance for No 20-Bit Gain Error (Unbuffered Mode) External Capacitance Gain 0 pF 50 pF 100 pF 500 pF 1000 pF 5000 pF 1 84.9K 21.1K 12.5K 3.2K 1.77K 440 2 42.0K 10.4K 6.1K 1.6K 880 220 4 20.5K 5.0K 2.9K 790K 430 110 8–128 8.8K 2.3K 1.3K 370 195 50 REV. A AD7719 –33– Data Output Coding When the ADC is configured for unipolar operation, the output coding is natural (straight) binary with a zero differential input voltage resulting in a code of 000 . . . 000, a midscale voltage resulting in a code of 100 . . . 000, and a full-scale input voltage resulting in a code of 111 . . . 111. The output code for any analog input voltage on the main ADC can be represented as follows: Code AIN GAIN N V = ( × × 2 ) (1.024 × REF ) Where AIN is the analog input voltage, GAIN is the PGA gain, i.e., 1 on the 2.56 V range and 128 on the 20 mV range, and N = 16 in 16-bit mode and N = 24 in 24-bit mode of operation. The output code for any analog input voltage on the aux ADC can be represented as follows: Code AIN GAIN N V = ( × × 2 ) REF Where AIN is the analog input voltage, GAIN is 1 or 2, determined by the ARN bit in the aux ADC control register, i.e., 1 on the VREF range and 2 on the VREF/2 range, and N = 16. When an ADC is configured for bipolar operation, the coding is offset binary with a negative full-scale voltage resulting in a code of 000 . . . 000, a zero differential voltage resulting in a code of 100 . . . 000, and a positive full-scale voltage resulting in a code of 111 . . . 111. The output code from the main ADC for any analog input voltage can be represented as follows: Code N AIN GAIN V = 2 × [( × (1 024 × REF )) + 1] –1 . Where AIN is the analog input voltage, GAIN is the PGA gain, i.e., 1 on the ±2.56 V range and 128 on the ±20 mV range, N = 16 in 16-bit mode, and N = 24 in 24-bit mode of operation. The output code from the aux ADC for any analog input voltage can be represented as follows: Code N AIN GAIN V = 2 × [( × REF ) + 1] –1 Where AIN is the analog input voltage, GAIN is 1 or 2, determined by the ARN bit in the aux ADC control register, i.e., 1 on the ±VREF range, 2 on the ±VREF/2 range, and N = 16. Burnout Currents The main ADC on the AD7719 contains two 100 nA constant current generators, one sourcing current from AVDD to AIN(+), and one sinking current from AIN(–) to AGND. The currents are switched to the selected analog input pair. Both currents are either on or off, depending on the Burnout Current Enable (BO) bit in the IOCON register. These currents can be used to verify that an external transducer is still operational before attempting to take measurements on that channel. Once the burnout currents are turned on, they will flow in the external transducer circuit, and a measurement of the input voltage on the analog input channel can be taken. If the resultant voltage measured is full-scale, the user needs to verify why this is the case. A full-scale reading could mean that the front end sensor is open circuit; it could also mean that the front end sensor is overloaded and is justified in outputting full-scale, or that the reference may be absent and the NOXREF bit is set, thus clamping the data to all 1s. When reading all 1s from the output, the user needs to check these three cases before making a judgment. If the voltage measured is 0 V, it may indicate that the transducer has short circuited. For normal operation, these burnout currents are turned off by writing a 0 to the BO bit in the IOCON register. The current sources work over the normal absolute input voltage range specifications with buffers on. Excitation Currents The AD7719 also contains two matched, software configurable 200 μA constant current sources. Both source current from AVDD that is directed to either the IOUT1 or IOUT2 pins of the device. These current sources are controlled via bits in the IOCON register. The configuration bits enable the current sources and can be configured to source 200 μA individually to both pins or a combination of both currents, i.e., 400 μA to either of the selected output pins. These current sources can be used to excite external resistive bridge or RTD sensors. Crystal Oscillator The AD7719 is intended for use with a 32.768 kHz watch crystal. A PLL internally locks onto a multiple of this frequency to provide a stable 4.194304 MHz clock for the ADC. The modulator sample rate is the same as the crystal oscillator frequency. The start-up time associated with 32 kHz crystals is typically 300 ms. The OSPD bit in the mode register can be used to prevent the oscillator from powering down when the AD7719 is placed in power-down mode. This avoids having to wait 300 ms after exiting power-down to start a conversion at the expense of raising the power-down current. Reference Input The AD7719 has a fully differential reference input capability for the main channel while the auxiliary channel accepts only a single-ended reference. On the main channel, the reference inputs REFIN1(+) and REFIN1(–) provide a differential reference input capability. The common-mode range for these differential inputs is from AGND to AVDD. The reference input is unbuffered, and therefore excessive R-C source impedances will introduce gain errors. The nominal reference voltage, VREF, (REFIN1(+) – REFIN1(–), for specified operation is 2.5 V, but the AD7719 is functional with reference voltages from 1 V to AVDD. In applications where the excitation (voltage or current) for the transducer on the analog input also drives the reference voltage for the part, the effect of the low frequency noise in the excitation source will be removed as the application is ratiometric. If the AD7719 is used in a nonratiometric application, a low noise reference should be used. Recommended reference voltage sources for the AD7719 include the AD780, REF43, and REF192. It should also be noted that the reference inputs provide a high impedance, dynamic load. Because the input impedance of each reference input is dynamic, resistor/capacitor combinations on these inputs can cause dc gain errors, depending on the output impedance of the source that is driving the reference inputs. Reference voltage sources like those recommended (e.g., AD780) will typically have low output impedances and are therefore tolerant to having decoupling capacitors on the REFIN1(+) without introducing gain errors in the system. Deriving the reference input voltage across an external resistor, as shown in Figure 19, will mean that the reference input sees a significant external source impedance. External decoupling on the REFIN1(+) and REFIN1(–) pins would not be recommended in this type of circuit configuration. The auxiliary channel conversion results are based on the voltage applied to REFIN2. This is a single-ended reference input specified for 2.5 V operation but functional with input voltages from 1 V to AVDD. REV. A AD7719 –34– Reference Detect The AD7719 includes on-chip circuitry to detect if the part has a valid reference on the main ADC for conversions or calibrations. If the voltage between the external REFIN1(+) and REFIN1(–) pins goes below 0.3 V or either the REFIN1(+) or REFIN1(–) inputs are open circuit, the AD7719 detects that it no longer has a valid reference. In this case, the NOXREF bit of the Status register is set to 1. If the AD7719 is performing normal conversions and the NOXREF bit becomes active, the conversion results revert to all 1s. Therefore, it is not necessary to continuously monitor the status of the NOXREF bit when performing conversions. It is only necessary to verify its status if the conversion result read from the ADC data register is all 1s. If the AD7719 is performing either an offset or gain calibration and the NOXREF bit becomes active, the updating of the respective calibration registers is inhibited to avoid loading incorrect coefficients to these registers, and the ERR0 bit in the Status register is set. If the user is concerned about verifying that a valid reference is in place every time a calibration is performed, the status of the ERR0 bit should be checked at the end of the calibration cycle. Reset Input The RESET input on the AD7719 resets all the logic, the digital filter, and the analog modulator while all on-chip registers are reset to their default state. RDY is driven high and the AD7719 ignores all communications to any of its registers while the RESET input is low. When the RESET input returns high, the AD7719 operates with its default setup conditions and it is necessary to set up all registers and carry out a system calibration if required after a RESET command. Power-Down Mode Loading 0, 0, 0 to the MD2, MD1, MD0 bits in the ADC mode register places the AD7719 in device power-down mode. Device power-down mode is the default condition for the AD7719 on power-up. Individual ADCs (main or auxiliary) can be put in power-down mode using the AD0EN in the main ADC control register (AD0CON) to power off the main ADC or the AD1EN in the auxiliary ADC control register (AD1CON) to power off the auxiliary ADC. The AD7719 retains the contents of all its on-chip registers (including the data register) while in powerdown or ADC disable mode. The device power-down mode does not affect the digital interface, and it does affect the status of the RDY pin. Putting the AD7719 into power-down or idle mode will reset the RDY line high. Placing the part in power-down mode reduces the total current (AIDD + DIDD) to 31 μA max when the part is operated at 5 V and the oscillator is allowed to run during power-down mode. With the oscillator shuts down, the total IDD is 3 μA max at 3 V and 9 μA max at 5 V. Idle Mode The AD7719 also contains an idle mode. The ADC defaults to this mode on completion of a calibration sequence and on the completion of a conversion when operating in single conversion mode. In idle mode, the power consumption of the AD7719 is not reduced below the normal mode dissipation. ADC Disable Mode This mode is entered by setting both the AD0EN and AD1EN bits in the main and max ADC control registers to 0 and setting the Mode bits (MD2, MD1, MD0) in the Mode register to non-0. In this mode, the internal PLL is enabled and the user can activate the current sources and power switches, but the power consumption of the ADC is reduced as both ADCs are disabled. In this mode, the AIDD is reduced to 0.15 mA and the DIDD is reduced to 0.35 mA max at 3 V and to 0.4 mA max with DVDD = 5 V. Calibration The AD7719 provides four calibration modes that can be programmed via the mode bits in the mode register. One of the major benefits of the AD7719 is that it is factory-calibrated as part of the final test process with the generated coefficients stored within the ADC. At power-on, the factory gain calibration coefficients are automatically loaded to the gain calibration registers on the AD7719. Each ADC (primary and auxiliary) has dedicated calibration register pairs as outlined in the AD0CON and AD1CON register descriptions. Given that the ADC is factory-calibrated and a chopping scheme is employed that gives excellent offset and drift performance, it is envisaged that in the majority of applications the user will not need to perform any field calibrations. However, the factory calibration values in the ADC calibration registers will be overwritten if any one of the four calibration options are initiated. Even though an internal offset calibration mode is described below, it should be recognized that both ADCs are chopped. This chopping scheme inherently minimizes offset and means that an internal offset calibration should never be required. Also, because factory 25°C gain calibration coefficients are automatically present at power-on, an internal full-scale calibration will only be required if the part is being operated at temperatures significantly different from 25°C or away from the calibration conditions. The AD7719 offers internal or system calibration facilities. For full calibration to occur on the selected ADC, the calibration logic must record the modulator output for two different input conditions. These are zero-scale and fullscale points derived by performing a conversion on the different input voltages provided to the input of the modulator during calibration. The result of the zero-scale calibration conversion is stored in the offset calibration registers for the appropriate ADC. The result of the full-scale calibration conversion is stored in the gain calibration registers for the appropriate ADC. With these readings, the calibration logic can calculate the offset and the gain slope for the input-to-output transfer function of the converter. During an internal zero-scale or full-scale calibration, the respective zero input and full-scale input are automatically connected to the ADC input pins internally to the device. A system calibration, however, expects the system zero-scale and system full-scale voltages to be applied to the external ADC pins before the calibration mode is initiated. In this way, external ADC errors are taken into account and minimized as a result of system calibration. It should also be noted that to optimize calibration accuracy, all AD7719 ADC calibrations are automatically carried out at the slowest update rate. REV. A AD7719 –35– Internally in the AD7719, the coefficients are normalized before being used to scale the words coming out of the digital filter. The offset calibration coefficient is subtracted from the result prior to the multiplication by the gain coefficient. From an operational point of view, a calibration should be treated like another ADC conversion. A zero-scale calibration (if required) should always be carried out before a full-scale calibration. System software should monitor the relevant ADC RDY0/1 bit in the Status register to determine end of calibration via a polling sequence or interrupt driven routine. Grounding and Layout Since the analog inputs and reference input on the main ADC are differential, most of the voltages in the analog modulator are common-mode voltages. The excellent common-mode rejection of the part will remove common-mode noise on these inputs. The analog and digital supplies to the AD7719 are independent and separately pinned out to minimize coupling between the analog and digital sections of the device. The AD7719 can be operated with 5 V analog and 3 V digital supplies, or vice versa. The digital filter will provide rejection of broadband noise on the power supplies, except at integer multiples of the modulator sampling frequency. The digital filter also removes noise from the analog and reference inputs provided these noise sources do not saturate the analog modulator. As a result, the AD7719 is more immune to noise interference than a conventional high resolution converter. However, because the resolution of the AD7719 is so high, and the noise levels from the AD7719 are so low, care must be taken with regard to grounding and layout. The printed circuit board that houses the AD7719 should be designed such that the analog and digital sections are separated and confined to certain areas of the board. This facilitates the use of ground planes that can be easily separated. A minimum etch technique is generally best for ground planes as it gives the best shielding. Although the AD7719 has separate pins for analog and digital ground, the AGND and DGND pins are tied together within the device via the substrate. The user must not tie these pins external to separate ground planes unless the ground planes are connected together near the AD7719. In systems where the AGND and DGND are connected somewhere else in the system, i.e., the system power supply, they should not be connected again at the AD7719 as a ground loop will result. In these situations, it is recommended that the AD7719’s AGND and DGND pins be tied to the AGND plane. In any layout, it is important that the user keep in mind the flow of currents in the system, ensuring that the return paths for all currents are as close as possible to the paths the currents took to reach their destinations. Avoid forcing digital currents to flow through the AGND sections of the layout. The PWRGND pin is tied internally to AGND on the AD7719. The PWRGND pad internally has a resistance of less then 50mΩ to the PWRGND pin, while the resistance back to the AGND pad is >3 Ω. This means that 19.5 mA of the maximum specified current (20 mA) will flow to PWRGND with the remaining 0.5 mA flowing to AGND. PWRGND and AGND should be tied together at the AD7719 and it is important to minimize the resistance on the ground return lines. Avoid running digital lines under the device as these will couple noise onto the die. The analog ground plane should be allowed to run under the AD7719 to prevent noise coupling. The power supply lines to the AD7719 should use as wide a trace as possible to provide low impedance paths and reduce the effects of glitches on the power supply line. Fast switching signals like clocks should be shielded with digital ground to avoid radiating noise to other sections of the board, and clock signals should never be run near the analog inputs. Avoid crossover of digital and analog signals. Traces on opposite sides of the board should run at right angles to each other. This will reduce the effects of feedthrough through the board. A microstrip technique is by far the best, but is not always possible with a double-sided board. In this technique, the component side of the board is dedicated to ground planes while signals are placed on the solder side. Good decoupling is important when using high resolution ADCs. All analog supplies should be decoupled with 10 μF tantalum in parallel with 0.1 μF capacitors to AGND. To achieve the best from these decoupling components, they have to be placed as close as possible to the device, ideally right up against the device. All logic chips should be decoupled with 0.1 μF ceramic capacitors to DGND. In systems where a common supply voltage is used to drive both the AVDD and DVDD of the AD7719, it is recommended that the system’s AVDD supply be used. This supply should have the recommended analog supply decoupling capacitors between the AVDD pin of the AD7719 and AGND, and the recommended digital supply decoupling capacitor between the DVDD pin of the AD7719 and DGND. APPLICATIONS The AD7719 provides a low cost, high resolution analog-todigital function. Because the analog-to-digital function is provided by a Σ-Δ architecture, it makes the part more immune to noisy environments, making it ideal for use in sensor measurement and industrial and process control applications. Given the architecture used in the AD7719, where the signal chain is chopped and the device is factory-calibrated at final test, field calibration can be avoided due to the extremely low offset and gain drifts exhibited by this converter. It also provides a programmable gain amplifier, a digital filter, and system calibration options. Thus, it provides far more system-level functionality than off-the-shelf integrating ADCs without the disadvantage of having to supply a high quality integrating capacitor. In addition, using the AD7719 in a system allows the system designer to achieve a much higher level of resolution because noise performance of the AD7719 is significantly better than that of integrating ADCs. The on-chip PGA allows the AD7719 to handle an analog input voltage range as low as 10 mV full-scale with VREF = 1.25 V. The differential inputs of the part allow this analog input range to have an absolute value anywhere between AGND + 100 mV and AVDD – 100 mV. It allows the user to connect the transducer directly to the input of the AD7719. The programmable gain front end on the AD7719 allows the part to handle unipolar analog input ranges from 0 mV to 20 mV to 0 V to 2.5 V and bipolar inputs of ±20 mV to ±2.5 V. Because the part operates from a single supply, these bipolar ranges are with respect to a biased-up differential input. Another key advantage of the AD7719 is that it contains two Σ-Δ converters operating in parallel; thus the user does not need to interrupt the main channel when a secondary measurement on a different variable needs to be performed. REV. A AD7719 –36– Pressure Measurement One typical application of the AD7719 is pressure measurement. Figure 19 shows the AD7719 used with a pressure transducer, the BP01 from Sensym. The pressure transducer is arranged in a bridge network and gives a differential output voltage between its OUT(+) and OUT(–) terminals. With rated full-scale pressure (in this case 300 mmHg) on the transducer, the differential output voltage is 3 mV/V of the input voltage (i.e., the voltage between its IN(+) and IN(–) terminals). Assuming a 5 V excitation voltage, the full-scale output range from the transducer is 15 mV. The excitation voltage for the bridge can be used to directly provide the reference for the ADC as the reference input range includes the supplies. Alternatively, a suitable resistor divider can be implemented that allows the full dynamic range of the input to be utilized in these application. This implementation is fully ratiometric, so variations in the excitation voltage do not introduce errors in the system. Choosing resistor values of 20 kΩ and 12 kΩ as per Figure 19 give a 1.875 V reference voltage for the AD7719 when the excitation voltage is 5 V. AD7719 IN+ OUT+ OUT– IN– 20k 12k EXCITATION VOLTAGE = 5V AVDD DVDD AIN1 AIN2 REFIN1(+) P1 PWRGND DGND AGND REFIN2(–) Figure 19. Pressure Measurement Using AD7719 Using the part with a programmed gain of 128 results in the full-scale input span of the AD7719 being 15 mV, which corresponds with the output span from the transducer. A second key advantage to using the AD7719 in transducer based applications is that the on-chip low-side power switch can be fully utilized in low power applications. The low-side power switch is connected in series with the cold side of the bridge. In normal operation, the switch is closed and measurements can be taken from the bridge. In applications where power is of concern, the AD7719 can be put in low power mode, substantially reducing the power burned in the application. In addition to this, the power switch can be opened while in low power mode thus avoiding the unnecessary burning of power in the front end transducer. When the AD7719 is taken back out of power-down and the power switch is closed, the user should ensure that the front end circuitry is fully settled before attempting a read from the AD7719. The circuit in Figure 20 shows a method that utilizes all three pseudodifferential input channels on the AD7719 main channel to temperature-compensate a pressure transducer. 5V OUT(+) OUT(–) IN(–) IN(+) I1 I2 PRESSURE BRIDGE XTAL1 XTAL2 IOUT1 6.25k AVDD REFIN(+) REFIN(–) AIN2 AIN1 AIN3 AIN4 AGND AD7719 250 Figure 20. Temperature-Compensating a Pressure Transducer In this application, pseudodifferential input channel AIN1/AIN4 is used to measure the bridge output while pseudodifferential channels AIN2/AIN4 and AIN3/AIN4 measure the voltage across the bridge. The voltage measured across the bridge will vary proportionally with temperature, and the delta in this voltage can be used to temperature-compensate the output of the pressure bridge. Temperature Measurement The AD7719 is also useful in temperature measurement applications; Figure 21 shows an RTD temperature measurement application. In this application, the transducer is an RTD (resistive temperature device), a PT100. The arrangement is a 4-lead RTD configuration. There are voltage drops across the lead resistances RL1 and RL4, but these simply shift the common-mode voltage. There is no voltage drop across lead resistances RL2 and RL3 as the input current to the AD7719 is very low, looking into a high input impedance buffer. RCM is included to shift the analog input voltage to ensure that it lies within the common-mode range (AGND + 100 mV to AVDD – 100 mV) of the ADC. In the application shown, the on-chip 200 μA current source provides the excitation current for the PT100 and also generates the reference voltage for the AD7719 via the 12.5 kΩ resistor. Variations in the excitation current do not affect the circuit as both the input voltage and the reference voltage vary ratiometrically with the excitation current. However, the 12.5 kΩ resistor must have a low temperature coefficient to avoid errors in the reference voltage over temperature. REV. A AD7719 –37– REFIN(–) IOUT1 5V 12.5k AVDD AIN2 AIN1 AD7719 REFIN(+) CONTROLLER IOUT2 DVDD DGND AGND PWRGND DRDY SCLK DIN DOUT CS XTAL1 XTAL2 RL1 RREF RL2 RL3 RL4 RCM RTD 200A Figure 21. 4-Wire RTD Temperature Measurement Using the AD7719 Figure 22 shows a further enhancement to the circuit shown in Figure 21. Generally, dc excitation has been accepted as the normal method of exciting resistive-based sensors like RTDs (resistance temperature detectors) in temperature measurement applications. With dc excitation, the excitation current through the sensor must be large enough so that the smallest temperature/resistance change to be measured results in a voltage change that is larger than the system noise, offset, and drift of the system. The purpose of switching the excitation source is to eliminate dc-induced errors. DC errors (EMF1 and EMF2) due to parasitic thermocouples produced by differential metal connections (solder and copper track) within the circuit are also eliminated when using this switching arrangement. This excitation is a form of synchronous detection where the sensor is excited with an alternating excitation source and the ADC only measures information in the same phase as the excitation source. REFIN(–) IOUT1 IOUT2 AVDD AIN2 AIN1 AIN3 AIN4 AD7719 REFIN(+) MUX1 RREF A A BUF AND PGA 200A I1 EMF1 RESISTIVE TRANSDUCER EMF2 Figure 22. Low Resistance Measurement AD7719 The switched polarity current source is developed using the on-chip current sources and external phase control switches (A and A) driven from the controller. During the conversion process, the AD7719 takes two conversion results, one on each phase. During Phase 1, the on-chip current source is directed to IOUT1 and flows top to bottom through the sensor and switch controlled by A. In Phase 2, the current source is directed to IOUT2 and flows in the opposite direction through the sensor and through switch controlled by A. In all cases, the current flows in the same direction through the reference resistor to develop the reference voltage for the ADC. All measurements are ratiometrically derived. The results of both conversions are combined within the microcontroller to produce one output measurement representing the resistance or temperature of the transducer. For example, if the RTD output during Phase 1 is 10 mV, a 1 mV circuit-induced dc error exists due to parasitic thermocouples, and the ADC measures 11 mV. During the second phase, the excitation current is reversed and the ADC measures –10 mV from the RTD and again sees 1 mV dc error, giving an ADC output of –9 mV during this phase. These measurements are processed in the controller (11 mV – (–9 mV)/2 = 10 mV), thus removing the dc-induced errors within the system. In the circuit shown in Figure 22, the resistance measurement is made using ratiometric techniques. Resistor RREF, which develops the ADC reference, must be stable over temperature to prevent reference-induced errors in the measurement output. 3-Wire RTD Configurations To fully optimize a 3-wire RTD configuration, two identically matched current sources are required. The AD7719, which contains two well-matched current sources, is ideally suited to these applications. One possible 3-wire configuration using the AD7719 is outlined in Figure 23. REFIN(–) IOUT1 DGND AGND 5V 12.5k AVDD AIN2 AIN1 AD7719 RL3 RCM REFIN(+) CONTROLLER IOUT2 DVDD DRDY SCLK DIN DOUT CS XTAL1 XTAL2 RL2 RTD 200A 200A RL1 Figure 23. 3-Wire RTD Configuration Using the AD7719 REV. A AD7719 –38– In this 3-wire configuration, the lead resistances will result in errors if only one current source is used because the 200 μA will flow through RL1, developing a voltage error between AIN1 and AIN2. In the scheme outlined below, the second RTD current source is used to compensate for the error introduced by the 200 μA flowing through RL1. The second RTD current flows through RL2. Assuming RL1 and RL2 are equal (the leads would normally be of the same material and of equal length), and IOUT1 and IOUT2 match, the error voltage across RL2 equals the error voltage across RL1, and no error voltage is developed between AIN1 and AIN2. Twice the voltage is developed across RL3, but since this is a common-mode voltage, it will not introduce errors. RCM is included so the current flowing through the combination of RL3 and RCM develops enough voltage that the analog input voltage seen by the AD7719 is within the common-mode range of the ADC. The reference voltage for the AD7719 is also generated using one of these matched current sources. This reference voltage is developed across the 12.5 kΩ resistor as shown, and applied to the differential reference inputs of the AD7719. This scheme ensures that the analog input voltage span remains ratiometric to the reference voltage. Any errors in the analog input voltage due to the temperature drift of the RTD current source is compensated for by the variation in the reference voltage. The typical drift matching between the two RTD current sources is less than 1 ppm/°C. The voltage on either IOUT pin can go to within 0.6 V of the AVDD supply. Smart Transmitters Smart transmitters are another key design-in area for the AD7719. The dual Σ-Δ converter, single-supply operation, 3-wire interface capabilities, and small package size are all of benefit in smart transmitters. Here, the entire smart transmitter must operate from the 4 to 20 mA loop. Tolerances in the loop mean that the amount of current available to power the transmitter is as low as 3.5 mA. Figure 24 shows a block diagram of a smart transmitter, which includes the AD7719. Not shown in Figure 24 is the isolated power source required to power the front end. The advantages of the AD7719 in these applications is the dual-channel operation, meaning that the user does not have to interrupt the main channel when measuring secondary variables, and therefore does not have the latency associated with the settling times of the digital filter. The fact that the AD7719 is factory-calibrated means that in the majority of applications, the user will not have to perform any field calibration given the excellent offset and gain drift performance of the device as a result of the signal chain chopping employed in the signal chain. MICROCONTROLLER REF OUT2 REF IN 10F 0.1F DVDD AVDD AIN1 AIN2 CS DOUT SCLK DIN DGND AGND AIN5 AIN5 COM REFIN2 REFIN(+) REFIN(–) 0.1F REF OUT1 CLOCK LATCH DATA 4.7F COM C1 C2 C3 VCC GND AD7719 AD421 BOOST VCC LV 0.01F 1k 1000pF LOOP POWER 10F 3.3V 1.25V DN25D 0.01F LOOP RTN COMP DRIVE MAIN VARIABLES SECONDARY VARIABLES AIN3 AIN4 Figure 24. Smart Transmitter Employing the AD7719 REV. A AD7719 –39– OUTLINE DIMENSIONS 28-Lead Standard Small Outline Package [SOIC] Wide Body (R-28) Dimensions shown in millimeters and (inches) CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN COMPLIANT TO JEDEC STANDARDS MS-013AE 0.32 (0.0126) 0.23 (0.0091) 8 0 0.75 (0.0295) 0.25 (0.0098) 45 1.27 (0.0500) 0.40 (0.0157) SEATING PLANE 0.30 (0.0118) 0.10 (0.0039) 0.51 (0.0201) 0.33 (0.0130) 2.65 (0.1043) 2.35 (0.0925) 1.27 (0.0500) BSC 28 15 1 14 18.10 (0.7126) 17.70 (0.6969) 10.65 (0.4193) 10.00 (0.3937) 7.60 (0.2992) 7.40 (0.2913) COPLANARITY 0.10 28-Lead Thin Shrink Small Outline Package [TSSOP] (RU-28) Dimensions shown in millimeters 4.50 4.40 4.30 28 15 1 14 9.80 9.70 9.60 6.40 BSC PIN 1 SEATING PLANE 0.15 0.05 0.30 0.19 0.65 BSC 1.20 MAX 0.20 0.09 0.75 0.60 0.45 8 0 COMPLIANT TO JEDEC STANDARDS MS-153AE COPLANARITY 0.10 –40– C02460–0–4/03(A) REV. A AD7719 Revision History Location Page 4/03—Data Sheet changed from REV. 0 to REV. A. Updated format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .Universal Changes to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39 Input Preamp Ladder Attenuator Output Amp Common Mode Control Bandwidth Limiting Circuitry Aux Amp Overvoltage Clamp GND 5,8 VCC 3,4 VCM_Aux 16 Serial Peripheral Interface 12 VDD Hi Gain or Low Gain 10 Step 2 dB/Step +IN -IN 6 7 10 9 11 SDIO CS SCLK 13 VCM 15 14 1 2 -OUT Aux +OUT Aux -OUT +OUT AUXILIARY OUTPUT MAIN OUT 72µ$'&¶ LMH6518 50: 50: 50: 50: Overvoltage Clamp Overvoltage Clamp LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 LMH6518 900 MHz, Digitally Controlled, Variable Gain Amplifier Check for Samples: LMH6518 1FEATURES DESCRIPTION The LMH6518 is a digitally controlled variable gain 2• Gain Range 40 dB • Gain Step Size 2 dB amplifier whose total gain can be varied from −1.16 dB to 38.8 dB for a 40 dB range in 2 dB steps. The • Combined Gain Resolution with −3 dB bandwidth is 900 MHz at all gains. Gain Gsample/Second ADC’s 8.5 mdB accuracy at each setting is typically 0.1 dB. When • Min Gain −1.16 dB used in conjunction with a Texas Instruments • Max Gain 38.8 dB Gsample/second (Gsps) ADC with adjustable full scale (FS) range, the LMH6518 gain adjustment will • −3 dB BW 900 MHz accommodate full scale input signals from 6.8 mVPP • Rise/Fall Time <500 ps to 920 mVPP to get 700 mVPP nominal at the ADC • Recovery Time <5 ns input. The Auxiliary output (“+OUT Aux” and “−OUT Aux”) follows the Main output and is intended for use • Propagation Delay Variation 100 ps in Oscilloscope trigger function circuitry but may have • HD2 @ 100 MHz −50 dBc other uses in other applications. • HD3 @ 100 MHz −53 dBc The LMH6518 gain is programmed via a SPI-1 • Input-Referred Noise (Max Gain) 0.98 nV/√Hz compatible serial bus. A signal path combined gain • Over-Voltage Clamps for Fast Recovery resolution of 8.5 mdB can be achieved when the LMH6518’s gain and the Gsps ADC’s FS input are • Power Consumption — Auxiliary Turned Off both manipulated. Inputs and outputs are DC- 1.1W0.75W coupled. The outputs are differential with individual Common Mode (CM) voltage control (for Main and APPLICATIONS Auxiliary outputs) and have a selectable bandwidth • Oscilloscope Programmable Gain Amplifier limiting circuitry (common to both Main and Auxiliary) of 20, 100, 200, 350, 650, 750 MHz or full bandwidth. • Differential ADC Drivers • High Frequency Single-Ended Input to Differential Conversion • Precision Gain Control Applications • Medical Applications • RF/IF Applications Functional Block Diagram 1 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. 2All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Copyright © 2008–2013, Texas Instruments Incorporated Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. Absolute Maximum Ratings (1)(2) ESD Tolerance (3) Human Body Model 2000V Machine Model 200V Charge Device Model 1000V Supply Voltage VCC (5V nominal) 5.5V VDD (3.3V nominal) 3.6V Differential Input ±1V Input Common Mode Voltage 1V to 4V VCM and VCM_Aux 2V SPI Inputs 3.6V Maximum Junction Temperature 150°C Storage Temperature Range −65°C to 150°C Soldering Information Infrared or Convection (20 sec.) 235°C Wave Soldering (10 sec.) 260°C (1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not ensured. For ensured specifications, see the Electrical Characteristics tables. (2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and specifications. (3) Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of JEDEC) Field-Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC). Operating Ratings (1) Supply Voltage VCC = 5V (±5%) VDD = 3.3V (±5%) Temperature Range −40°C to 85°C (1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not ensured. For ensured specifications, see the Electrical Characteristics tables. Thermal Properties Temperature Range (1) −40°C to 85°C Junction-to-Ambient Thermal Resistance (θJA), WQFN (1) 40°C/W (1) The maximum power dissipation is a function of TJ(MAX), θJA and TA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) – TA)/ θJA. All numbers apply for package soldered directly into a 2 layer PC board with zero air flow. Package should be soldered unto a 6.8 mm2 copper area as shown in the “recommended land pattern” shown in the package drawing. Electrical Characteristics (1) Unless otherwise specified, all limits are ensured for TA = 25°C, Input CM = 2.5V, VCM = 1.2V, VCM_Aux = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main & Auxiliary Outputs), both Main and Auxiliary Output Specifications, full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (2). Electrical Characteristics Definition of Terms and Specifications for abbreviations used in the datasheet. Boldface limits apply at the temperature extremes. (1) Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating of the device such that TJ = TA. No specification of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ > TA. (2) “Full Power” setting is with Auxiliary output turned on. 2 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Electrical Characteristics (1) (continued) Unless otherwise specified, all limits are ensured for TA = 25°C, Input CM = 2.5V, VCM = 1.2V, VCM_Aux = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main & Auxiliary Outputs), both Main and Auxiliary Output Specifications, full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (2). Electrical Characteristics Definition of Terms and Specifications for abbreviations used in the datasheet. Boldface limits apply at the temperature extremes. Symbol Parameter Condition Min(3) Typ(4) Max(3) Units Dynamic Performance LSBW −3 dB Bandwidth All Gains 900 MHz Peaking Peaking All Gains 1 dB GF_0.1 dB ±0.1 dB Gain Flatness All Gains 150 MHz GF_1 dB ±1 dB Gain Flatness All Gains 400 MHz TRS Rise Time 460 ps TRL Fall Time 450 OS Overshoot Main Output 9 % ts_1 Settling Time Main Output, ±0.5% 10 ns ts_2 Main Output, ±0.05% 14 t_recover Recovery Time(5) All Gains <5 ns PD Propagation Delay VOUT = 0.7 VPP, All Gains 1.2 ns PD_VAR Propagation Delay Variation Gain Varied 100 ps Noise, Distortion, and RF Specifications en_1 Input Noise Spectral Density Max Gain, 10 MHz 0.98 nV/√Hz en_2 Preamp LG and 0 dB Ladder, 4.1 10 MHz eno_1 RMS Output Noise Max Gain, 100 Hz to 400 MHz 1.7 mV eno_2 Preamp LG, 0 dB Ladder, 100 Hz 940 μV to 400 MHz NF_1 Noise Figure Max Gain, RS = 50Ω each Input, 3.8 10 MHz dB NF_2 Preamp LG, 0 dB Ladder, RS = 50Ω 13.5 each Input, 10 MHz HD2/ HD3_1 2nd/ 3rd Harmonic Distortion(6) Main Output, 100 MHz, All Gains −50/ −53 HD2/ HD3_2 Auxiliary Output, 100 MHz, All Gains −48/ −50 dBc HD2/ HD3_3 Main Output, 250 MHz, All Gains −44/ −50 HD2/ HD3_4 Auxiliary Output, 250 MHz, All Gains −42/ −42 IMD3 Intermodulation Distortion (6) f = 250 MHz, Main output −65 dBc OIP3_1 Intermodulation Intercept (6) Main Output, 250 MHz 26 dBm P_1dB_main −1 dB Compression Main Output, 250 MHz, 0 dB Ladder 1.8 Main Output, 250 MHz, 20 dB Ladder 1.0 P_1dB_aux Auxiliary Output, 250 MHz, 1.65 VPP 0 dB Ladder Auxiliary Output, 250 MHz, 1.0 20 dB Ladder Gain Parameters AV_DIFF_MAX Max Gain 38.1 38.8 39.5 dB AV_DIFF_MIN Min Gain −1.91 −1.16 −0.40 dB (3) Limits are 100% production tested at 25°C unless otherwise specified. Limits over the operating temperature range are ensured through correlation using Statistical Quality Control (SQC) methods. (4) Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will also depend on the application and configuration. The typical values are not tested and are not ensured on shipped production material. (5) Recovery time” is the slower of the Main and Auxiliary outputs. Output swing of 700 mVPP shifted up or down by 50% (0.35V) by introducing an offset. Measured values correspond to the time it takes to return to within ±1% of 0.7 VPP (±7 mV). (6) Distortion data taken under single ended input condition. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 3 Product Folder Links: LMH6518 LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Electrical Characteristics (1) (continued) Unless otherwise specified, all limits are ensured for TA = 25°C, Input CM = 2.5V, VCM = 1.2V, VCM_Aux = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main & Auxiliary Outputs), both Main and Auxiliary Output Specifications, full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (2). Electrical Characteristics Definition of Terms and Specifications for abbreviations used in the datasheet. Boldface limits apply at the temperature extremes. Symbol Parameter Condition Min(3) Typ(4) Max(3) Units Gain_Step Gain Step Size All Gains including Preamp Step 1.8 2 2.2 dB Gain Step Size with ADC (See ADC FS Adjusted 8.5 mdB Applications Information) Gain_Range Gain Range 39 40 41 dB TC_AV_DIFF Gain Temp Coefficient (7) All Gains −0.8 mdB/°C Gain_ACC Absolute Gain Accuracy Compared to theoretical from 0.75 — +0.75 dB Max Gain in 2 dB steps Matching Gain_match Gain Matching Main/Auxiliary All Gains ±0.1 ±0.2 dB BW_match −3 dB Bandwidth Matching All Gains 5 % Main/Auxiliary RT_match Rise Time Matching Main/ Auxiliary All Gains 5 % PD_match Propagation Delay Matching All Gains 100 ps Main/Auxiliary Analog I/O CMRR_1 CM Rejection Ratio (see Table 1) Preamp HG, 0 dB Ladder, 1.9V < 45 86 CMVR < 3.1V dB CMRR_2 Preamp LG, 0 dB Ladder, 1.9V < 40 55 CMVR < 3.1V CMVR_1 Input Common Mode Voltage Range Preamp HG, All Ladder Steps, CMRR 1.9 — 3.1 ≥ 45 dB V CMVR_2 Preamp LG, All Ladder Steps, CMRR 1.9 — 3.1 ≥ 40 dB |ΔVO_CM|ΔI_CM| All Gains, 2V < CMVR < 3V −60 −100 dB CMRR_CM CM Rejection Ratio relative to VCM (see Preamp LG, 0 dB 101 dB Table 1) Zin_diff Differential Input Impedance All Gains 150||1.5 Z KΩ || in_CM CM Input impedance Preamp HG 420||1.7 pF Preamp LG 900||1.7 FSOUT1 Full Scale Voltage Swing Main Output, THD @ 100 MHz ≤ 770(8) 800 −40 dBc, All Gains FSOUT2 Main Output, Clamped, 0 dB Ladder 1800 1960 FSOUT3 Auxiliary Output, THD @ 100 MHz ≤ 770(8) 800 mVPP −40 dBc All Gains FSOUT4 Auxiliary Output, Clamped,0 dB 1600 1760 Ladder VOUT_MAX1 Voltage range at each output pin Main Output, All gains, VCM = 1.2V 0.5 1.8 V (clamped) OUT_MAX2 Auxiliary Output, All Gains, 0.8 2.2 VCM = 1.2V V VOUT_MAX3 Main Output, All Gains, VCM = 1.45V 2.05 VOUT_MAX4 Auxiliary output, All gains, 2.45 VCM = 1.45V ZOUT_DIFF Differential Output Impedance All Gains 92 100 108 Ω VOOS Output Offset Voltage All Gains ±15 ±40 mV VOOS_shift1 Output Offset Voltage Shift Preamp LG to Preamp HG 13.7 mV VOOS_shift2 All Gains, Excluding Preamp Step 12.7 (7) Drift determined by dividing the change in parameter at temperature extremes by the total temperature change. (8) Specified by design. 4 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 vO_CM vOUT 250 MHz, 'VO_CM 'VOUT DC, LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Electrical Characteristics (1) (continued) Unless otherwise specified, all limits are ensured for TA = 25°C, Input CM = 2.5V, VCM = 1.2V, VCM_Aux = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main & Auxiliary Outputs), both Main and Auxiliary Output Specifications, full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (2). Electrical Characteristics Definition of Terms and Specifications for abbreviations used in the datasheet. Boldface limits apply at the temperature extremes. Symbol Parameter Condition Min(3) Typ(4) Max(3) Units TCVOOS Output Offset Voltage Drift(9) Preamp HG, 0 dB Ladder −24 μV/°C Preamp LG, 0 dB Ladder −7 IB Input Bias Current(10) +40 +100 +140 μA VOCM Output CM Voltage Range All Gains 0.95 1.20 1.45 V VOS_CM Output CM Offset Voltage All Gains ±15 ±30 mV TC_VOS_CM CM Offset Voltage Temp Coefficient All Gains +55 μV/°C BAL_Error_DC Output Gain Balance Error −78 dB BAL_Error_AC −45 PB Phase Balance Error (See Table 1) 250 MHz ±0.8 deg PSRR Differential Power Supply Rejection(see Preamp HG, 0 dB Ladder −60 −87 Table 1) dB Preamp LG, 0 dB Ladder −50 −70 PSRR_CM CM Power Supply Rejection(see Preamp LG, 0 dB −55 −71 dB Table 1) VCM_I VCM Input Bias Current(11) All Gains ±1 ±10 ±20 nA VCM_AUX_I VCM_AUX Input Bias Current(11) All Gains ±1 ±10 ±20 (9) Drift determined by dividing the change in parameter at temperature extremes by the total temperature change. (10) Positive current is current flowing into the device. (11) Positive current is current flowing into the device. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 5 Product Folder Links: LMH6518 LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Electrical Characteristics (1) (continued) Unless otherwise specified, all limits are ensured for TA = 25°C, Input CM = 2.5V, VCM = 1.2V, VCM_Aux = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main & Auxiliary Outputs), both Main and Auxiliary Output Specifications, full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (2). Electrical Characteristics Definition of Terms and Specifications for abbreviations used in the datasheet. Boldface limits apply at the temperature extremes. Symbol Parameter Condition Min(3) Typ(4) Max(3) Units Digital I/O & Timing VIH Input Logic High VDD-0.6 V VIL Input Logic Low 0.5 V VOH Output Logic High VDD V VOL Output Logic Low 0 V RHi_Z Output Resistance High Impedance Mode 5 MΩ I_in Input Bias Current <1 μA FSCLK SCLK Rate 10 MHz FSCLK_DT SCLK Duty Cyle 45 50 55 % TS SDIO Setup Time 25 ns TH SDIO Hold Time 25 ns TCES CS Enable Setup Time From CS asserted to rising edge of 25 ns SCLK tCDS CS Disable Setup Time From CS de-asserted to rising edge 25 ns of SCLK TIAG Inter-Acess Gap 3 Cycles of SCLK Power Requirements IS1 Supply Current VCC 195 210 225 230 mA IS1_off VCC Aux off 150 165 170 IDD VDD 180 350 400 μA Bandwidth Limiting Filter Specifications Filter Parameter Condition Min Typ Max Units 20 MHz Pass Band Tolerance (All Gains) −3 dB Bandwidth −0, +20 % 100 MHz Pass Band Tolerance (All Gains) −3 dB Bandwidth −0, +20 % 200 MHz Pass Band Tolerance (All Gains) −3 dB Bandwidth −0, +20 % 350 MHz Pass Band Tolerance (Preamp LG, 0 dB −3 dB Bandwidth ±10 Ladder) % Pass Band Tolerance (All Gains) ±25 650 MHz Pass Band Tolerance (Preamp LG, 0 dB −3 dB Bandwidth ±10 Ladder) % Pass Band Tolerance (All Gains) ±25 750 MHz Pass Band Tolerance (Preamp LG, 0 dB −3 dB Bandwidth ±10 Ladder) % Pass Band Tolerance (All Gains) ±25 6 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 'VO_CM 'VOUT LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Table 1. Definition of Terms and Specifications 1. AV_CM (dB) Change in output offset voltage (ΔVOOS) with respect to the change in input common mode voltage (ΔVI_CM) 2. AV_DIFF (dB) Gain with 100Ω differential load 3. CM Common Mode 4. CMRR (dB) Common Mode rejection defined as: AV_DIFF (dB) - AV_CM (dB) 5. CMRR_CM Common Mode rejection relative to VCM defined as: ΔVOOS /ΔVCM 6. HG Preamp High Gain 7. Ladder Ladder Attenuator setting (0-20 dB) 8. LG Preamp Low Gain 9. Max Gain Gain = 38.8 dB 10. Min Gain Gain = −1.16 dB 11. +Out Positive Main Output 12. −Out Negative Main Output 13. +Out Aux Positive Auxiliary Output 14. −Out Aux Negative Auxiliary Output 15. PB Phase Balance defined as the phase difference between the complimentary outputs relative to 180° 16. PSRR Input referred VOOS shift divided by change in VCC 17. PSRR_CM Output common mode voltage change (ΔVO_CM) with respect to VCC voltage change (ΔVCC) 18. VCM Input pin voltage that sets Main output CM 19. VCM_Aux Input pin voltage that sets Auxiliary output CM 20. VI_CM Input CM voltage (average of +IN and −IN) 21. ΔVIN (V) Differential voltage across device inputs 22. VOOS DC offset voltage. Differential output voltage measured with inputs shorted together to VCC/2 23. VO_CM Output common mode voltage (DC average of V+OUT and V−OUT) 24. VOS_CM CM offset voltage: VO_CM - VCM 25. ΔVO_CM Variation in output common mode voltage (VO_CM) 26. Balance Error. Measure of the output swing balance of “+OUT” and “−OUT”, as reflected on the output common mode voltage (VO_CM), relative to the differential output swing (VOUT). Calculated as output common mode voltage change (ΔVO_CM) divided by the output differential voltage change (ΔVOUT, which is nominally around 700 mVPP) 27. ΔVOUT Change in differential output voltage (Corrected for DC offset (VOOS)) Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 7 Product Folder Links: LMH6518 -OUT AUX 16 15 14 13 10 11 12 8 7 6 5 4 3 2 1 CS SDIO SCLK VDD VCM -OUT +OUT VCM_AUX +OUT AUX VCC VCC GND +IN -IN GND 9 LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com PIN OUT Pin Out Function P1 = +OUT Aux Auxiliary positive output P2 = −OUT Aux Auxiliary negative output P3 = VCC (5V) Analog power supply P4 = VCC (5V) Analog power supply P5 = GND Ground, electrically connected to the WQFN heat sink P6 = +IN Positive Input P7 = −IN Negative Input P8 = GND Ground, electrically connected to the WQFN heat sink P9 = CS SPI interface, Chip Select, Active low P10 = SDIO SPI interface, Serial Data Input/Output P11 = SCLK SPI interface, Clock P12 = VDD (3.3V) Digital power supply P13 = VCM Input from ADC to control main output CM P14 = −OUT Main negative output P15 = +OUT Main positive output P16 = VCM_Aux Input to control auxiliary output CM Connection Diagram Figure 1. 16-Pin-Top View See Package Number RGH0016A 8 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 1 10 100 1G FREQUENCY (MHz) -25 -20 -15 -10 -5 0 5 NORMALIZED GAIN (dB) Full BW 750 MHz 650 MHz 350 MHz 200 MHz 100 MHz 20 MHz Response (HG, 0 dB) VOUT = 0.1 VPP 10 100 1G FREQUENCY (MHz) -6 -5 -4 -3 -2 -1 0 1 2 3 NORMALIZED GAIN (dB) Full BW HG, 0 dB LG, 0 dB HG, 20 dB LG, 20 dB 1 10 100 1G FREQUENCY (MHz) -25 -20 -15 -10 -5 0 5 NORMALIZED GAIN (dB) Full BW 750 MHz 650 MHz 350 MHz 200 MHz 100 MHz 20 MHz Response (HG, 0 dB) 1 10 100 1G FREQUENCY (MHz) -25 -20 -15 -10 -5 0 5 NORMALIZED GAIN (dB) Full BW 750 MHz 650 MHz 350 MHz 200 MHz 100 MHz 20 MHz Response (LG, 0 dB) VOUT = 0.1 VPP 1 10 100 1G FREQUENCY (MHz) -25 -20 -15 -10 -5 0 5 NORMALIZED GAIN (dB) Full BW 750 MHz 650 MHz 350 MHz 200 MHz 100 MHz 20 MHz Response (LG, 0 dB) 1 10 100 1G FREQUENCY (MHz) -300 -250 -200 -150 -100 -50 0 PHASE (°) Full BW 20 MHz 750 MHz 100 MHz 200 MHz 350 MHz 650 MHz Phase (LG, 0 dB) LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Typical Performance Characteristics Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). Response (LG, 0 dB) Phase (LG, 0 dB) Figure 2. Figure 3. Response (HG, 0 dB) Small Signal Response (LG, 0 dB) Figure 4. Figure 5. Response vs. Small Signal Response (HG, 0 dB) Gain Figure 6. Figure 7. (1) “Full Power” setting is with Auxiliary output turned on. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 9 Product Folder Links: LMH6518 0 2 4 6 8 10 12 14 16 18 20 -3 -2.5 -2 -1.5 -1 -0.5 0 0.5 NORMALIZED GAIN (dB) FREQUENCY (MHz) All Gains 20 MHz Filter 0 50 100 150 200 250 300 FREQUENCY (MHz) -180 -130 -80 -30 20 PHASE (°) HG, 0 dB, 10 dB, 20 dB LG, 0 dB, 10 dB, 20 dB 20 MHz Filter 10 100 1G FREQUENCY (MHz) -6 -5 -4 -3 -2 -1 0 1 2 NORMALIZED GAIN (dB) 85°C, HG 25°C, HG 85°C, LG 25°C, LG -40°C, HG -40°C, LG 10 dB Ladder 10 100 1G FREQUENCY (MHz) -6 -5 -4 -3 -2 -1 0 1 2 NORMALIZED GAIN (dB) -350 -300 -250 -200 -150 -100 -50 0 50 PHASE (°) Phase Gain Aux, HG Main, LG Aux, HG Main, HG Main, LG Aux, LG 10 dB Ladder 0 200 400 600 800 1000 -300 -250 -200 -150 -100 -50 0 PHASE (°) FREQUENCY (MHz) Full BW LG, 0 dB LG, 20 dB HG, 0 dB HG, 20 dB 10 100 1G FREQUENCY (MHz) -6 -5 -4 -3 -2 -1 0 1 2 NORMALIZED GAIN (dB) 85°C, HG 85°C, LG 25°C, HG 25°C, LG -40°C, HG -40°C, LG 10 dB Ladder LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Typical Performance Characteristics (continued) Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). Phase vs. Gain Response Over Temperature Figure 8. Figure 9. Main vs. Auxiliary Response Over Temperature Auxiliary Response Figure 10. Figure 11. Response Phase vs. vs. Gain Gain Figure 12. Figure 13. 10 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 0 2 4 6 8 10 12 14 16 18 20 0.5 0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3 2.5 LADDER ATTENUATION (dB) INPUT REFERRED (nV/ Hz) 0 20 40 60 80 100 OUTPUT REFERRED (nV/ Hz) Input Referred Output Referred f = 10 MHz Preamp HG INPUT REFERRED (nV/ Hz) OUTPUT REFERRED (nV/ Hz) 0 2 4 6 8 10 12 14 16 18 20 0 2 4 6 8 10 12 14 16 18 LADDER ATTENUATION (dB) 0 5 10 15 20 25 30 35 40 45 Input Referred Output Referred f = 10 MHz Preamp LG 1 10 100 1000 FREQUENCY (MHz) -15 -10 -5 0 5 10 15 PHASE (°) 0 3 6 9 12 15 18 GROUP DELAY (ns) Linear Phase Deviation Group Delay 1 10 100 1G FREQUENCY (MHz) -90 -80 -70 -60 -50 -40 -30 -20 -10 GAIN (dB) Phase Gain LG, 0 dB -7 -6 -5 -4 -3 -2 -1 0 1 PHASE (°) 10 100 1G -6 -5 -4 -3 -2 -1 0 1 NORMALIZED GAIN (dB) FREQUENCY (MHz) 650 MHz Filter LG, 0 dB HG, 20 dB HG, 0 dB LG, 20 dB 0 200 400 600 800 1000 -300 -250 -200 -150 -100 -50 0 PHASE (°) FREQUENCY (MHz) 650 MHz Filter LG, 0 dB LG, 20 dB HG, 0 dB HG, 20 dB LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Typical Performance Characteristics (continued) Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). Response Phase vs. vs. Gain Gain Figure 14. Figure 15. Balance Error Linear Phase Deviation and Group Delay Figure 16. Figure 17. Noise Noise vs. vs. Ladder Attenuation Ladder Attenuation Figure 18. Figure 19. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 11 Product Folder Links: LMH6518 0 4 8 12 16 20 -30 -35 -40 -45 -50 -55 -60 -65 -70 HD (dBc) LADDER ATTENUATION (dB) 500 MHz 10 MHz 100 MHz 20 MHz 250 MHz 50 MHz HG 0 4 8 12 16 20 -50 -55 -60 -65 -70 -75 -80 -85 HD (dBc) LADDER ATTENUATION (dB) LG 10 MHz 20 MHz 50 MHz 100 MHz 250 MHz 500 MHz 1 100 10k 1M FREQUENCY (kHz) 1 10 100 10 1k 100k CURRENT NOISE (pA/ Hz) LG HG 0 4 8 12 16 20 -30 -35 -40 -45 -50 -55 -60 -65 -70 -75 HD (dBc) LADDER ATTENUATION (dB) 10 MHz 20 MHz 500 MHz 100 MHz 250 MHz 50 MHz LG 0 5 10 15 20 0 2 4 8 6 10 12 14 16 18 20 22 24 26 28 LADDER ATTENUATION (dB) NOISE FIGURE (dB) f = 10 MHz RS = 50: on each input Preamp HG Preamp LG 1 100 10k 1M FREQUENCY (kHz) 0 1 10 1000 10 1k 100k 100 VOLTAGE NOISE (nV/ Hz) HG, 20 dB LG, 0 dB HG, 0 dB LG, 20 dB LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Typical Performance Characteristics (continued) Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). Noise Figure Input Voltage Noise vs. vs. Gain Frequency Figure 20. Figure 21. Input Current Noise HD2 vs. vs. Frequency Ladder Attenuation Figure 22. Figure 23. HD3 HD2 vs. vs. Ladder Attenuation Ladder Attenuation Figure 24. Figure 25. 12 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 LADDER ATTENUATION (dB) 0 4 8 12 16 -2 2 6 10 14 18 42 GAIN (dB) 20 22 26 30 34 38 HG -40°C to 85°C LG -40°C HG 0 4 8 12 16 20 -0.15 -0.1 -0.05 0 0.05 0.1 0.15 0.2 0.25 GAIN ACCURACY (dB) LADDER ATTENUATION (dB) Relative to HG/0 dB @ 25°C LG -40°C 25°C 85°C 25°C 85°C -7 -6 -5 -4 -3 -2 -1 0 -65 -67 -69 -71 -73 -75 -77 -79 -81 -83 -85 HARMONIC DISTORTION (dBc) OUTPUT POWER (dBFS) HG, 10 dB 65 MHz HD2 HD3 0 4 8 12 16 20 -50 -55 -60 -65 -70 -75 -80 -85 -90 -95 HARMONIC DISTORTION (dBc) LADDER ATTENUATION (dB) LG, 65 MHz Aux HD3 HD2 Aux Main 0 4 8 12 16 20 -50 -55 -60 -65 -70 -75 -80 -85 HARMONIC DISTORTION (dBc) LADDER ATTENUATION (dB) HG, 65 MHz HD3 HD2 Main Aux Aux 0 4 8 12 16 20 -50 -55 -60 -65 -70 -75 HD (dBc) LADDER ATTENUATION (dB) 10 MHz 20 MHz 50 MHz 100 MHz 250 MHz 500 MHz HG LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Typical Performance Characteristics (continued) Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). HD3 vs. Ladder Attenuation Main and Auxiliary Distortion Comparison Figure 26. Figure 27. Distortion vs. Main and Auxiliary Distortion Comparison Output Power Figure 28. Figure 29. Gain Gain Accuracy vs. vs. Ladder Attenuation Ladder Attenuation Figure 30. Figure 31. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 13 Product Folder Links: LMH6518 1.5 5 2.5 3 3.5 -50 -40 -30 -20 -10 0 10 20 VOOS (mV) VI_CM LG, 0 dB LG, 20 dB HG, 0 dB HG, 20 dB 85°C 0 4 8 12 16 20 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 VOUT (VPP) LADDER ATTENUATION (dB) LG HG Aux, HG Aux, LG f = 250 MHz RL = 100 1.5 5 2.5 3 3.5 -50 -40 -30 -20 -10 0 10 20 VOOS (mV) VI_CM LG, 0 dB LG, 20 dB HG, 0 dB HG, 20 dB -40°C 1.5 5 2.5 3 3.5 -50 -40 -30 -20 -10 0 10 20 VOOS (mV) VI_CM LG, 0 dB LG, 20 dB HG, 0 dB HG, 20 dB 25°C -40°C 0 4 8 12 16 20 -0.15 -0.1 -0.05 0 0.05 0.1 0.15 0.2 0.25 GAIN ACCURACY (dB) LADDER ATTENUATION (dB) Relative to HG/0 dB @ 25°C LG -40°C 25°C 85°C 25°C 85°C HG 0 4 8 12 16 20 -0.094 -0.093 -0.092 -0.091 -0.09 -0.089 -0.088 GAIN (dB) LADDER ATTENUATION (dB) HG LG Aux Gain ± Main Gain LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Typical Performance Characteristics (continued) Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). Auxiliary Gain Accuracy Gain Matching vs. vs. Ladder Attenuation Ladder Attenuation Figure 32. Figure 33. AV_CM AV_CM Figure 34. Figure 35. −1 dB Compression vs. AV_CM Ladder Attenuation Figure 36. Figure 37. 14 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 HG -40°C LG 0 4 8 12 16 20 -20 -15 -10 -5 0 5 10 15 20 VOOS (mV) LADDER ATTENUATION (dB) 85°C 25°C -40°C LG 0 4 8 12 16 20 -20 -15 -10 -5 0 5 10 15 20 VOOS (mV) LADDER ATTENUATION (dB) 85°C 25°C HG TIME (1 ns/DIV) -0.4 -0.3 -0.2 -0.1 0 0.1 0.2 0.3 0.4 VOUT (V) HI to LO LO to HI HI to LO LO to HI Input = 0.2V/DIV LG, 20 dB Output Input TIME (1 ns/DIV) -0.4 -0.3 -0.2 -0.1 0 0.1 0.2 0.3 0.4 VOUT (V) HI to LO LO to HI HI to LO LO to HI Input = 2 mV/DIV HG, 0 dB Output Input TIME (1 ns/DIV) -0.4 -0.3 -0.2 -0.1 0 0.1 0.2 0.3 0.4 VOUT (V) HI to LO HI to LO LO to HI LO to HI Input = 20 mV/DIV LG, 0 dB Output Input TIME (1 ns/DIV) -0.4 -0.3 -0.2 -0.1 0 0.1 0.2 0.3 0.4 VOUT (V) HI to LO LO to HI HI to LO LO to HI Input = 20 mV/DIV HG, 20 dB Output Input LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Typical Performance Characteristics (continued) Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). Step Response Step Response Figure 38. Figure 39. Step Response Step Response Figure 40. Figure 41. Output Offset Voltage (Typical Unit 1) Output Offset Voltage (Typical Unit 2) Figure 42. Figure 43. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 15 Product Folder Links: LMH6518 85°C 1.5 2 2.5 3 3.5 0 0.02 0.04 0.06 0.08 0.1 0.12 0.14 0.16 0.18 0.2 INPUT BIAS CURRENT (mA) VI_CM (V) 25°C -40°C 85°C 4.5 4.7 4.9 5.1 5.3 5.5 1000 1200 1400 1600 1800 2000 2200 2400 AUXILIARY VOLTAGE (mV) VCC (V) 25°C -40°C RL = 100: VCM_Aux = 1.2V No CM Load +OUT Aux and -OUT Aux 4.5 4.7 4.9 5.1 5.3 5.5 195 200 205 210 215 220 225 ICC (mA) VCC (V) -40°C 85°C 25°C 2.8 2.9 3 3.1 3.2 3.3 3.4 3.5 3.6 VDD (V) 0.1 0.12 0.14 0.16 0.18 0.2 0.22 IDD (mA) 85°C 25°C -40°C 85°C -40°C LG 0 4 8 12 16 20 -20 -15 -10 -5 0 5 10 15 20 VOOS (mV) LADDER ATTENUATION (dB) 25°C HG 0.7 0.9 1.1 1.3 1.5 1.7 -15 -13 -11 -9 -7 -5 -3 -1 1 3 5 VOS_CM (mV) VCM (V) 85°C 25°C -40°C LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Typical Performance Characteristics (continued) Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). VOS_CM vs. Output Offset Voltage (Typical Unit 3) VCM Figure 44. Figure 45. Supply Current Supply Current vs. vs. Supply Voltage Supply Voltage Figure 46. Figure 47. Input Bias Current vs. Input CM Auxiliary Output Voltage (Hi-Z Mode) Figure 48. Figure 49. 16 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 0 100 200 300 400 500 TIME (ns) -1.5 -1.0 -0.5 0 0.5 1.0 1.5 ERROR (%) 0 dB Ladder Attenuation 50% Overdrive Preamp HG or LG -10 -6 -2 2 6 10 600 800 1000 1200 1400 1600 1800 OUTPUT VOLTAGE (V) DELTA-VIN (mV) +OUT -OUT -40°C 85°C 25°C -40°C 25°C, 85°C Main or Auxiliary Output HG, 0 dB -100 -60 -20 20 60 100 800 900 1000 1100 1200 1300 1400 1500 1600 OUTPUT VOLTAGE (V) DELTA-VIN (mV) +OUT -OUT -40°C 25°C, 85°C 25°C, 85°C -40°C Main or Auxiliary Output HG, 20 dB -100 -60 -20 20 60 100 600 800 1000 1200 1400 1600 1800 OUTPUT VOLTAGE (V) DELTA-VIN (mV) +OUT -OUT -40°C 85°C 25°C -40°C 25°C, 85°C Main or Auxiliary Output LG, 0 dB -1 -0.6 -0.2 0.2 0.6 1 800 900 1000 1100 1200 1300 1400 1500 1600 OUTPUT VOLTAGE (V) DELTA-VIN (V) +OUT 25°C, 85°C -OUT -40°C 25°C, 85°C -40°C Main or Auxiliary Output LG, 20 dB -5 0 5 10 15 20 25 30 35 40 -10 -5 0 5 10 15 20 25 ERROR from NOMINAL FILTER BW (%) GAIN (dB) LG HG 350 MHz 650 MHz 750 MHz 750 MHz 350 MHz 650 MHz LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Typical Performance Characteristics (continued) Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). Filter BW Output vs. vs. Gain Input Figure 50. Figure 51. Output Output vs. vs. Input Input Figure 52. Figure 53. Output vs. Input Overdrive Recovery Time (Return to Zero) Figure 54. Figure 55. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 17 Product Folder Links: LMH6518 0 100 200 300 400 500 TIME (ns) -1.5 -1.0 -0.5 0 0.5 1.0 1.5 ERROR (%) 20 dB Ladder Attenuation 50% Overdrive Preamp HG or LG LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Typical Performance Characteristics (continued) Unless otherwise specified, Input CM = 2.5V, VCM = 1.2V, VCM AUX = 1.2V, Single-ended input drive, VCC = 5V, VDD = 3.3V, RL = 100Ω differential (both Main & Auxiliary Outputs), VOUT = 0.7 VPP differential (both Main and Auxiliary Outputs), Main output specification (Auxiliary is labeled “Auxiliary”), full bandwidth setting, gain = 18.8 dB (Preamp LG, 0 dB ladder attenuation), Full Power setting (1). Overdrive Recovery Time (Return to Zero) Figure 56. 18 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 ) to +1.41 dB 700 mV (= 20 x log 805 mV ) 700 mV (= 20 x log 595 mV = 8.5 mdB 0.56 ± 2 x 512 0.84 ± 0.56 0.56 + 2 x 512 0.84 ± 0.56 ¨ ¨ © § ¨ ¨ © § ¨ ¨ © § ¨ ¨ © § Gain Resolution = 20 log = 42.6 dB) 920 mVPP (20 x log 6.8 mVPP Ladder Attenuator 10 Steps, 2 dB/ Step 0 to -20 dB Pre-amp 10 dB or 30 dB Output Amp 8.86 dB +Out -Out +In -In 50: 50: LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 APPLICATIONS INFORMATION FUNCTIONAL DESCRIPTION AND DYNAMIC RANGE IN OSCILLOSCOPE APPLICATIONS Here is a block diagram of the LMH6518’s Main Output signal path: Figure 57. LMH6518 Signal Path Block Diagram The Auxiliary output (not shown) uses another but similar Output Amp that taps into the Ladder Attenuator output. In this document, Preamp gain of 30 dB is referred to as “Preamp HG” (High Gain) and Preamp gain of 10 dB as “Preamp LG” (Low Gain). The LMH6518’s 2 dB/step gain resolution and 40 dB adjustment range (from −1.16 dB to 38.8 dB) allows this device to be used with the TI GSample/second ADCs which have Full Scale, FS, adjustment (through their Extended Control Mode or ECM) to provide near-continuous variability (8.5 mdB resolution) to cover a 42.6 dB (1) FS input range. The Texas Instruments GSample/second ECM control allows the ADC FS to be set using the ADC SPI bus. The ADC FS voltage range is from 560 mV to 840 mV with 9 bits of FS voltage control. The ADC ECM gain resolution can be calculated as follows: (2) The recommended ADC FS operating range is, however, narrower and it is from 595 mV to 805 mV with 700 mVPP as the mid-point. Raising the value of ADC FS voltage is tantamount to reducing the signal path gain to accommodate a larger input and vice versa, thus providing a method of gain fine-adjust. The ADC ECM gain adjustment is −1.21 dB (3) Because the ADC FS fine-adjust range of 2.62 dB (= 1.41 dB + 1.21 dB) is larger than the LMH6518’s 2 dB/step resolution, there is always at least one LMH6518 gain setting to accommodate any FS signal from 6.8 mVPP to 920 mVPP, at the LMH6518 input, with 0.62 dB (= 2.62-2) overlap. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 19 Product Folder Links: LMH6518 (20 x log = 3.5 dB) 200 MHz 450 MHz Attenuation (dB) = 20 x log FSMAX (VPP) 800 mVPP Maximum LMH6518 FS Input = 0.7 VPP 10 (-1.16 ± 1.21) dB 20 = 920 mVPP Minimum LMH6518 FS Input = 0.7 VPP 10 (38.8 + 1.41) dB 20 = 6.8 mVPP LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Assuming a nominal 0.7VPP output, the LMH6518’s minimum FS input swing is limited by the maximum signal path gain possible and vice versa: (4) (or 8 mVPP with no ADC fine adjust) (5) (or 800 mVPP with no ADC FS adjust) To accommodate a higher FS input, an additional attenuator is needed before the LMH6518. This front-end attenuator is shown in the Figure 62 with its details shown in Figure 71. The highest minimum attenuation level is determined by the largest FS input signal (FSmax): (6) So, to accommodate 80 VPP, 40 dB minimum attenuation is needed before the LMH6518. In a typical oscilloscope application, the voltage range encountered is from 1 mV/DIV to 10 V/DIV with 8 vertical divisions visible on the screen. One of the primary concerns in a digital oscilloscope is SNR which translates to display trace width/ thickness. Typically, oscilloscope manufacturers need the noise level to be low enough so that the “no-input” visible trace width is less than 1% of FS. Experience has shown that this corresponds to a minimum SNR of 52 dB. The factors that influence SNR are: • Scope front end noise (Front-end attenuator + scope probe Hi-Z buffer which is discussed later in this document and shown in Figure 62) • LMH6518 • ADC LMH6518 related SNR factors are: • Bandwidth • Preamp used (Preamp High Gain or Low Gain) • Ladder Attenuation • Signal level SNR increases with the inverse square root of the bandwidth. So, reducing bandwidth from 450 MHz to 200 MHz, for example, improves SNR by 3.5 dB (7) The other factors listed above, preamp and ladder attenuation, depend on the signal level and also impact SNR. The combined effect of these factors is summarized in Figure 58 where SNR is plotted as a function of the LMH6518 FS input voltage (assuming scope bandwidth of 200 MHz) and not including the ADC and the front end noise: 20 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 100x (= ) 80 VPP 0.8 VPP ) 0.8 VPP 24 mVPP (= 20 x log 0.001 1 INPUT FS (V) 38 46 54 62 SNR (dBFS) 0.01 0.1 58 50 42 40 44 48 52 56 60 6 14 18 10 2 0 4 8 12 16 20 LADDER ATTENUATION (dB) Preamp HG Preamp LG 200 MHz Filter 22 -2 LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Figure 58. LMH6518 SNR & Ladder Attenuation used vs. Input As can be seen from Figure 58, SNR of at least 52 dB is maintained for FS inputs above 24 mVPP (3 mV/DIV on a scope) assuming the LMH6518’s internal 200 MHz filter is enabled. Most oscilloscope manufacturers relax the SNR specifications to 40 dB for the highest gain (lowest scope voltage setting). From Figure 58, LMH6518’s minimum SNR is 43.5 dB, thereby meeting the relaxed SNR specification for the lower range of scope front panel voltages. In Figure 58, the step-change in SNR near Input FS of 90 mVPP is the transition point from Preamp LG to Preamp HG with a subsequent 3 dB difference due to the Preamp HG/ 20 dB ladder attenuation’s lower output noise compared to Preamp LG/ 2 dB ladder attenuation’s noise. Judicious choice of front end attenuators can ensure that the 52 dB SNR specification is maintained for scope FS inputs ≥ 24 mVPP by confining the LMH6518 gain range to the lower 30.5 dB (8) from the total range of 40 dB (= 38.8 - (−1.16)) possible. Here is an example: To cover the range of 1 mV/DIV to 10 V/DIV (80 dB range), here is a configuration which affords good SNR: Table 2. Oscilloscope Example Including Front-End Attenuators Row Scope FS Input “S”, Scope Vertical Preamp Ladder Attenuation “A”, Front-end Minimum SNR (dB) (VPP) Scale (V/DIV) Range (dB) attenuation (V/V) with 200 MHz filter 1 8m-24m 1m-3m HG 0-10 1 44 2 24m-80m 3m-10m HG 10-20 1 52.0 3 80m-0.8 10m-0.1 LG 0-20 1 53.4 4 0.8-8 0.1-1 LG 0-20 10 53.4 5 8-80 1-10 LG 0-20 100 53.4 In Table 2, the highest FS input in Row 5, Column 2 (80 VPP), and the LMH6518’s highest FS input allowed (0.8 VPP) set the (9) front-end attenuator value. The 100x attenuator will allow high SNR operation to 30.5 dB down, as explained earlier, or 2.4 VPP at scope input. In that same table, Rows 1-3 with no front-end attenuation (1x) cover the scope FS input range from 8 mVPP-800 mVPP. That leaves the scope FS input range of 0.8 VPP-2.4 VPP. If the 100x attenuator were used for the entire scope FS range of 0.8 VPP-80 VPP, SNR would dip below 52 dB for a portion of that range. Another attenuation level is thus required to maintain the SNR specification requirement of 52 dB. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 21 Product Folder Links: LMH6518 x 1.05 x 1020 G FSE = S x 8 A 10 110 mV oK = -21.6 + 20 x log = 17.57 dB K = 20 x log = 0.95 x 700 mVPP 8 x S(V/div) A -21.6 + 20 x log A S(V/div) LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com One possible attenuation partitioning is to select the additional attenuator value to cover a 20 dB range above 0.8 VPP FS (to 8 VPP) with the 100x attenuator covering the remaining 20 dB range from 8 VPP to 80 VPP. Mapping 8 VPP FS scope input to 0.8 VPP at LMH6518 input means the additional attenuator is 10x, as shown in Table 2, Row 4. The remaining scope input range of 8 VPP-80 VPP would then be covered by the 100x front-end attenuator derived earlier. The entire scope input range is now covered with SNR maintained about 52 dB for scope FS input ≥ 24 mVPP, as shown in Table 2. SETTINGS AND ADC SPI CODE (ECM) Covering the range from 1 mV/DIV to 10 V/DIV requires the following to be adjusted within the digital oscilloscope: • Front-End Attenuator • LMH6518 Preamp • LMH6518 Ladder Attenuation • ADC FS Value (ECM) The LMH6518 Product Folder contains a spreadsheet which allows one to calculate the front-end attenuator, LMH6518 Preamp gain (HG or LG) and ladder attenuation, and ADC FS setting based on the scope vertical scale (S in V/DIV). Here is the step by step procedure that explains the operations performed by the said spreadsheet based on the scope vertical scale setting (S in V/div) and front-end attenuation “A” (from Table 2). A numerical example is also worked out for more clarification: 1. Determine the required signal path gain, K: (10) assuming the full scale signal occupies 95% of the 0.7 VPP FS (for 5% overhead) which occupies 8 vertical scope divisions). Required condition: −2.37 dB ≤ K ≤ 40.3 dB Example: With S = 110 mV/DIV, Table 2 shows that A = 10 V/V: (11) 2. Determine the LMH6518 gain, G: – G is the closest LMH6518 gain, to the value of K where: – G = (38.8 – 2n)dB; n = 0, 1, 2, …, 20 – For this example, the closest G to K = 17.57 dB is 16.8 dB (with n = 11). The next LMH6518 gain, 18.8 dB (with n = 10) would be incorrect as 16.8 is closer. If 18.8 dB were mistakenly chosen, the ADC FS setting would be out of range. – Therefore: G = 16.8 dB 3. Determine Preamp (HG or LG) & Ladder Attenuation: – If G ≥ 18.8 dB → Preamp is HG and Ladder Attenuation = 38.8 - G – If G < 18.8 dB → Preamp is LG and Ladder Attenuation = 18.8 - G – For this example, with G = 16.8 → Preamp LG and Ladder Attenuation = 2 dB (= 18.8-16.8). 4. Determine the required ADC FS voltage, FSE: (12) 22 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 ECM (ratio) = 0.6393 - 0.56 0.28 = 0.283 ECM (ratio) = FSE - 0.56 0.28 16.8 FSE = = 639.3 mV S x 8 10 x 1.05 x 1020 LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 The “1.05” factor is to add 5% FS overhead margin to avoid ADC overdrive. (13) Required condition: 0.56V ≤ FSE ≤ 0.84V Recommend condition: 0.595V ≤ FSE ≤ 0.805V for optimum ADC FS 5. Determine the ADC ECM code ratio: where • 0.28V= (0.84-0.56)V • 0.56V is the lower end of the ADC FS adjustability • For this example: (14) – Required condition: 0 ≤ ECM (ratio) ≤ 1 6. Determine the ECM binary code to be sent on ADC SPI bus: – Convert the ECM value represented by the ratio calculated above, to binary: – ECM (binary) = DEC2BIN{ECM(ratio)* 511, 9} – Where “DEC2BIN” is a spreadsheet function which converts the decimal ECM ratio, from step 5 above, multiplied by 511 distinct levels, into binary 9 bits. NOTE The Web based spreadsheet computes ECM without the use of “DEC2BIN” function to ease usage by all spreadsheet users who may not have this function installed. – For this example: ECM (binary) = DEC2BIN(0.283*511, 9) = 010010000. This would be the number to be sent to the ADC on the SPI bus to program the ADC to the proper FS voltage. INPUT/OUTPUT CONSIDERATIONS The LMH6518’s ideal Input/Output Conditions, considered individually, are listed below: Table 3. LMH6518's Ideal Input/Output Conditions Impedance from Common Mode Differential Input Load Impedance (Ω) Differential Output Common Mode each input to Input (V) (VPP) (V) Output (V) ground (Ω) ≤50 1.5 to 3.1 <0.8 100 (differential)/ 50 <0.77 0.95-1.45 (single ended) In addition to the individual conditions listed in Table 3, the Input/Output terminal conditions should match differentially (i.e. +IN to −IN and +OUT to −OUT), as well, for best performance. The input is differential but can be driven single-ended as long as the conditions of Table 3 are met and there is good matching between the driven and the undriven inputs from DC to the highest frequency of interest. If not, there could be a settling time impact among other possible performance degradations. The datasheet specifications are with single-ended input, unless specified. Here is the recommended bench-test schematic to drive one input and to bias the other input with good matching in mind: Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 23 Product Folder Links: LMH6518 LMH6518 -IN +IN R 63.4 2 W R 82.5 3 W J1 Input (from 50W source) (Ground Referenced) +5V R 76.8 5 W R 76.8 4 W R 63.4 1 W V (-3.3V) EE LMH6518 -IN +IN C1 1 nF R1 100: R3 24.9: R2 R5 49.9: 200: R4 200: J1 Input (from 50: source) (2.5V CM) +5V C2 1 nF LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Figure 59. Recommended Single-Ended Bench-Test Input Drive from 50Ω Source With the schematic of Figure 59, each LMH6518 input sees 25Ω to ground at the higher frequencies when the capacitors look like shorts. This impedance increases to 125Ω at DC for both inputs, thereby preserving the required matching at any frequency. This configuration, using properly selected R’s and C’s, allows four times less biasing power dissipation than when the undriven input is biased with an effective 25Ω from the LMH6518 input to ground. It is possible to drive the LMH6518 input from a ground referenced 50Ω source by providing level shift circuitry on the driven input. Figure 60 shows a circuit where ½ the input signal reaches the LMH6518 input while the negative supply voltage (VEE) ensures that the 50Ω source at J1 does not experience any biasing current while providing 50Ω termination to the source. The driven input (+IN) is biased to 2.5V (VCC/2): Figure 60. LMH6518 Driven by a Ground Referenced Source In the schematic of Figure 60, the equivalent impedance from each LMH6518 input to ground is around 38Ω. This configuration’s power consumption of ∼0.5W (in R1 - R5) is higher than that of Figure 59 because of additional power dissipated to perform the level shifting. Additional 50Ω attenuators can be placed between J1 and R2/R3 junction in Figure 60 in order to accommodate higher input voltages. It is also possible to shift the LMH6518 output common mode level using a level shift approach similar to that of Figure 60. The circuit in Figure 61 shows an implementation where the LMH6518’s nominal 1.2V CM output, set by a 1.2V on VCM input from the Gsample/s ADC, is shifted lower for proper interface to different ADC's which require VCM = 0V and have high input impedance: 24 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 50: 50: VOUT -5V R1 0.35 VPP, 0V DC +5V Vx R2 +5V -5V R3 R3 131.3: R1 172.7: R2 41.4: 0.7 VPP, 1.2V DC 0.43 VPP, 1.2V DC +OUT -OUT To ADC LMH6518 LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Figure 61. Output CM Shift Scheme With the scheme of Figure 61, Vx is kept at 1.2V, by proper selection of external resistor values, so that the LMH6518 outputs are not CM-loaded. As was the case with input level shifting, this output level shifting also consumes additional power (0.58W). Output Swing, Clamping, and Operation Beyond Full Scale One of the major concerns in interfacing to low voltage ADC’s (such as the Gsample/s ADC’s that the LMH6518 is intended to drive) is ensuring that the ADC input is not violated with excessive drive. For this reason, plus the very important requirement of an oscilloscope to recover quickly and gracefully from an overdrive condition, the LMH6518 is fitted with three overvoltage clamps; one at the Preamp output and one at Main and Auxiliary outputs each. The Preamp clamp is responsible for preventing the Preamp from saturation (to minimize recovery time) with large ladder attenuation when Preamp output swing is at its highest. On the other hand, the output clamps, perform this function when the Ladder attenuation is lower and hence the output amplifier is closer to saturation, and prolonged recovery, if not properly clamped. The combination of these clamps results in Figure 51, Figure 52, Figure 53, and Figure 54 where it is possible to observe where output limiting starts due to the clamp action. LMH6518 owes its fast recovery time (< 5 ns) from 50% overdrive to the said clamps. Figure 51, Figure 52, Figure 53, and Figure 54, in Typical Performance Characteristics, can be used to determine the LMH6518 linear swing beyond full scale. This information sets the overdrive limit for both oscilloscope waveform capture and for signal triggering. The Preamp clamp is set tighter than the output clamp, evidenced by lower output swing with 20 dB Ladder attenuation than with 0 dB. With high ladder attenuation (20 dB) defining the limit, the graphs show that the “+Out” and “−Out” difference of 0.4V is well inside the clamp range, thereby ensuring 0.8 VPP of unhindered output swing. This corresponds to an overdrive capability of approximately ±7% beyond full scale. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 25 Product Folder Links: LMH6518 Switch > J1 Oscilloscope Input 900 k: 90 k: 50: 10 k: LNA FPGA or MPU DAC +IN -IN SPI +OUT -OUT VCM +OUT Aux -OUT Aux VCM_Aux LMH6518 U1 Trigger Circuit +IN 1 -IN 1 VCMO Gsample/sec 8-Bit ADC SPI (Full Scale Voltage Control) Attenuation = 100x Channel 1 JFET Lo-Noise Amp Attenuation = 1x Attenuation = 10x Fine Gain Adjust VCC Hi-Z/50: Switch Attenuator Block VCC 200: 1 nF 200: LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Here is a block diagram for how the LMH6518 is used in an oscilloscope: Figure 62. Digital Oscilloscope Front-End From Figure 62, the signal path consists of the input impedance switch, the attenuator switch, Low Noise Amplifier (LNA, JFET amplifier) to drive the LMH6518 input (+IN), and the DAC to provide offset adjust. The LNA must have the following characteristics: • Set U1’s common mode level to VCC/2 (∼2.5V) • Very low drift (1 mV shift at LNA output could translate into 88 mV shift at LMH6518 output at max gain, or ∼13% of FS). • Low output impedance (≤ 50Ω) to drive U1, for good settling behavior • Low Noise (<0.98 nV/√Hz) to reduce the impact on the LMH6518 Noise Figure. Note that Figure 62 does not show the necessary capacitors across the resistors in the front-end attenuators (see Figure 71). These capacitors provide frequency response compensation and limit the noise contribution from the resistors so that they do not impact the signal path noise. For more information about front-end attenuator design, including frequency compensation, see REFERENCE for additional resources. • Gain of 1 V/V (or very close to 1 V/V) • Excellent frequency response flatness from DC to > 500-800 MHz to not impact the time domain performance The undriven input (−IN) is biased to VCC/2 using a voltage driver. The impedance driving the LMH6518’s −IN should be closely matched to the LNA’s output impedance for good settling time performance. APPENDIX A shows one possible implementation of the LNA buffer along with performance data. When the LMH6518’s Auxiliary output is not used, it is possible to disable this output using SPI-1 (see LOGIC FUNCTIONS for SPI register map). Electrical Characteristics shows that by doing so, device power dissipation decreases by the reduction in supply current of about 60 mA. As can be seen in Figure 63, in the absence of heavy common loading, the Auxiliary output will be at a voltage close to 1.7V (VCC = 5V). With higher supply voltages, the Auxiliary voltage will also increase and it is important to make sure any circuitry tied to this output is capable of handling the 2.3V possible under VCC worst case condition of 5.5V. 26 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 85°C 4.5 4.7 4.9 5.1 5.3 5.5 1000 1200 1400 1600 1800 2000 2200 2400 AUXILIARY VOLTAGE (mV) VCC (V) 25°C -40°C RL = 100: VCM_Aux = 1.2V No CM Load +OUT Aux and -OUT Aux LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Figure 63. Auxiliary Output Voltage as a Function of VCC LOGIC FUNCTIONS The following LMH6518 functions are controlled using the SPI-1 compatible bus: • Filters (20, 100, 200, 350, 650, 750 MHz or full bandwidth) • Power Mode (Full Power or Auxiliary Hi-Z (high impedance) • Preamp (HG or LG) • Attenuation Ladder (0-20 dB, 10 states) • LMH6518 state “Write” or “Read” back The SPI-1 bus uses 3.3V logic. “SDIO” is the serial digital input-output which can write to the LMH6518 or read back from it. “SCLK” is the bus clock with chip select function controlled by “CS” SPI-1 PIN DESCRIPTIONS Pin Name Type Function and Connection CS Input Serial Chip Select: While this signal is asserted SCLK is used to accept serial data present on SDIO and to source serial data on SDIO. When this signal is de-asserted, SDIO is ignored and SDIO is in TRI-STATE mode. SCLK Input Serial Clock: Serial data are shifted into and out of the device synchronous with this clock signal. SCLK transitions with CS de-asserted are ignored. SCLK to be stopped when not needed to minimize digital crosstalk. SDIO Input-Output Serial Data-In or Data-out: Serial data are shifted into the device (8 bit Command and 16 bit Data) on this pin while CS signal is asserted during Write operation. Serial data are shifted out of the device on this pin during a read operation while CS signal is asserted. At other times, and after one complete Access Cycle (24 bits, see Figure 64 and Figure 65), this input is ignored. This output is in TRI-STATE mode when CS is deasserted. This pin is bi-directional. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 27 Product Folder Links: LMH6518 SCLK Valid SDIO Data Tsu Th SCLK Valid Data Valid Data Tod SDIO Inter-Access Gap XXX 0 X X X X X X X XXX C7 C6 C5 C4 C3 C2 C1 C0 1 2 3 4 Command Field Data Field 16 bits LMH6518 Bus in Tri-State MSB LSB DI5 D1 D0 8 9 24 25 Single Access Cycle SCLK LMH6518 Bus in Tri-State SDIO CS Inter-Access Gap XXX 1 X X X X X X X XXX C7 C6 C5 C4 C3 C2 C1 C0 1 2 Command Field Data Field MSB 16 bits LSB D15 D1 D0 8 9 24 25 Single Access Cycle SCLK LMH6518 Bus in Tri-State SDIO CS 3 4 LMH6518 Bus in Tri-State LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Figure 64. Serial Interface Protocol- Read Operation Figure 65. Serial Interface Protocol- Write Operation Figure 66. Read Timing Figure 67. Write Timing 28 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Table 4. Data Field Filter Pre-amp Ladder Attenuation D15 D14 D13 D12 D11 D10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0 (MSB) (LSB) X 0 0 0 0 0=Full Power 0 See Table 6 0 0=LG See Table 7 1=Aux Hi-Z 1=HG NOTE Bits D5, D9, D11-D14 must be “0”. Otherwise, device operation is undefined and specifications are not ensured. Table 5. Default Power-On Reset Condition Filter Pre-amp Ladder Attenuation D15 D14 D13 D12 D11 D10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0 (MSB) (LSB) 0 0 0 0 0 0 0 0 0 0 0 0 1 0 1 0 Table 6. Filer Selection Data Field Filter Filter BW D8 D7 D6 (MHz) 0 0 0 Full 0 0 1 20 0 1 0 100 0 1 1 200 1 0 0 350 1 0 1 650 1 1 0 750 1 1 1 Unallowed NOTE All filters are low pass single pole roll-off and operate on both Main and Auxiliary outputs. These filters are intended as signal path bandwidth and/ or noise limiting. Table 7. Ladder Attenuation Data Field Ladder Attenuation Ladder Attenuation (dB) D3 D2 D1 D0 0 0 0 0 0 0 0 0 1 −2 0 0 1 0 −4 0 0 1 1 −6 0 1 0 0 −8 0 1 0 1 −10 0 1 1 0 −12 0 1 1 1 −14 1 0 0 0 −16 1 0 0 1 −18 1 0 1 0 −20 1 0 1 1 Unallowed 1 1 0 0 Unallowed 1 1 0 1 Unallowed 1 1 1 0 Unallowed Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 29 Product Folder Links: LMH6518 x 100°C = 1.45 mV {5.9 LSB) PV (9.5 mV + 50 °C ) 2.5V (= 210 LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Table 7. Ladder Attenuation Data Field (continued) Ladder Attenuation Ladder Attenuation (dB) 1 1 1 1 Unallowed NOTE An “Unallowed” SPI-1 state may result in undefined operation where device behavior is not ensured. OSCILLOSCOPE TRIGGER APPLICATIONS With the Auxiliary output of the LMH6518 offering a second output that follows the Main one (except for a slightly reduced distortion performance), the oscilloscope trigger function can be implemented by tapping this output. The “VCM_Aux” input of the LMH6518 allows the Auxiliary common mode to be set. The trigger function can be physically located at a distance from the main signal path, if need be, by taking advantage of the differential Auxiliary output and rejecting any board related common mode interference pick-up at the receive end. If Trigger circuitry is physically close to the LMH6518, the circuit diagram shown in Figure 68 allows operation using only one of two Auxiliary outputs. The unused output does need to be terminated properly using R1, R11 combination. U3 (DAC101C085) generates a 0- 2.5V trigger level, with 2.4 mV resolution (15) or 0.7% (= 2.4 mV x 100/0.35 VPP) of FS, which is compared to the LMH6518 “+Out Aux” by using an ultra-fast comparator, U2 (LMH7220). U2’s complimentary LVDS output is terminated in the required 100Ω load (R10), for best performance, where the LVDS Trigger output is available. The LMH7220’s offset voltage (±9.5 mV) and offset voltage drift (±50 μV/°C) error will be 5.9 LSB (16) of the Trigger DAC (U3). The offset voltage related portion of this error can be nulled-out, if necessary, during the oscilloscope initial calibration. To do so, the LMH6518 input is terminated properly with no input applied and U3 output is adjusted around VCM_Aux voltage (1.2V ±10 mV) while looking for U2’s output transition. U3’s output, relative to VCM_Aux at transition corresponds to U2’s offset error which can be factored into the Trigger readings and thus eliminated, leaving only the Offset voltage temperature drift component (= 2 LSB). 30 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 (= 50 mV x 100) 0.7V 2 U1 LMH6518 +5V 3,4 5,8 16 +5V VCM_Aux R3 3.83 k: 1% R4 1.20 k: 1% 1 2 +OUT Aux -OUT Aux R11 237: R12 237: R1 75: R2 75: - + U2 LMH7220 U3 DAC101 C085 U4 LP3985 10 bit DAC (I 2 C) SDA SCL +5V 2.5V VA VOUT VREF 0-2.5V R10 100: Trigger Output (LVDS) 2 3 4 5 +5V 6 1 +5V LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 Figure 68. Single-Ended Trigger from LMH6518 Auxiliary Output U2’s minimum Toggle Rate specification of 750 Mb/s with ±50 mV overdrive allow the oscilloscope to trigger on repetitive waveforms well above the 500 MHz oscilloscope bandwidth applications, when the input signal is at least 14.3% of FS swing (17) The worst case single event minimum discernable pulse width is set by the LMH7220’s propagation delay specification of 3.63 ns (20 mV overdrive). Both the Main and the Auxiliary outputs can recover gracefully and quickly from a 50% overdrive condition as tabulated in Electrical Characteristics under overdrive Recovery Time. Overdrive conditions beyond 50%, however, could result in longer recovery times due to the interaction between an internal clamp and the common mode feedback loop that sets the output common mode voltage. This may have an impact on both the displayed waveform and the oscilloscope Trigger. The result could be a loss of Trigger pulse and/or visual distortion of the displayed waveform. To avoid this scenario, the oscilloscope should detect an excessive overdrive and go into trigger-loss mode. Done this way, the oscilloscope display would show the last waveform that did not violate the overdrive condition. Preferably there would be a visual indicator on the screen that alerts the user of the situation so that he can correct the excessive condition to return to normal display. Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 31 Product Folder Links: LMH6518 R14 R14 + R21 1 + R5 R1 || R2 Gain (DC) = ¨ ¨ © § ¨ ¨ © § #1 V/V Scope Input Input Attenuators Not Shown C6 5 pF R21 678 k: R14 322 k: - + ½ U1 LMV842 C7 20 nF C3 100 nF R22 1 M: R15 678 k: R11 322 k: LMH6518 +IN Q0 BFQ67 J8 MMBF5486 R20 500: J10 MMBF5486 R16 20: - + ½ U1 LMV842 -5V +5V LMH6518 -IN R5 +5V 500 k: R1 500 k: R3 500 k: R4 500 k: R0 500 k: R9 200: R6 200: -10V +10V R2 Adjust R2 for gain matching between DC and AC C0 1 nF C5 1 nF R17 100: R49 15: R50 15: Offset Control DAC R8 0: LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com APPENDIX A Here is the schematic drawing for a possible implementation of the LNA buffer shown in Figure 62: Figure 69. JFET LNA Implementation CIRCUIT OPERATION This circuit uses an N-Channel JFET (J10) in Source-Follower configuration, to buffer the input signal, with J8 acting as a constant current source. This buffer presents a fixed input impedance (1 MΩ||10 pF) with a gain close to 1 V/V. The signal path is AC coupled through C7 with DC (and low frequency) at LMH6518 +IN maintained through the action of U1. NPN transistor Q0 is an emitter follower which isolates the buffer from the load (LMH6518 input and board traces). The undriven input of the LMH6518, −IN, is biased to 2.5V by R6, R9 voltage divider. The Lower ½ of U1 inverts this voltage and the upper ½ of U1 compares it to the combination of the driven output level at LMH6518 +IN and the scaled version of scope input at R14, R21 junction, and adjusts J10 Gate accordingly to set the LMH6518 +IN. This control loop has a frequency response that covers DC to a few Hz, limited by the roll-off capacitor C3 and R15 combination (1st order approximation). DC and low frequency gain is given by: (18) With the values in Figure 69 → R2 ≈ 452 kΩ: 32 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 R21 R14 = R15 R11 LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 For a flat frequency response, the DC (low frequency) gain needs to be lowered to match the less-than-1 V/V AC (high frequency) path gain through the JFETs. This can be done by increasing the value of R2. By choosing the values of R15 and R11 so that (19) the frequency response at J10 Gate (and consequently the output) will remain flat when C7 starts to conduct. Offset correction is done by varying the voltage at R4, using a DAC or equivalent as shown, in order to shift the LMH6518 +IN voltage relative to −IN. The result is a circuit which shifts the ground referenced scope input to 2.5V (VCC/2) CM with adjustable offset and without any JFET or BJT related offsets. Note that the front-end attenuator (not shown) lower leg resistance should be increased for proper divider-ratio to account for the 1 MΩ shunt due to the series combination of R21 and R14. For example, a 10:1 front-end attenuator could be formed by a series 900 kΩ and a shunt 111 kΩ for a scope BNC input impedance of 1 MΩ (= 900K + (111K || 1M)). Table 8 lists other possible JFET candidates that fall in the range of speed (ft) and low noise needed: Table 8. Suitable JFET Candidates Specifications Company Part Number VP (V) Idss gm (mS) Input C noise (1) Break Calculated ft (mA) (pF) (nV/RtHz) down (V) (MHz) Interfet IF140 −2.2 10 5.5 2.3 4 −20 380 Interfet IF142 −2.2 10 5.5 2.3 4 −25 380 Interfet 2N5397/8 −2.5 13 8 5 2.5 −25 254 Interfet 2N5911/2 −2.5 13 8 5 2.5 254 Interfet J308/9/10 −2.3 21 17 5.8 −25 466 Philips BF513 -3 15 10 5 318 Fairchild MMBF5486 −4 14 7 4 2.5 −25 278 Vishay Siliconix SST441 −3.5 13 6 3.5 4 −35 272 (1) Noise data at ∼ Idss/2 The LNA noise could degrade the scope’s SNR if it is comparable to the input referred noise of the LMH6518. LNA noise is influenced by the following operating conditions: a. JFET equivalent input noise b. BJT Base current Reducing either “a” or “b” above, or both, reduces noise. One way to reduce “a” is to increase R8 (currently set to 0Ω). This will reduce the noise impact of J8 but requires a JFET which has a higher Idss rating in order to maintain the operating current of J10 so that J10’s noise contribution is minimized. Reducing the BJT Base current can be accomplished with increasing R20 at the expenses of higher rise/fall times. A higher β will also reduce the Base current (keep in mind that β and ft at the operating Collector current is what matters). Figure 70 shows the impact of the JFET buffer noise on SNR, compared to SNR in Figure 58, assuming either 3 nV/√Hz or 1.5 nV/√Hz buffer noise for comparison: Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 33 Product Folder Links: LMH6518 R1 900 k: C5 2-5 pF R2 111 k: C1 8 pF C2 65 pF R3 990 k: C6 2-5 pF R4 10.1 k: C3 8 pF C4 780 pF 10:1 100:1 R_LNA 1 M: C_LNA 10 pF JFET LNA 1:1 10:1 100:1 1:1 -2 2 6 10 14 18 22 26 30 34 38 0 2 4 6 8 10 12 SNR IMPACT (dB) GAIN (dB) 42 LNA Noise = 1.5 nV/ LNA Noise = 3 nV/ Hz Hz LMH6518 SNOSB21C –MAY 2008–REVISED JULY 2013 www.ti.com Figure 70. LNA Buffer SNR Impact ATTENUATOR DESIGN Figure 71 shows a front-end attenuator designed to work with the JFET LNA of Figure 69. Figure 71. Front End Attenuator for Figure 69 JFET LNA R_LNA” and “C_LNA” are the input impedance components of the JFET LNA. The 10:1 and 100:1 attenuators bottom resistors (R2 and R4) are adjusted higher to compensate for the LNA’s 1 MΩ input impedance, compared to the case where a high-input-impedance LNA is used. The two switches used on the input and output of the attenuator block must be low capacitance, high isolation switches in order to reduce any speed or crosstalk impact. C1-C4 provide the proper frequency response (and step response) by creating “zeros” that flatten the response for wide-band operation. For the 10:1 attenuator, R1C1 = R2C2. The same applies to the 100:1 attenuator. The shunt capacitors C1-C4 have a very important other benefit in that they roll-off the resistor thermal noise at a low frequency (low pass response, −3 dB down at ∼20 kHz) thereby eliminating any significant noise contribution from the attenuation resistors. Otherwise, the channel noise would be dominated by the attenuator resistor thermal noise. C2 and C6 trimmer capacitors can be adjusted to match the input capacitance regardless of attenuator used. REFERENCE 1. Wideband amplifiers by Peter Staric and Erik Margan, published by Springer in 2006. (Section 5.2). 34 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated Product Folder Links: LMH6518 LMH6518 www.ti.com SNOSB21C –MAY 2008–REVISED JULY 2013 REVISION HISTORY Hooman: Corrected PSRR condition from "HG" to "LG" per CMS C1305178. Changes from Revision A (March 2013) to Revision B Page • Changed layout of National Data Sheet to TI format .......................................................................................................... 34 Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 35 Product Folder Links: LMH6518 PACKAGE OPTION ADDENDUM www.ti.com 24-Jul-2013 Addendum-Page 1 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Qty Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3) Op Temp (°C) Device Marking (4/5) Samples LMH6518SQ/NOPB ACTIVE WQFN RGH 16 1000 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 85 L6518SQ LMH6518SQE/NOPB ACTIVE WQFN RGH 16 250 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 85 L6518SQ LMH6518SQX/NOPB ACTIVE WQFN RGH 16 4500 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 85 L6518SQ (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. 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PACKAGE OPTION ADDENDUM www.ti.com 24-Jul-2013 Addendum-Page 2 In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Reel Diameter (mm) Reel Width W1 (mm) A0 (mm) B0 (mm) K0 (mm) P1 (mm) W (mm) Pin1 Quadrant LMH6518SQ/NOPB WQFN RGH 16 1000 178.0 12.4 4.3 4.3 1.3 8.0 12.0 Q1 LMH6518SQE/NOPB WQFN RGH 16 250 178.0 12.4 4.3 4.3 1.3 8.0 12.0 Q1 LMH6518SQX/NOPB WQFN RGH 16 4500 330.0 12.4 4.3 4.3 1.3 8.0 12.0 Q1 PACKAGE MATERIALS INFORMATION www.ti.com 24-Jul-2013 Pack Materials-Page 1 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LMH6518SQ/NOPB WQFN RGH 16 1000 213.0 191.0 55.0 LMH6518SQE/NOPB WQFN RGH 16 250 213.0 191.0 55.0 LMH6518SQX/NOPB WQFN RGH 16 4500 367.0 367.0 35.0 PACKAGE MATERIALS INFORMATION www.ti.com 24-Jul-2013 Pack Materials-Page 2 MECHANICAL DATA RGH0016A www.ti.com SQA16A (Rev A) IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. 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Packaged in TO-220AB, TO-220FPAB, D2PAK and I2PAK. DESCRIPTION TO-220AB STPS10H100CT A1 A2 K Symbol Parameter Value Unit VRRM Repetitive peak reverse voltage 100 V IF(RMS) RMS forward current 10 A IF(AV) Average forward current d = 0.5 TO-220AB D2PAK / I2PAK Tc = 165°C per diode per device 5 10 A TO-220FPAB Tc = 160°C IFSM Surge non repetitive forward current tp = 10 ms sinusoidal 180 A IRRM Repetitive peak reverse current tp = 2 μs square F = 1kHz 1 A PARM Repetitive peak avalanche power tp = 1μs Tj = 25°C 7200 W Tstg Storage temperature range - 65 to + 175 °C Tj Maximum operating junction temperature * 175 °C dV/dt Critical rate of rise of reverse voltage 10000 V/μs ABSOLUTE RATINGS (limiting values, per diode) A1 A2 K A1 A2 K D2PAK STPS10H100CG K A1 A2 K I2PAK STPS10H100CR * : dPtot dTj Rth j a < - 1 ( ) thermal runaway condition for a diode on its own heatsink A1 A2 K TO-220FPAB STPS10H100CFP STPS10H100CT/CG/CR/CFP 2/7 Symbol Parameter Value Unit Rth (j-c) Junction to case D2PAK / I2PAK TO-220AB Per diode 2.2 °C/W Total 1.3 Rth (c) Coupling 0.3 Rth (j-c) Junction to case TO-220FPAB Per diode 4.5 °C/W Total 3.5 Rth (c) Coupling 2.5 When the diodes 1 and 2 are used simultaneously : D Tj(diode 1) = P(diode1) x Rth(j-c)(Per diode) + P(diode 2) x Rth(c) THERMAL RESISTANCES Symbol Parameter Tests conditions Min. Typ. Max. Unit IR * Reverse leakage current Tj = 25°C VR = VRRM 3.5 μA Tj = 125°C 1.3 4.5 mA VF ** Forward voltage drop Tj = 25°C IF = 5 A 0.73 V Tj = 125°C 0.57 0.61 Tj = 25°C IF = 10 A 0.85 Tj = 125°C 0.66 0.71 Pulse test : * tp = 5 ms, d < 2% ** tp = 380 μs, d < 2% To evaluate the maximum conduction losses use the following equation : P = 0.51 x IF(AV) + 0.02 x IF 2 (RMS) STATIC ELECTRICAL CHARACTERISTICS (per diode) 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 PF(av)(W) IF(av) (A) T d=tp/T tp d = 0.2 d = 0.5 d = 1 d = 0.05 d = 0.1 Fig. 1: Average forward power dissipation versus average forward current (per diode). 0 25 50 75 100 125 150 175 0 1 2 3 4 5 6 Tamb(°C) IF(av)(A) Rth(j-a)=Rth(j-c) Rth(j-a)=15°C/W D²PAK/I²PAK/TO-220AB TO-220FPAB Fig. 2: Average forward current versus ambient temperature (d=0.5, per diode). STPS10H100CT/CG/CR/CFP 3/7 1E-3 1E-2 1E-1 1E+0 0 20 40 60 80 100 120 IM(A) Tc=50°C Tc=75°C Tc=125°C t(s) IM t d=0.5 Fig. 5-1: Non repetitive surge peak forward current versus overload duration (maximum values, per diode) 1E-3 1E-2 1E-1 1E+0 0.0 0.2 0.4 0.6 0.8 1.0 Zth(j-c)/Rth(j-c) tp(s) T d=tp/T tp Single pulse d = 0.1 d = 0.2 d = 0.5 Fig. 6-1: Relative variation of thermal impedance junction to case versus pulse duration (per diode). 1E-3 1E-2 1E-1 1E+0 0 10 20 30 40 50 60 70 80 t(s) IM(A) Tc=50°C Tc=75°C Tc=125°C IM t d=0.5 Fig. 5-2: Non repetitive surge peak forward current versus overload duration (maximum values, per diode)(TO-220FPAB) 1E-3 1E-2 1E-1 1E+0 1E+1 0.0 0.2 0.4 0.6 0.8 1.0 tp(s) Zth(j-c)/Rth(j-c) T d=tp/T tp Single pulse d = 0.1 d = 0.2 d = 0.5 Fig. 6-2: Relative variation of thermal impedance junction to case versus pulse duration (per diode).(TO-220FPAB) 0 0.2 0.4 0.6 0.8 1 1.2 0 25 50 75 100 125 150 Tj(°C) P (t) P (25°C) ARM p ARM Fig. 4: Normalized avalanche power derating versus junction temperature. 0.001 0.01 0.01 0.1 1 0.1 10 100 1000 1 tp(μs) P (t) P (1μs) ARM p ARM Fig. 3: Normalized avalanche power derating versus pulse duration. STPS10H100CT/CG/CR/CFP 4/7 0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 0.1 1.0 10.0 100.0 IFM(A) Tj=125°C Typical values Tj=125°C Tj=25°C Tj=150°C Typical values VFM(V) Fig. 9: Forward voltage drop versus forward current (maximum values, per diode). 0 2 4 6 8 10 12 14 16 18 20 0 10 20 30 40 50 60 70 80 Rth(j-a) (°C/W) S(Cu) (cm²) Fig. 10: Thermal resistance junction to ambient versus copper surface under tab (Epoxy printed circuit board FR4, copper thickness: 35μm) 0 10 20 30 40 50 60 70 80 90 100 1E-2 1E-1 1E+0 1E+1 1E+2 1E+3 1E+4 IR(μA) Tj=125°C Tj=25°C Tj=150°C Tj=100°C VR(V) Fig. 7: Reverse leakage current versus reverse voltage applied (typical values, per diode). 1 2 5 10 20 50 100 10 100 1000 C(pF) F=1MHz Tj=25°C VR(V) Fig. 8: Junction capacitance versus reverse voltage applied (typical values, per diode). 5/7 STPS10H100CT/CG/CR/CFP PACKAGE MECHANICAL DATA D2PAK A C2 D R A2 M V2 C A1 G L L3 L2 B B2 E * * FLAT ZONE NO LESS THAN 2mm REF. DIMENSIONS Millimeters Inches Min. Max. Min. Max. A 4.40 4.60 0.173 0.181 A1 2.49 2.69 0.098 0.106 A2 0.03 0.23 0.001 0.009 B 0.70 0.93 0.027 0.037 B2 1.14 1.70 0.045 0.067 C 0.45 0.60 0.017 0.024 C2 1.23 1.36 0.048 0.054 D 8.95 9.35 0.352 0.368 E 10.00 10.40 0.393 0.409 G 4.88 5.28 0.192 0.208 L 15.00 15.85 0.590 0.624 L2 1.27 1.40 0.050 0.055 L3 1.40 1.75 0.055 0.069 M 2.40 3.20 0.094 0.126 R 0.40 typ. 0.016 typ. V2 0° 8° 0° 8° FOOT PRINT in millimeters 8.90 3.70 1.30 5.08 16.90 10.30 6/7 STPS10H100CT/CG/CR/CFP PACKAGE MECHANICAL DATA I2PAK e D L L1 L2 b1 b b2 E A c2 A1 c REF. DIMENSIONS Millimeters Inches Min. Max. Min. Max. A 4.40 4.60 0.173 0.181 A1 2.49 2.69 0.098 0.106 b 0.70 0.93 0.028 0.037 b1 1.14 1.17 0.044 0.046 b2 1.14 1.17 0.044 0.046 c 0.45 0.60 0.018 0.024 c2 1.23 1.36 0.048 0.054 D 8.95 9.35 0.352 0.368 e 2.40 2.70 0.094 0.106 E 10.0 10.4 0.394 0.409 L 13.1 13.6 0.516 0.535 L1 3.48 3.78 0.137 0.149 L2 1.27 1.40 0.050 0.055 PACKAGE MECHANICAL DATA TO-220FPAB H L3 L2 L4 L6 G G1 F F1 L5 D E L7 A B Dia F2 REF. DIMENSIONS Millimeters Inches Min. Max. Min. Max. A 4.4 4.6 0.173 0.181 B 2.5 2.7 0.098 0.106 D 2.5 2.75 0.098 0.108 E 0.45 0.70 0.018 0.027 F 0.75 1 0.030 0.039 F1 1.15 1.70 0.045 0.067 F2 1.15 1.70 0.045 0.067 G 4.95 5.20 0.195 0.205 G1 2.4 2.7 0.094 0.106 H 10 10.4 0.393 0.409 L2 16 Typ. 0.63 Typ. L3 28.6 30.6 1.126 1.205 L4 9.8 10.6 0.386 0.417 L5 2.9 3.6 0.114 0.142 L6 15.9 16.4 0.626 0.646 L7 9.00 9.30 0.354 0.366 Dia. 3.00 3.20 0.118 0.126 7/7 STPS10H100CT/CG/CR/CFP Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. The ST logo is a registered trademark of STMicroelectronics © 2003 STMicroelectronics - Printed in Italy - All rights reserved. STMicroelectronics GROUP OF COMPANIES Australia - Brazil - Canada - China - Finland - France - Germany Hong Kong - India - Israel - Italy - Japan - Malaysia - Malta - Morocco - Singapore Spain - Sweden - Switzerland - United Kingdom - United States. http://www.st.com Ordering type Marking Package Weight Base qty Delivery mode STPS10H100CT STPS10H100CT TO-220AB 2.20g 50 Tube STPS10H100CFP STPS10H100CFP TO-220FPAB 2.0 g 50 Tube STPS10H100CG STPS10H100CG D2PAK 1.48g 50 Tube STPS10H100CG-TR STPS10H100CG D2PAK 1.48g 1000 Tape and reel STPS10H100CR STPS10H100CR I2PAK 1.49g 50 Tube n Epoxy meets UL94,V0 PACKAGE MECHANICAL DATA TO-220AB A C D L7 Dia L5 L6 L9 L4 F H2 G G1 L2 F2 F1 E M REF. DIMENSIONS Millimeters Inches Min. Max. Min. Max. A 4.40 4.60 0.173 0.181 C 1.23 1.32 0.048 0.051 D 2.40 2.72 0.094 0.107 E 0.49 0.70 0.019 0.027 F 0.61 0.88 0.024 0.034 F1 1.14 1.70 0.044 0.066 F2 1.14 1.70 0.044 0.066 G 4.95 5.15 0.194 0.202 G1 2.40 2.70 0.094 0.106 H2 10 10.40 0.393 0.409 L2 16.4 typ. 0.645 typ. L4 13 14 0.511 0.551 L5 2.65 2.95 0.104 0.116 L6 15.25 15.75 0.600 0.620 L7 6.20 6.60 0.244 0.259 L9 3.50 3.93 0.137 0.154 M 2.6 typ. 0.102 typ. Diam. 3.75 3.85 0.147 0.151 n Cooling method: C. n Recommended torque value: 0.55 m.N n Maximum torque value 0.70 m.N LM19 www.ti.com SNIS122E –MAY 2001–REVISED MARCH 2013 LM19 2.4V, 10μA, TO-92 Temperature Sensor Check for Samples: LM19 1FEATURES DESCRIPTION The LM19 is a precision analog output CMOS 2• Rated for Full −55°C to +130°C Range integrated-circuit temperature sensor that operates • Available in a TO-92 Package over a −55°C to +130°C temperature range. The • Predictable Curvature Error power supply operating range is +2.4 V to +5.5 V. • Suitable for Remote Applications The transfer function of LM19 is predominately linear, • UL Recognized Component yet has a slight predictable parabolic curvature. The accuracy of the LM19 when specified to a parabolic transfer function is ±2.5°C at an ambient temperature APPLICATIONS of +30°C. The temperature error increases linearly • Cellular Phones and reaches a maximum of ±3.8°C at the • Computers temperature range extremes. The temperature range is affected by the power supply voltage. At a power • Power Supply Modules supply voltage of 2.7 V to 5.5 V the temperature • Battery Management range extremes are +130°C and −55°C. Decreasing • FAX Machines the power supply voltage to 2.4 V changes the negative extreme to −30°C, while the positive • Printers remains at +130°C. • HVAC The LM19's quiescent current is less than 10 μA. • Disk Drives Therefore, self-heating is less than 0.02°C in still air. • Appliances Shutdown capability for the LM19 is intrinsic because its inherent low power consumption allows it to be KEY SPECIFICATIONS powered directly from the output of many logic gates or does not necessitate shutdown at all. • Accuracy at +30°C ±2.5 °C (max) • Accuracy at +130°C & −55°C ±3.5 to ±3.8 °C (max) • Power Supply Voltage Range +2.4V to +5.5V • Current Drain 10 μA (max) • Nonlinearity ±0.4 % (typ) • Output Impedance 160 Ω (max) • Load Regulation – 0μA < IL< +16 μA 1 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. 2All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Copyright © 2001–2013, Texas Instruments Incorporated Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. LM19 SNIS122E –MAY 2001–REVISED MARCH 2013 www.ti.com Typical Application Output Voltage vs Temperature VO = (−3.88×10−6×T2) + (−1.15×10−2×T) + 1.8639 or where: T is temperature, and VO is the measured output voltage of the LM19. Figure 1. Full-Range Celsius (Centigrade) Temperature Sensor (−55°C to +130°C) Operating from a Single Li-Ion Battery Cell Temperature (T) Typical VO +130°C +303 mV +100°C +675 mV +80°C +919 mV +30°C +1515 mV +25°C +1574 mV 0°C +1863.9 mV −30°C +2205 mV −40°C +2318 mV −55°C +2485 mV Connection Diagram Figure 2. TO-92 Package Number LP These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 2 Submit Documentation Feedback Copyright © 2001–2013, Texas Instruments Incorporated Product Folder Links: LM19 LM19 www.ti.com SNIS122E –MAY 2001–REVISED MARCH 2013 Absolute Maximum Ratings(1) Supply Voltage +6.5V to −0.2V Output Voltage (V+ + 0.6 V) to −0.6 V Output Current 10 mA Input Current at any pin(2) 5 mA Storage Temperature −65°C to +150°C Maximum Junction Temperature (TJMAX) +150°C ESD Susceptibility(3) Human Body Model 2500 V Machine Model 250 V Lead Temperature TO-92 Package Soldering (3 seconds dwell) +240°C (1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not ensure specific performance limits. For ensured specifications and test conditions, see the Electrical Characteristics. The specified specifications apply only for the test conditions listed. Some performance characteristics may degrade when the device is not operated under the listed test conditions. (2) When the input voltage (VI) at any pin exceeds power supplies (VI < GND or VI > V+), the current at that pin should be limited to 5 mA. (3) The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. The machine model is a 200 pF capacitor discharged directly into each pin. Operating Ratings(1) Specified Temperature Range TMIN ≤ TA ≤ TMAX 2.4 V ≤ V+≤ 2.7 V −30°C ≤ TA ≤ +130°C 2.7 V ≤ V+≤ 5.5 V −55°C ≤ TA ≤ +130°C Supply Voltage Range (V+) +2.4 V to +5.5 V Thermal Resistance, θJA (2) TO-92 150°C/W (1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not ensure specific performance limits. For ensured specifications and test conditions, see the Electrical Characteristics. The specified specifications apply only for the test conditions listed. Some performance characteristics may degrade when the device is not operated under the listed test conditions. (2) The junction to ambient thermal resistance (θJA) is specified without a heat sink in still air. Copyright © 2001–2013, Texas Instruments Incorporated Submit Documentation Feedback 3 Product Folder Links: LM19 LM19 SNIS122E –MAY 2001–REVISED MARCH 2013 www.ti.com Electrical Characteristics Unless otherwise noted, these specifications apply for V+ = +2.7 VDC. Boldface limits apply for TA = TJ = TMIN to TMAX ; all other limits TA = TJ = 25°C; Unless otherwise noted. Parameter Conditions Typical(1) LM19C Units Limits(2) (Limit) Temperature to Voltage Error TA = +25°C to +30°C ±2.5 °C (max) VO = (−3.88×10−6×T2) TA = +130°C ±3.5 °C (max) + (−1.15×10−2×T) + 1.8639V(3) TA = +125°C ±3.5 °C (max) TA = +100°C ±3.2 °C (max) TA = +85°C ±3.1 °C (max) TA = +80°C ±3.0 °C (max) TA = 0°C ±2.9 °C (max) TA = −30°C ±3.3 °C (min) TA = −40°C ±3.5 °C (max) TA = −55°C ±3.8 °C (max) Output Voltage at 0°C +1.8639 V Variance from Curve ±1.0 °C Non-Linearity(4) −20°C ≤ TA ≤ +80°C ±0.4 % Sensor Gain (Temperature Sensitivity −30°C ≤ TA ≤ +100°C −11.77 −11.0 mV/°C (min) or Average Slope) to equation: −12.6 mV/°C (max) VO=−11.77 mV/°C×T+1.860V Output Impedance 0 μA ≤ IL ≤ +16 μA(5) (6) 160 Ω (max) Load Regulation(7) 0 μA ≤ IL ≤ +16 μA(5) (6) −2.5 mV (max) Line Regulation(8) +2. 4 V ≤ V+ ≤ +5.0V +3.7 mV/V (max) +5.0 V ≤ V+ ≤ +5.5 V +11 mV (max) Quiescent Current +2. 4 V ≤ V+ ≤ +5.0V 4.5 7 μA (max) +5.0V ≤ V+ ≤ +5.5V 4.5 9 μA (max) +2. 4 V ≤ V+ ≤ +5.0V 4.5 10 μA (max) Change of Quiescent Current +2. 4 V ≤ V+ ≤ +5.5V +0.7 μA Temperature Coefficient of Quiescent −11 nA/°C Current Shutdown Current V+ ≤ +0.8 V 0.02 μA (1) Typicals are at TJ = TA = 25°C and represent most likely parametric norm. (2) Limits are ensured to AOQL (Average Outgoing Quality Level). (3) Accuracy is defined as the error between the measured and calculated output voltage at the specified conditions of voltage, current, and temperature (expressed in°C). (4) Non-Linearity is defined as the deviation of the calculated output-voltage-versus-temperature curve from the best-fit straight line, over the temperature range specified. (5) Negative currents are flowing into the LM19. Positive currents are flowing out of the LM19. Using this convention the LM19 can at most sink −1 μA and source +16 μA. (6) Load regulation or output impedance specifications apply over the supply voltage range of +2.4V to +5.5V. (7) Regulation is measured at constant junction temperature, using pulse testing with a low duty cycle. Changes in output due to heating effects can be computed by multiplying the internal dissipation by the thermal resistance. (8) Line regulation is calculated by subtracting the output voltage at the highest supply input voltage from the output voltage at the lowest supply input voltage. 4 Submit Documentation Feedback Copyright © 2001–2013, Texas Instruments Incorporated Product Folder Links: LM19 MAX Limit MIN Limit Typical -100 -50 0 50 100 150 -5 -4 -3 -2 -1 0 1 2 3 4 5 ERROR ( ºC) TEMPERATURE (ºC) LM19 www.ti.com SNIS122E –MAY 2001–REVISED MARCH 2013 Typical Performance Characteristics Temperature Error vs. Temperature Thermal Response in Still Air LM19 TRANSFER FUNCTION The LM19's transfer function can be described in different ways with varying levels of precision. A simple linear transfer function, with good accuracy near 25°C, is VO= −11.69 mV/°C × T + 1.8663 V (1) Over the full operating temperature range of −55°C to +130°C, best accuracy can be obtained by using the parabolic transfer function VO = (−3.88×10−6×T2) + (−1.15×10−2×T) + 1.8639 (2) solving for T: (3) A linear transfer function can be used over a limited temperature range by calculating a slope and offset that give best results over that range. A linear transfer function can be calculated from the parabolic transfer function of the LM19. The slope of the linear transfer function can be calculated using the following equation: m = −7.76 × 10−6× T − 0.0115 where • T is the middle of the temperature range of interest and m is in V/°C. (4) For example for the temperature range of Tmin = −30 to Tmax = +100°C: T = 35°C and m = −11.77 mV/°C The offset of the linear transfer function can be calculated using the following equation: b = (VOP(Tmax) + VOP(T) − m × (Tmax+T))/2 where • VOP(Tmax) is the calculated output voltage at Tmax using the parabolic transfer function for VO. • VOP(T) is the calculated output voltage at T using the parabolic transfer function for VO. (5) Using this procedure the best fit linear transfer function for many popular temperature ranges was calculated in Table 1. As shown in Table 1 the error that is introduced by the linear transfer function increases with wider temperature ranges. Copyright © 2001–2013, Texas Instruments Incorporated Submit Documentation Feedback 5 Product Folder Links: LM19 LM19 SNIS122E –MAY 2001–REVISED MARCH 2013 www.ti.com Table 1. First Order Equations Optimized For Different Temperature Ranges Temperature Range Linear Equation Maximum Deviation of Linear Equation from T VO= Parabolic Equation (°C) min (°C) Tmax (°C) −55 +130 −11.79 mV/°C × T + 1.8528 V ±1.41 −40 +110 −11.77 mV/°C × T + 1.8577 V ±0.93 −30 +100 −11.77 mV/°C × T + 1.8605 V ±0.70 -40 +85 −11.67 mV/°C × T + 1.8583 V ±0.65 −10 +65 −11.71 mV/°C × T + 1.8641 V ±0.23 +35 +45 −11.81 mV/°C × T + 1.8701 V ±0.004 +20 +30 −11.69 mV/°C × T + 1.8663 V ±0.004 Mounting The LM19 can be applied easily in the same way as other integrated-circuit temperature sensors. It can be glued or cemented to a surface. The temperature that the LM19 is sensing will be within about +0.02°C of the surface temperature to which the LM19's leads are attached. This presumes that the ambient air temperature is almost the same as the surface temperature; if the air temperature were much higher or lower than the surface temperature, the actual temperature measured would be at an intermediate temperature between the surface temperature and the air temperature. To ensure good thermal conductivity the backside of the LM19 die is directly attached to the GND pin. The tempertures of the lands and traces to the other leads of the LM19 will also affect the temperature that is being sensed. Alternatively, the LM19 can be mounted inside a sealed-end metal tube, and can then be dipped into a bath or screwed into a threaded hole in a tank. As with any IC, the LM19 and accompanying wiring and circuits must be kept insulated and dry, to avoid leakage and corrosion. This is especially true if the circuit may operate at cold temperatures where condensation can occur. Printed-circuit coatings and varnishes such as Humiseal and epoxy paints or dips are often used to ensure that moisture cannot corrode the LM19 or its connections. The thermal resistance junction to ambient (θJA) is the parameter used to calculate the rise of a device junction temperature due to its power dissipation. For the LM19 the equation used to calculate the rise in the die temperature is as follows: TJ = TA + θJA [(V+ IQ) + (V+ − VO) IL] where • IQ is the quiescent current and ILis the load current on the output. (6) Since the LM19's junction temperature is the actual temperature being measured care should be taken to minimize the load current that the LM19 is required to drive. Table 2 summarizes the rise in die temperature of the LM19 without any loading, and the thermal resistance for different conditions. Table 2. Temperature Rise of LM19 Due to Self-Heating and Thermal Resistance (θJA) TO-92 TO-92 no heat sink small heat fin θJA TJ − TA θJA TJ − TA (°C/W) (°C) (°C/W) (°C) Still air 150 TBD TBD TBD Moving air TBD TBD TBD TBD 6 Submit Documentation Feedback Copyright © 2001–2013, Texas Instruments Incorporated Product Folder Links: LM19 LM19 www.ti.com SNIS122E –MAY 2001–REVISED MARCH 2013 Capacitive Loads The LM19 handles capacitive loading well. Without any precautions, the LM19 can drive any capacitive load less than 300 pF as shown in Figure 3. Over the specified temperature range the LM19 has a maximum output impedance of 160 Ω. In an extremely noisy environment it may be necessary to add some filtering to minimize noise pickup. It is recommended that 0.1 μF be added from V+ to GND to bypass the power supply voltage, as shown in Figure 4. In a noisy environment it may even be necessary to add a capacitor from the output to ground with a series resistor as shown in Figure 4. A 1 μF output capacitor with the 160 Ω maximum output impedance and a 200 Ω series resistor will form a 442 Hz lowpass filter. Since the thermal time constant of the LM19 is much slower, the overall response time of the LM19 will not be significantly affected. Figure 3. LM19 No Decoupling Required for Capacitive Loads Less than 300 pF Table 3. LM19 with Filter for Noisy Environment and Capacitive Loading greater than 300 pF R (Ω) C (μF) 200 1 470 0.1 680 0.01 1 k 0.001 Either placement of resistor as shown above is just as effective. Figure 4. LM19 with Filter for Noisy Environment and Capacitive Loading greater than 300 pF Copyright © 2001–2013, Texas Instruments Incorporated Submit Documentation Feedback 7 Product Folder Links: LM19 4.1V R1 R3 R2 LM4040 U3 0.1 PF R4 VOUT V+ VT VTemp + - U1 V+ LM19 U2 (High = overtemp alarm) VT1 VT2 VTEMP VOUT VT1 = R1 + R2||R3 (4.1)R2 VT2 = R2 + R1||R3 (4.1)R2||R3 LM7211 LM19 SNIS122E –MAY 2001–REVISED MARCH 2013 www.ti.com Applications Circuits Figure 5. Centigrade Thermostat Figure 6. Conserving Power Dissipation with Shutdown Figure 7. Suggested Connection to a Sampling Analog to Digital Converter Input Stage Most CMOS ADCs found in ASICs have a sampled data comparator input structure that is notorious for causing grief to analog output devices such as the LM19 and many op amps. The cause of this grief is the requirement of instantaneous charge of the input sampling capacitor in the ADC. This requirement is easily accommodated by the addition of a capacitor. Since not all ADCs have identical input stages, the charge requirements will vary necessitating a different value of compensating capacitor. This ADC is shown as an example only. If a digital output temperature is required please refer to devices such as the LM74. 8 Submit Documentation Feedback Copyright © 2001–2013, Texas Instruments Incorporated Product Folder Links: LM19 LM19 www.ti.com SNIS122E –MAY 2001–REVISED MARCH 2013 REVISION HISTORY Changes from Revision D (March 2013) to Revision E Page • Changed layout of National Data Sheet to TI format ............................................................................................................ 8 Copyright © 2001–2013, Texas Instruments Incorporated Submit Documentation Feedback 9 Product Folder Links: LM19 PACKAGE OPTION ADDENDUM www.ti.com 18-Oct-2013 Addendum-Page 1 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Qty Eco Plan (2) Lead/Ball Finish (6) MSL Peak Temp (3) Op Temp (°C) Device Marking (4/5) Samples LM19CIZ/LFT4 ACTIVE TO-92 LP 3 2000 Green (RoHS & no Sb/Br) SN | CU SN N / A for Pkg Type LM19 CIZ LM19CIZ/NOPB ACTIVE TO-92 LP 3 1800 Green (RoHS & no Sb/Br) SN | CU SN N / A for Pkg Type -55 to 130 LM19 CIZ (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. 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DS22003B-page 1 MCP3421 Features • 18-bit ΔΣ ADC in a SOT-23-6 package • Differential input operation • Self calibration of Internal Offset and Gain per each conversion • On-board Voltage Reference: - Accuracy: 2.048V ± 0.05% - Drift: 5 ppm/°C • On-board Programmable Gain Amplifier (PGA): - Gains of 1,2,4 or 8 • On-board Oscillator • INL: 10 ppm of FSR (FSR = 4.096V/PGA) • Programmable Data Rate Options: - 3.75 SPS (18 bits) - 15 SPS (16 bits) - 60 SPS (14 bits) - 240 SPS (12 bits) • One-Shot or Continuous Conversion Options • Low current consumption: - 145 μA typical (VDD= 3V, Continuous Conversion) - 39 μA typical (VDD= 3V, One-Shot Conversion with 1 SPS) • Supports I2C Serial Interface: - Standard, Fast and High Speed Modes • Single Supply Operation: 2.7V to 5.5V • Extended Temperature Range: -40°C to 125°C Typical Applications • Portable Instrumentation • Weigh Scales and Fuel Gauges • Temperature Sensing with RTD, Thermistor, and Thermocouple • Bridge Sensing for Pressure, Strain, and Force. Package Types Description The MCP3421 is a single channel low-noise, high accuracy ΔΣ A/D converter with differential inputs and up to 18 bits of resolution in a small SOT-23-6 package. The on-board precision 2.048V reference voltage enables an input range of ±2.048V differentially (Δ voltage = 4.096V). The device uses a two-wire I2C compatible serial interface and operates from a single 2.7V to 5.5V power supply. The MCP3421 device performs conversion at rates of 3.75, 15, 60, or 240 samples per second (SPS) depending on the user controllable configuration bit settings using the two-wire I2C serial interface. This device has an on-board programmable gain amplifier (PGA). The user can select the PGA gain of x1, x2, x4, or x8 before the analog-to-digital conversion takes place. This allows the MCP3421 device to convert a smaller input signal with high resolution. The device has two conversion modes: (a) Continuous mode and (b) One-Shot mode. In One-Shot mode, the device enters a low current standby mode automatically after one conversion. This reduces current consumption greatly during idle periods. The MCP3421 device can be used for various high accuracy analog-to-digital data conversion applications where design simplicity, low power, and small footprint are major considerations. Block Diagram 1 2 3 4 5 VIN+ 6 VSS SCL VINVDD SDA SOT-23-6 Top View VSS VDD VIN+ VINSCL SDA Voltage Reference Clock (2.048V) I2C Interface Gain = 1, 2, 4, or 8 VREF ΔΣ ADC PGA Converter Oscillator 18-Bit Analog-to-Digital Converter with I2C Interface and On-Board Reference MCP3421 DS22003B-page 2 © 2006 Microchip Technology Inc. 1.0 ELECTRICAL CHARACTERISTICS Absolute Maximum Ratings† VDD...................................................................................7.0V All inputs and outputs w.r.t VSS ............... –0.3V to VDD+0.3V Differential Input Voltage ...................................... |VDD - VSS| Output Short Circuit Current .................................Continuous Current at Input Pins ....................................................±2 mA Current at Output and Supply Pins ............................±10 mA Storage Temperature.....................................-65°C to +150°C Ambient Temp. with power applied ...............-55°C to +125°C ESD protection on all pins ................ ≥ 6 kV HBM, ≥ 400V MM Maximum Junction Temperature (TJ) ..........................+150°C †Notice: Stresses above those listed under “Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at those or any other conditions above those indicated in the operational listings of this specification is not implied. Exposure to maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS Electrical Specifications: Unless otherwise specified, all parameters apply for TA = -40°C to +85°C, VDD = +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. All ppm units use 2*VREF as full-scale range. Parameters Sym Min Typ Max Units Conditions Analog Inputs Differential Input Range — ±2.048/PGA — V VIN = VIN+ - VINCommon- Mode Voltage Range (absolute) (Note 1) VSS-0.3 — VDD+0.3 V Differential Input Impedance (Note 2) ZIND (f) — 2.25/PGA — MΩ During normal mode operation Common Mode input Impedance ZINC (f) — 25 — MΩ PGA = 1, 2, 4, 8 System Performance Resolution and No Missing Codes (Note 8) 12 — — Bits DR = 240 SPS 14 — — Bits DR = 60 SPS 16 — — Bits DR = 15 SPS 18 — — Bits DR = 3.75 SPS Data Rate (Note 3) DR 176 240 328 SPS S1,S0 = ‘00’, (12 bits mode) 44 60 82 SPS S1,S0 = ‘01’, (14 bits mode) 11 15 20.5 SPS S1,S0 = ‘10’, (16 bits mode) 2.75 3.75 5.1 SPS S1,S0 = ‘11’, (18 bits mode) Output Noise — 1.5 — μVRMS TA = 25°C, DR = 3.75 SPS, PGA = 1, VIN = 0 Note 1: Any input voltage below or greater than this voltage causes leakage current through the ESD diodes at the input pins. This parameter is ensured by characterization and not 100% tested. 2: This input impedance is due to 3.2 pF internal input sampling capacitor. 3: The total conversion speed includes auto-calibration of offset and gain. 4: INL is the difference between the endpoints line and the measured code at the center of the quantization band. 5: Includes all errors from on-board PGA and VREF. 6: Full Scale Range (FSR) = 2 x 2.048/PGA = 4.096/PGA. 7: This parameter is ensured by characterization and not 100% tested. 8: This parameter is ensured by design and not 100% tested. © 2006 Microchip Technology Inc. DS22003B-page 3 MCP3421 Integral Nonlinearity (Note 4) INL — 10 35 ppm of FSR DR = 3.75 SPS (Note 6) Internal Reference Voltage VREF — 2.048 — V Gain Error (Note 5) — 0.05 0.35 % PGA = 1, DR = 3.75 SPS PGA Gain Error Match (Note 5) — 0.1 — % Between any 2 PGA gains Gain Error Drift (Note 5) — 5 40 ppm/°C PGA=1, DR=3.75 SPS Offset Error VOS — 15 40 μV Tested at PGA = 1 VDD = 5.0V and DR = 3.75 SPS Offset Drift vs. Temperature — 50 — nV/°C VDD = 5.0V Common-Mode Rejection — 105 — dB at DC and PGA =1, — 110 — dB at DC and PGA =8, TA = +25°C Gain vs. VDD — 5 — ppm/V TA = +25°C, VDD = 2.7V to 5.5V, PGA = 1 Power Supply Rejection at DC — 100 — dB TA = +25°C, VDD = 2.7V to 5.5V, PGA = 1 Power Requirements Voltage Range VDD 2.7 — 5.5 V Supply Current during Conversion IDDA — 155 190 μA VDD = 5.0V — 145 — μA VDD = 3.0V Supply Current during Standby Mode IDDS — 0.1 0.5 μA I2C Digital Inputs and Digital Outputs High level input voltage VIH 0.7 VDD — VDD V Low level input voltage VIL — — 0.3VDD V Low level output voltage VOL — — 0.4 V IOL = 3 mA, VDD = +5.0V Hysteresis of Schmitt Trigger for inputs (Note 7) VHYST 0.05VDD — — V fSCL = 100 kHz Supply Current when I2C bus line is active IDDB — — 10 μA Input Leakage Current IILH — — 1 μA VIH = 5.5V IILL -1 — — μA VIL = GND Pin Capacitance and I2C Bus Capacitance Pin capacitance CPIN — — 10 pF I2C Bus Capacitance Cb — — 400 pF Thermal Characteristics Specified Temperature Range TA -40 — +85 °C Operating Temperature Range TA -40 — +125 °C Storage Temperature Range TA -65 — +150 °C ELECTRICAL CHARACTERISTICS (CONTINUED) Electrical Specifications: Unless otherwise specified, all parameters apply for TA = -40°C to +85°C, VDD = +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. All ppm units use 2*VREF as full-scale range. Parameters Sym Min Typ Max Units Conditions Note 1: Any input voltage below or greater than this voltage causes leakage current through the ESD diodes at the input pins. This parameter is ensured by characterization and not 100% tested. 2: This input impedance is due to 3.2 pF internal input sampling capacitor. 3: The total conversion speed includes auto-calibration of offset and gain. 4: INL is the difference between the endpoints line and the measured code at the center of the quantization band. 5: Includes all errors from on-board PGA and VREF. 6: Full Scale Range (FSR) = 2 x 2.048/PGA = 4.096/PGA. 7: This parameter is ensured by characterization and not 100% tested. 8: This parameter is ensured by design and not 100% tested. MCP3421 DS22003B-page 4 © 2006 Microchip Technology Inc. 2.0 TYPICAL PERFORMANCE CURVES Note: Unless otherwise indicated, TA = -40°C to +85°C, VDD = +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. FIGURE 2-1: INL vs. Supply Voltage (VDD). FIGURE 2-2: INL vs. Temperature. FIGURE 2-3: Offset Error vs. Temperature. FIGURE 2-4: Noise vs. Input Voltage. FIGURE 2-5: Total Error vs. Input Voltage. FIGURE 2-6: Gain Error vs. Temperature. Note: The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. .000 .001 .002 .003 .004 .005 2.5 3 3.5 4 4.5 5 5.5 VDD (V) PGA = 1 PGA = 2 PGA = 8 PGA = 4 Integral Nonlinearity (% of FSR) 0 0.001 0.002 0.003 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (oC) Integral Nonlinearity (% of FSR) VDD = 5 V VDD = 2.7V PGA = 1 -20 -15 -10 -5 0 5 10 15 20 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (°C) Offset Error (μV) VDD = 5V PGA = 1 PGA = 2 PGA = 4 PGA = 8 0.0 2.5 5.0 7.5 10.0 -100 -75 -50 -25 0 25 50 75 100 Input Voltage (% of Full-Scale) Noise (μV, rms) PGA = 1 PGA = 2 PGA = 8 PGA = 4 TA = +25°C VDD = 5V -3.0 -2.0 -1.0 0.0 1.0 2.0 3.0 -100 -75 -50 -25 0 25 50 75 100 Input Voltage (% of Full-Scale) Total Error (mV) PGA = 1 PGA = 2 PGA = 8 PGA = 4 -0.4 -0.3 -0.2 -0.1 0 0.1 0.2 0.3 0.4 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (°C) Gain Error (% of FSR) VDD = 5.0V PGA = 1 PGA = 2 PGA = 8 PGA = 4 © 2006 Microchip Technology Inc. DS22003B-page 5 MCP3421 Note: Unless otherwise indicated, TA = -40°C to +85°C, VDD = +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. FIGURE 2-7: IDDA vs. Temperature. FIGURE 2-8: IDDS vs. Temperature. FIGURE 2-9: IDDB vs. Temperature. FIGURE 2-10: OSC Drift vs. Temperature. FIGURE 2-11: Frequency Response. 100 120 140 160 180 200 220 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (oC) IDDA (μA) VDD = 5V VDD = 2.7V 0 100 200 300 400 500 600 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (oC) IDDS (nA) VDD = 2.7V VDD = 5V 0 1 2 3 4 5 6 7 8 9 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (oC) IDDB (A) VDD = 5V VDD = 4.5V VDD = 3.3V VDD = 2.7V -1 0 1 2 3 4 5 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (°C) Oscillator Drift (%) VDD = 5.0V VDD = 2.7V Data Rate = 3.75 SPS -120 -110 -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 0 0.1 1 10 100 1000 10000 Input Signal Frequency (Hz) Magnitude (dB) 0.1 1 10 100 1k 10k MCP3421 DS22003B-page 6 © 2006 Microchip Technology Inc. 3.0 PIN DESCRIPTIONS TABLE 3-1: PIN FUNCTION TABLE 3.1 Analog Inputs (VIN+, VIN-) VIN+ and VIN- are differential signal input pins. The MCP3421 device accepts a fully differential analog input signal which is connected on the VIN+ and VINinput pins. The differential voltage that is converted is defined by VIN = (VIN+ - VIN-) where VIN+ is the voltage applied at the VIN+ pin and VIN- is the voltage applied at the VIN- pin. The input signal level is amplified by the programmable gain amplifier (PGA) before the conversion. The differential input voltage should not exceed an absolute of (2* VREF/PGA) for accurate measurement, where VREF is the internal reference voltage (2.048V) and PGA is the PGA gain setting. The converter output code will saturate if the input range exceeds (2* VREF/PGA). The absolute voltage range on each of the differential input pins is from VSS-0.3V to VDD+0.3V. Any voltage above or below this range will cause leakage currents through the Electrostatic Discharge (ESD) diodes at the input pins. This ESD current can cause unexpected performance of the device. The common mode of the analog inputs should be chosen such that both the differential analog input range and the absolute voltage range on each pin are within the specified operating range defined in Section 1.0 “Electrical Characteristics” and Section 4.0 “Description of Device Operation”. 3.2 Supply Voltage (VDD, VSS) VDD is the power supply pin for the device. This pin requires an appropriate bypass capacitor of about 0.1 μF (ceramic) to ground. An additional 10 μF capacitor (tantalum) in parallel is also recommended to further attenuate high frequency noise present in some application boards. The supply voltage (VDD) must be maintained in the 2.7V to 5.5V range for specified operation. VSS is the ground pin and the current return path of the device. The user must connect the VSS pin to a ground plane through a low impedance connection. If an analog ground path is available in the application PCB (printed circuit board), it is highly recommended that the VSS pin be tied to the analog ground path or isolated within an analog ground plane of the circuit board. 3.3 Serial Clock Pin (SCL) SCL is the serial clock pin of the I2C interface. The MCP3421 acts only as a slave and the SCL pin accepts only external serial clocks. The input data from the Master device is shifted into the SDA pin on the rising edges of the SCL clock and output from the MCP3421 occurs at the falling edges of the SCL clock. The SCL pin is an open-drain N-channel driver. Therefore, it needs a pull-up resistor from the VDD line to the SCL pin. Refer to Section 5.3 “I2C Serial Communications” for more details of I2C Serial Interface communication. 3.4 Serial Data Pin (SDA) SDA is the serial data pin of the I2C interface. The SDA pin is used for input and output data. In read mode, the conversion result is read from the SDA pin (output). In write mode, the device configuration bits are written (input) though the SDA pin. The SDA pin is an opendrain N-channel driver. Therefore, it needs a pull-up resistor from the VDD line to the SDA pin. Except for start and stop conditions, the data on the SDA pin must be stable during the high period of the clock. The high or low state of the SDA pin can only change when the clock signal on the SCL pin is low. Refer to Section 5.3 “I2C Serial Communications” for more details of I2C Serial Interface communication. Pin No Sym Function 1 VIN+ Non-Inverting Analog Input Pin 2 VSS Ground Pin 3 SCL Serial Clock Input Pin of the I2C Interface 4 SDA Bidirectional Serial Data Pin of the I2C Interface 5 VDD Positive Supply Voltage Pin 6 VIN- Inverting Analog Input Pin © 2006 Microchip Technology Inc. DS22003B-page 7 MCP3421 4.0 DESCRIPTION OF DEVICE OPERATION 4.1 General Overview The MCP3421 is a low-power, 18-Bit Delta-Sigma A/D converter with an I2C serial interface. The device contains an on-board voltage reference (2.048V), programmable gain amplifier (PGA), and internal oscillator. The user can select 12, 14, 16, or 18 bit conversion by setting the configuration register bits. The device can be operated in Continuous Conversion or One-Shot Conversion mode. In the Continuous Conversion mode, the device converts the inputs continuously. While in the One-Shot Conversion mode, the device converts the input one time and stays in the low-power standby mode until it receives another command for a new conversion. During the standby mode, the device consumes less than 0.1 μA typical. 4.2 Power-On-Reset (POR) The device contains an internal Power-On-Reset (POR) circuit that monitors power supply voltage (VDD) during operation. This circuit ensures correct device start-up at system power-up and power-down events. The POR has built-in hysteresis and a timer to give a high degree of immunity to potential ripples and noises on the power supply. A 0.1 μF decoupling capacitor should be mounted as close as possible to the VDD pin for additional transient immunity. The threshold voltage is set at 2.2V with a tolerance of approximately ±5%. If the supply voltage falls below this threshold, the device will be held in a reset condition. The typical hysteresis value is approximately 200 mV. The POR circuit is shut-down during the low-power standby mode. Once a power-up event has occurred, the device requires additional delay time (approximately 300 μs) before a conversion can take place. During this time, all internal analog circuitries are settled before the first conversion occurs. Figure 4-1 illustrates the conditions for power-up and power-down events under typical start-up conditions. When the device powers up, it automatically resets and sets the configuration bits to default settings. The default configuration bit conditions are a PGA gain of 1 V/V and a conversion speed of 240 SPS in Continuous Conversion mode. When the device receives an I2C General Call Reset command, it performs an internal reset similar to a Power-On-Reset event. FIGURE 4-1: POR Operation. 4.3 Internal Voltage Reference The device contains an on-board 2.048V voltage reference. This reference voltage is for internal use only and not directly measurable. The specifications of the reference voltage are part of the device’s gain and drift specifications. Therefore, there is no separate specification for the on-board reference. 4.4 Analog Input Channel The differential analog input channel has a switched capacitor structure. The internal sampling capacitor (3.2 pF) is charged and discharged to process a conversion. The charging and discharging of the input sampling capacitor creates dynamic input currents at the VIN+ and VIN- input pins, which is inversely proportional to the internal sampling capacitor and internal frequency. The current is also a function of the differential input voltages. Care must be taken in setting the common-mode voltage and input voltage ranges so that the input limits do not exceed the ranges specified in Section 1.0 “Electrical Characteristics”. 4.5 Digital Output Code The digital output code produced by the MCP3421 is a function of PGA gain, input signal, and internal reference voltage. In a fixed setting, the digital output code is proportional to the voltage difference between the two analog inputs. The output data format is a binary two’s complement. With this code scheme, the MSB can be considered a sign indicator. When the MSB is a logic ‘0’, it indicates a positive value. When the MSB is a logic ‘1’, it indicates a negative value. The following is an example of the output code: (a) for a negative full-scale input voltage: 100...000 (b) for a zero differential input voltage: 000...000 (c) for a positive full-scale input voltage: 011...111. The MSB is always transmitted first through the serial port. The number of data bits for each conversion is 18, 16, 14, or 12 bits depending on the conversion mode selection. VDD 2.2V 2.0V 300 μS Reset Start-up Normal Operation Reset Time MCP3421 DS22003B-page 8 © 2006 Microchip Technology Inc. The output codes will not roll-over if the input voltage exceeds the maximum input range. In this case, the code will be locked at 0111...11 for all voltages greater than +(VREF - 1 LSB) and 1000...00 for voltages less than -VREF. Table 4-2 shows an example of output codes of various input levels using 18 bit conversion mode. Table 4-3 shows an example of minimum and maximum codes for each data rate option. The output code is given by: EQUATION 4-1: The LSB of the code is given by: EQUATION 4-2: TABLE 4-1: LSB SIZE OF VARIOUS BIT CONVERSION MODES TABLE 4-2: EXAMPLE OF OUTPUT CODE FOR 18 BITS TABLE 4-3: MINIMUM AND MAXIMUM CODES 4.6 Self-Calibration The device performs a self-calibration of offset and gain for each conversion. This provides reliable conversion results from conversion-to-conversion over variations in temperature as well as power supply fluctuations. 4.7 Input Impedance The MCP3421 uses a switched-capacitor input stage using a 3.2 pF sampling capacitor. This capacitor is switched (charged and discharged) at a rate of the sampling frequency that is generated by the on-board clock. The differential mode impedance varies with the PGA settings. The typical differential input impedance during a normal mode operation is given by: Since the sampling capacitor is only switching to the input pins during a conversion process, the above input impedance is only valid during conversion periods. In a low power standby mode, the above impedance is not presented at the input pins. Therefore, only a leakage current due to ESD diode is presented at the input pins. The conversion accuracy can be affected by the input signal source impedance when any external circuit is connected to the input pins. The source impedance adds to the internal impedance and directly affects the time required to charge the internal sampling capacitor. Therefore, a large input source impedance connected to the input pins can increase the system performance errors such as offset, gain, and integral nonlinearity (INL) errors. Ideally, the input source impedance should be zero. This can be achievable by using an operational amplifier with a closed-loop output impedance of tens of ohms. Bit Resolutions LSB (V) 12 bits 1 mV 14 bits 250 μV 16 bits 62.5 μV 18 bits 15.625 μV Input Voltage (V) Digital Code ≥ VREF 011111111111111111 VREF - 1 LSB 011111111111111111 2 LSB 000000000000000010 1 LSB 000000000000000001 0 000000000000000000 -1 LSB 111111111111111111 -2 LSB 111111111111111110 - VREF 100000000000000000 < -VREF 100000000000000000 Output Code (Max Code + 1) (VIN+ – VIN-) 2.048V = × -------------------------------------- LSB 2 × 2.048V 2N = -------------------------- Where: N = the number of bits Number of Bits Data Rate Minimum Code Maximum Code 12 240 SPS -2048 2047 14 60 SPS -8192 8191 16 15 SPS -32768 32767 18 3.75 SPS -131072 131071 Note: Maximum n-bit code = 2n-1 - 1 Minimum n-bit code = -1 x 2n-1 ZIN(f) = 2.25 MΩ/PGA © 2006 Microchip Technology Inc. DS22003B-page 9 MCP3421 4.8 Aliasing and Anti-aliasing Filter Aliasing occurs when the input signal contains timevarying signal components with frequency greater than half the sample rate. In the aliasing conditions, the device can output unexpected output codes. For applications that are operating in electrical noise environments, the time-varying signal noise or high frequency interference components can be easily added to the input signals and cause aliasing. Although the MCP3421 device has an internal first order sinc filter, its’ filter response may not give enough attenuation to all aliasing signal components. To avoid the aliasing, an external anti-aliasing filter, which can be accomplished with a simple RC low-pass filter, is typically used at the input pins. The low-pass filter cuts off the high frequency noise components and provides a band-limited input signal to the MCP3421 input pins. MCP3421 DS22003B-page 10 © 2006 Microchip Technology Inc. 5.0 USING THE MCP3421 DEVICE 5.1 Operating Modes The user operates the device by setting up the device configuration register and reads the conversion data using serial I2C interface commands. The MCP3421 operates in two modes: (a) Continuous Conversion Mode or (b) One-Shot Conversion Mode (single conversion). The selection is made by setting the O/C bit in the Configuration Register. Refer to Section 5.2 “Configuration Register” for more information. 5.1.1 CONTINUOUS CONVERSION MODE (O/C BIT = 1) The MCP3421 device performs a Continuous Conversion if the O/C bit is set to logic “high”. Once the conversion is completed, the result is placed at the output data register. The device immediately begins another conversion and overwrites the output data register with the most recent data. The device also clears the data ready flag (RDY bit = 0) when the conversion is completed. The device sets the ready flag bit (RDY bit = 1), if the latest conversion result has been read by the Master. 5.1.2 ONE-SHOT CONVERSION MODE (O/C BIT = 0) Once the One-Shot Conversion (single conversion) Mode is selected, the device performs a conversion, updates the Output Data register, clears the data ready flag (RDY = 0), and then enters a low power standby mode. A new One-Shot Conversion is started again when the device receives a new write command with RDY = 1. This One-Shot Conversion Mode is recommended for low power operating applications. During the low current standby mode, the device consumes less than 1 μA typical. For example, if user collects 18 bit conversion data once a second in One-Shot Conversion mode, the device draws only about one fourth of its total operating current. In this example, the device consumes approximately 39 μA (= ~145 μA/3.75 SPS), if the device performs only one conversion per second (1 SPS) in 18-bit conversion mode with 3V power supply. © 2006 Microchip Technology Inc. DS22003B-page 11 MCP3421 5.2 Configuration Register The MCP3421 has an 8-bit wide configuration register to select for: PGA gain, conversion rate, and conversion mode. This register allows the user to change the operating condition of the device and check the status of the device operation. The user can rewrite the configuration byte any time during the device operation. Register 5-1 shows the configuration register bits. REGISTER 5-1: CONFIGURATION REGISTER R/W-1 R/W-0 R/W-0 R/W-1 R/W-0 R/W-0 R/W-0 R/W-0 RDY C1 C0 O/C S1 S0 G1 G0 1 * 0 * 0 * 1 * 0 * 0 * 0 * 0 * bit 7 bit 0 * Default Configuration after Power-On Reset Legend: R = Readable bit W = Writable bit U = Unimplemented bit, read as ‘0’ -n = Value at POR ‘1’ = Bit is set ‘0’ = Bit is cleared x = Bit is unknown bit 7 RDY: Ready Bit This bit is the data ready flag. In read mode, this bit indicates if the output register has been updated with a new conversion. In One-Shot Conversion mode, writing this bit to “1” initiates a new conversion. Reading RDY bit with the read command: 1 = Output register has not been updated. 0 = Output register has been updated with the latest conversion data. Writing RDY bit with the write command: Continuous Conversion mode: No effect One-Shot Conversion mode: 1 = Initiate a new conversion. 0 = No effect. bit 6-5 C1-C0: Channel Selection Bits These are the Channel Selection bits, but not used in the MCP3421 device. bit 4 O/C: Conversion Mode Bit 1 = Continuous Conversion Mode. Once this bit is selected, the device performs data conversions continuously. 0 = One-Shot Conversion Mode. The device performs a single conversion and enters a low power standby mode until it receives another write/read command. bit 3-2 S1-S0: Sample Rate Selection Bit 00 = 240 SPS (12 bits), 01 = 60 SPS (14 bits), 10 = 15 SPS (16 bits), 11 = 3.75 SPS (18 bits) bit 1-0 G1-G0: PGA Gain Selector Bits 00 = 1 V/V, 01 = 2 V/V, 10 = 4 V/V, 11 = 8 V/V MCP3421 DS22003B-page 12 © 2006 Microchip Technology Inc. In read mode, the RDY bit in the configuration byte indicates the state of the conversion: (a) RDY = 1 indicates that the data bytes that have just been read were not updated from the previous conversion. (b) RDY = 0 indicates that the data bytes that have just been read were updated. If the configuration byte is read repeatedly by clocking continuously after the first read (i.e., after the 5th byte in the 18-bit conversion mode), the state of the RDY bit indicates whether the device is ready with new conversion data. See Figure 5-2. For example, RDY = 0 means new conversion data is ready for reading. In this case, the user can send a stop bit to exit the current read operation and send a new read command to read out updated conversion data. See Figures 5-2 and 5-3 for reading conversion data. The user can rewrite the configuration byte any time for a new setting. Tables 5-1 and 5-2 show the examples of the configuration bit operation. 5.3 I2C Serial Communications The MCP3421 device communicates with Master (microcontroller) through a serial I2C (Inter-Integrated Circuit) interface and supports standard (100 kbits/sec), fast (400 kbits/sec) and high-speed (3.4 Mbits/sec) modes. The serial I2C is a bidirectional 2-wire data bus communication protocol using opendrain SCL and SDA lines. The MCP3421 can only be addressed as a slave. Once addressed, it can receive configuration bits or transmit the latest conversion results. The serial clock pin (SCL) is an input only and the serial data pin (SDA) is bidirectional. An example of a hardware connection diagram is shown in Figure 6-1. The Master starts communication by sending a START bit and terminates the communication by sending a STOP bit. The first byte after the START bit is always the address byte of the device, which includes the device code, the address bits, and the R/W bit. The device code for the MCP3421 device is 1101. The address bits (A2, A1, A0) are pre-programmed at the factory. In general, the address bits are specified by the customer when they order the device. The three address bits are programmed to “000” at the factory, if they are not specified by the customer. Figure 5-1 shows the details of the MCP3421 address byte. During a low power standby mode, SDA and SCL pins remain at a floating condition. More details of the I2C bus characteristic is described in Section 5.6 “I2C Bus Characteristics”. 5.3.1 DEVICE ADDRESSING The address byte is the first byte received following the START condition from the Master device. The MCP3421 device code is 1101. The device code is followed by three address bits (A2, A1, A0) which are programmed at the factory. The three address bits allow up to eight MCP3421 devices on the same data bus line. The (R/W) bit determines if the Master device wants to read the conversion data or write to the Configuration register. If the (R/W) bit is set (read mode), the MCP3421 outputs the conversion data in the following clocks. If the (R/W) bit is cleared (write mode), the MCP3421 expects a configuration byte in the following clocks. When the MCP3421 receives the correct address byte, it outputs an acknowledge bit after the R/W bit. Figure 5-1 shows the MCP3421 address byte. See Figures 5-2 and 5-3 for the read and write operations of the device. TABLE 5-1: CONFIGURATION BITS FOR WRITING R/W O/C RDY Operation 0 0 0 No effect if all other bits remain the same - operation continues with the previous settings 0 0 1 Initiate One-Shot Conversion 0 1 0 Initiate Continuous Conversion 0 1 1 Initiate Continuous Conversion TABLE 5-2: CONFIGURATION BITS FOR READING R/W O/C RDY Operation 1 0 0 New conversion data in One- Shot conversion mode has been just read. The RDY bit remains low until set by a new write command. 1 0 1 One-Shot Conversion is in progress, The conversion data is not updated yet. The RDY bit stays high. 1 1 0 New conversion data in Continuous Conversion mode has been just read. The RDY bit changes to high after this read. 1 1 1 The conversion data in Continuous Conversion mode was already read. The latest conversion data is not ready. The RDY bit stays high until a new conversion is completed. © 2006 Microchip Technology Inc. DS22003B-page 13 MCP3421 FIGURE 5-1: MCP3421 Address Byte. 5.3.2 READING DATA FROM THE DEVICE When the Master sends a read command (R/W = 1), the MCP3421 outputs the conversion data bytes and configuration byte. Each byte consists of 8 bits with one acknowledge (ACK) bit. The ACK bit after the address byte is issued by the MCP3421 and the ACK bits after each conversion data bytes are issued by the Master. When the device is configured for 18-bit conversion mode, the device outputs three data bytes followed by a configuration byte. The first 7 data bits in the first data byte are the MSB of the conversion data. The user can ignore the first 6 data bits, and take the 7th data bit (D17) as the MSB of the conversion data. The LSB of the 3rd data byte is the LSB of the conversion data (D0). If the device is configured for 12, 14, or 16 bit-mode, the device outputs two data bytes followed by a configuration byte. In 16 bit-conversion mode, the MSB of the first data byte is the MSB (D15) of the conversion data. In 14-bit conversion mode, the first two bits in the first data byte can be ignored (they are the MSB of the conversion data), and the 3rd bit (D13) is the MSB of the conversion data. In 12-bit conversion mode, the first four bits can be ignored (they are the MSB of the conversion data), and the 5th bit (D11) of the byte represents the MSB of the conversion data. Table 5-3 shows an example of the conversion data output of each conversion mode. The configuration byte follows the output data byte. The device outputs the configuration byte as long as the SCL pulses are received. The device terminates the current outputs when it receives a Not-Acknowledge (NAK), a repeated start or a stop bit at any time during the output bit stream. It is not required to read the configuration byte. However, the user may read the configuration byte to check the RDY bit condition to confirm whether the just received data bytes are updated conversion data. The user may continuously send clock (SCL) to repeatedly read the configuration bytes to check the RDY bit status. Figures 5-2 and 5-3 show the timing diagrams of the reading. 5.3.3 WRITING A CONFIGURATION BYTE TO THE DEVICE When the Master sends an address byte with the R/W bit low (R/W = 0), the MCP3421 expects one configuration byte following the address. Any byte sent after this second byte will be ignored. The user can change the operating mode of the device by writing the configuration register bits. If the device receives a write command with a new configuration setting, the device immediately begins a new conversion and updates the conversion data. Start bit Read/Write bit Address Byte R/W ACK 1 1 0 1 X X X Device Code Address Bits (Note 1) Address Acknowledge bit Address Note 1: Specified by customer and programmed at the factory. If not specified by the customer, programmed to ‘000’. TABLE 5-3: EXAMPLE OF CONVERSION DATA OUTPUT OF EACH CONVERSION MODE Conversion Mode Conversion Data Output 18-bits MMMMMMMD16 (1st data byte) - D15 ~ D8 (2nd data byte) - D7 ~ D0 (3rd data byte) - Configuration byte 16-bits MD14~D8 (1st data byte) - D7 ~ D0 (2nd data byte) - Configuration byte 14-bits MMMD12~D8 (1st data byte) - D7 ~ D0 (2nd data byte) - Configuration byte 12-bits MMMMMD10D9D8 (1st data byte) - D7 ~ D0 (2nd data byte) - Configuration byte Note: M is MSB of the data byte. MCP3421 DS22003B-page 14 © 2006 Microchip Technology Inc. FIGURE 5-2: Timing Diagram For Reading From The MCP3421 With 18-Bit Mode. 1 9 1 9 1 9 1 9 1 9 1 9 1 1 0 1 A2 A1 A0 D ACK by RDY O/C MCP3421 7 Start Bit by R/W Master Repeat of D17 (MSB) 2nd Byte Upper Data Byte (Data on Clocks 1-6th can be ignored) ACK by Master ACK by Master ACK by Master ACK by Master 17 D 16 D 15 D 14 D 13 D 12 D 11 D 10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0 C1 C0 S1 S0 G1 G0 1st Byte MCP3421 Address Byte 3rd Byte Middle Data Byte 4th Byte Lower Data Byte 5th Byte Configuration Byte (Optional) C1 C0 S1 S0 G1 G0 NAK by Master Stop Bit by Master (Optional) Nth Repeated Byte: Configuration Byte Note: – MCP3421 device code is 1101. – Address Bits A2- A0 = 000 are programmed at the factory unless customer requests specific codes. – Stop bit or NAK bit can be issued any time during reading. – Data bits on clocks 1 - 6th in 2nd byte are repeated MSB and can be ignored. SCL SDA RDY O/C © 2006 Microchip Technology Inc. DS22003B-page 15 MCP3421 FIGURE 5-3: Timing Diagram For Reading From The MCP3421 With 12-Bit to 16-Bit Modes. 1 1 0 1 A2 A1 A0 ACK by MCP3421 Start Bit by Master 2nd Byte Middle Data Byte ACK by Master ACK by Master ACK by Master D 15 D 14 D 13 D 12 D 11 D 10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0 C1 C0 S1 S0 G1 G0 1st Byte MCP3421 Address Byte 3rd Byte Lower Data Byte 4th Byte Configuration Byte (Optional) C1 C0 S1 S0 G1 G0 NAK by Master Stop Bit by Master (Optional) Nth Repeated Byte: Configuration Byte Note: – MCP3421 device code is 1101. – Address Bits A2- A0 = 000 are programmed at the factory unless customer requests specific codes. – Stop bit or NAK bit can be issued any time during reading. – In 14 - bit mode: D15 and D14 are repeated MSB and can be ignored. – In 12 - bit mode: D15 - D12 are repeated MSB and can be ignored. 1 9 1 9 1 9 1 9 SCL SDA 1 9 R/W RDY O/C RDY O/C MCP3421 DS22003B-page 16 © 2006 Microchip Technology Inc. FIGURE 5-4: Timing Diagram For Writing To The MCP3421. 5.4 General Call The MCP3421 acknowledges the general call address (0x00 in the first byte). The meaning of the general call address is always specified in the second byte. Refer to Figure 5-5. The MCP3421 supports the following general calls: 5.4.1 GENERAL CALL RESET The general call reset occurs if the second byte is ‘00000110’ (06h). At the acknowledgement of this byte, the device will abort current conversion and perform an internal reset similar to a power-on-reset (POR). 5.4.2 GENERAL CALL CONVERSION The general call conversion occurs if the second byte is ‘00001000’ (08h). All devices on the bus initiate a conversion simultaneously. For the MCP3421 device, the configuration will be set to the One-Shot Conversion mode and a single conversion will be performed. The PGA and data rate settings are unchanged with this general call. FIGURE 5-5: General Call Address Format. For more information on the general call, or other I2C modes, please refer to the Phillips I2C specification. 1 9 1 9 Stop Bit by 1 1 0 1 A2 A1 A0 R/W ACK by MCP3421 RDY C1 C0 O/C S1 S0 G1 G0 1st Byte: 2nd Byte: Master ACK by MCP3421 MCP3421 Address Byte Configuration Byte Start Bit by Master with Write command Note: – Stop bit can be issued any time during writing. – MCP3421 device code is 1101. – Address Bits A2- A0 = 000 are programmed at factory unless customer requests different codes. SCL SDA Note: The I2C specification does not allow to use “00000000” (00h) in the second byte. LSB First Byte ACK 0 0 0 0 0 0 0 0 A x x x x x x x x A (General Call Address) Second Byte ACK © 2006 Microchip Technology Inc. DS22003B-page 17 MCP3421 5.5 High-Speed (HS) Mode The I2C specification requires that a high-speed mode device must be ‘activated’ to operate in high-speed mode. This is done by sending a special address byte of 00001XXX following the START bit. The XXX bits are unique to the High-Speed (HS) mode Master. This byte is referred to as the High-Speed (HS) Master Mode Code (HSMMC). The MCP3421 device does not acknowledge this byte. However, upon receiving this code, the MCP3421 switches on its HS mode filters and communicates up to 3.4 MHz on SDA and SCL. The device will switch out of the HS mode on the next STOP condition. For more information on the HS mode, or other I2C modes, please refer to the Phillips I2C specification. 5.6 I2C Bus Characteristics The I2C specification defines the following bus protocol: • Data transfer may be initiated only when the bus is not busy. • During data transfer, the data line must remain stable whenever the clock line is HIGH. Changes in the data line while the clock line is HIGH will be interpreted as a START or STOP condition. Accordingly, the following bus conditions have been defined using Figure 5-6. 5.6.1 BUS NOT BUSY (A) Both data and clock lines remain HIGH. 5.6.2 START DATA TRANSFER (B) A HIGH to LOW transition of the SDA line while the clock (SCL) is HIGH determines a START condition. All commands must be preceded by a START condition. 5.6.3 STOP DATA TRANSFER (C) A LOW to HIGH transition of the SDA line while the clock (SCL) is HIGH determines a STOP condition. All operations can be ended with a STOP condition. 5.6.4 DATA VALID (D) The state of the data line represents valid data when, after a START condition, the data line is stable for the duration of the HIGH period of the clock signal. The data on the line must be changed during the LOW period of the clock signal. There is one clock pulse per bit of data. Each data transfer is initiated with a START condition and terminated with a STOP condition. 5.6.5 ACKNOWLEDGE The Master (microcontroller) and the slave (MCP3421) use an acknowledge pulse as a hand shake of communication for each byte. The ninth clock pulse of each byte is used for the acknowledgement. The acknowledgement is achieved by pulling-down the SDA line “LOW” during the 9th clock pulse. The clock pulse is always provided by the Master (microcontroller) and the acknowledgement is issued by the receiving device of the byte (Note: The transmitting device must release the SDA line (“HIGH”) during the acknowledge pulse.). For example, the slave (MCP3421) issues the acknowledgement (bring down the SDA line “LOW”) after the end of each receiving byte, and the master (microcontroller) issues the acknowledgement when it reads data from the Slave (MCP3421). When the MCP3421 is addressed, it generates an acknowledge after receiving each byte successfully. The Master device (microcontroller) must provide an extra clock pulse (9th pulse of each byte) for the acknowledgement from the MCP3421 (slave). The MCP3421 (slave) pulls-down the SDA line during the acknowledge clock pulse in such a way that the SDA line is stable low during the high period of the acknowledge clock pulse. During reads, the Master (microcontroller) can terminate the current read operation by not providing an acknowledge bit on the last byte that has been clocked out from the MCP3421. In this case, the MCP3421 releases the SDA line to allow the master (microcontroller) to generate a STOP or repeated START condition. FIGURE 5-6: Data Transfer Sequence on the Serial Bus. SCL SDA (A) (B) (D) (D) (C) (A) START CONDITION ADDRESS OR ACKNOWLEDGE VALID DATA ALLOWED TO CHANGE STOP CONDITION MCP3421 DS22003B-page 18 © 2006 Microchip Technology Inc. TABLE 5-4: I2C SERIAL TIMING SPECIFICATIONS Electrical Specifications: Unless otherwise specified, all limits are specified for TA = -40 to +85°C, VDD = +2.7V, +3.3V or +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. Parameters Sym Min Typ Max Units Conditions Standard Mode Clock frequency fSCL 0 — 100 kHz Clock high time THIGH 4000 — — ns Clock low time TLOW 4700 — — ns SDA and SCL rise time (Note 1) TR — — 1000 ns From VIL to VIH SDA and SCL fall time (Note 1) TF — — 300 ns From VIH to VIL START condition hold time THD:STA 4000 — — ns After this period, the first clock pulse is generated. Repeated START condition setup time TSU:STA 4700 — — ns Only relevant for repeated Start condition Data hold time (Note 3) THD:DAT 0 — 3450 ns Data input setup time TSU:DAT 250 — — ns STOP condition setup time TSU:STO 4000 — — ns STOP condition hold time THD:STD 4000 — — ns Output valid from clock (Notes 2 and 3) TAA 0 — 3750 ns Bus free time TBUF 4700 — — ns Time between START and STOP conditions. Fast Mode Clock frequency TSCL 0 — 400 kHz Clock high time THIGH 600 — — ns Clock low time TLOW 1300 — — ns SDA and SCL rise time (Note 1) TR 20 + 0.1Cb — 300 ns From VIL to VIH SDA and SCL fall time (Note 1) TF 20 + 0.1Cb — 300 ns From VIH to VIL START condition hold time THD:STA 600 — — ns After this period, the first clock pulse is generated Repeated START condition setup time TSU:STA 600 — — ns Only relevant for repeated Start condition Data hold time (Note 4) THD:DAT 0 — 900 ns Data input setup time TSU:DAT 100 — — ns STOP condition setup time TSU:STO 600 — — ns STOP condition hold time THD:STD 600 — — ns Output valid from clock (Notes 2 and 3) TAA 0 — 1200 ns Bus free time TBUF 1300 — — ns Time between START and STOP conditions. Input filter spike suppression (Note 5) TSP 0 — 50 ns SDA and SCL pins Note 1: This parameter is ensured by characterization and not 100% tested. 2: This specification is not a part of the I2C specification. This specification is equivalent to the Data Hold Time (THD:DAT) plus SDA Fall (or rise) time: TAA = THD:DAT + TF (OR TR). 3: If this parameter is too short, it can create an unintended Start or Stop condition to other devices on the bus line. If this parameter is too long, Clock Low time (TLOW) can be affected. 4: For Data Input: This parameter must be longer than tSP. If this parameter is too long, the Data Input Setup (TSU:DAT) or Clock Low time (TLOW) can be affected. For Data Output: This parameter is characterized, and tested indirectly by testing TAA parameter. 5: This parameter is ensured by characterization and not 100% tested. This parameter is not available for Standard Mode. © 2006 Microchip Technology Inc. DS22003B-page 19 MCP3421 High Speed Mode Clock frequency fSCL 0 — 3.4 1.7 MHz MHz Cb = 100 pF Cb = 400 pF Clock high time THIGH 60 120 — — ns ns Cb = 100 pF Cb = 400 pF Clock low time TLOW 160 320 — — ns Cb = 100 pF Cb = 400 pF SCL rise time (Note 1) TR — — 40 80 ns From VIL to VIH,Cb = 100 pF Cb = 400 pF SCL fall time (Note 1) TF — — 40 80 ns From VIH to VIL,Cb = 100 pF Cb = 400 pF SDA rise time (Note 1) TR: DAT — — 80 160 ns From VIL to VIH,Cb = 100 pF Cb = 400 pF SDA fall time (Note 1) TF: DATA — — 80 160 ns From VIH to VIL,Cb = 100 pF Cb = 400 pF START condition hold time THD:STA 160 — — ns After this period, the first clock pulse is generated Repeated START condition setup time TSU:STA 160 — — ns Only relevant for repeated Start condition Data hold time (Note 4) THD:DAT 00 — 70 150 ns Cb = 100 pF Cb = 400 pF Data input setup time TSU:DAT 10 — — ns STOP condition setup time TSU:STO 160 — — ns STOP condition hold time THD:STD 160 — — ns Output valid from clock (Notes 2 and 3) TAA — — 150 310 ns Cb = 100 pF Cb = 400 pF Bus free time TBUF 160 — — ns Time between START and STOP conditions. Input filter spike suppression (Note 5) TSP 0 — 10 ns SDA and SCL pins TABLE 5-4: I2C SERIAL TIMING SPECIFICATIONS (CONTINUED) Electrical Specifications: Unless otherwise specified, all limits are specified for TA = -40 to +85°C, VDD = +2.7V, +3.3V or +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. Parameters Sym Min Typ Max Units Conditions Note 1: This parameter is ensured by characterization and not 100% tested. 2: This specification is not a part of the I2C specification. This specification is equivalent to the Data Hold Time (THD:DAT) plus SDA Fall (or rise) time: TAA = THD:DAT + TF (OR TR). 3: If this parameter is too short, it can create an unintended Start or Stop condition to other devices on the bus line. If this parameter is too long, Clock Low time (TLOW) can be affected. 4: For Data Input: This parameter must be longer than tSP. If this parameter is too long, the Data Input Setup (TSU:DAT) or Clock Low time (TLOW) can be affected. For Data Output: This parameter is characterized, and tested indirectly by testing TAA parameter. 5: This parameter is ensured by characterization and not 100% tested. This parameter is not available for Standard Mode. MCP3421 DS22003B-page 20 © 2006 Microchip Technology Inc. FIGURE 5-7: I2C Bus Timing Data. TF SCL SDA TSU:STA TSP THD:STA TLOW THIGH THD:DAT TAA TSU:DAT TR TSU:STO TBUF © 2006 Microchip Technology Inc. DS22003B-page 21 MCP3421 6.0 BASIC APPLICATION CONFIGURATION The MCP3421 device can be used for various precision analog-to-digital converter applications. The device operates with very simple connections to the application circuit. The following sections discuss the examples of the device connections and applications. 6.1 Connecting to the Application Circuits 6.1.1 INPUT VOLTAGE RANGE The fully differential input signals can be connected to the VIN+ and VIN- input pins. The input range should be within absolute common mode input voltage range: VSS - 0.3V to VDD + 0.3V. Outside this limit, the ESD protection diode at the input pin begins to conduct and the error due to input leakage current increases rapidly. Within this limit, the differential input VIN (= VIN+ - VIN-) is boosted by the PGA before a conversion takes place. The MCP3421 can not accept negative input voltages on the input pins. Figures 6-1 and 6-2 show typical connection examples for differential inputs and a singleended input, respectively. For the single-ended input, the input signal is applied to one of the input pins (typically connected to the VIN+ pin) while the other input pin (typically VIN- pin) is grounded. The input signal range of the single-ended configuration is from 0V to 2.048V. All device characteristics hold for the single-ended configuration, but this configuration loses one bit resolution because the input can only stand in positive half scale. Refer to Section 1.0 “Electrical Characteristics”. 6.1.2 BYPASS CAPACITORS ON VDD PIN For accurate measurement, the application circuit needs a clean supply voltage and must block any noise signal to the MCP3421 device. Figure 6-1 shows an example of using two bypass capacitors (a 10 μF tantalum capacitor and a 0.1 μF ceramic capacitor) in parallel on the VDD line. These capacitors are helpful to filter out any high frequency noises on the VDD line and also provide the momentary bursts of extra currents when the device needs from the supply. These capacitors should be placed as close to the VDD pin as possible (within one inch). If the application circuit has separate digital and analog power supplies, the VDD and VSS of the MCP3421 should reside on the analog plane. 6.1.3 CONNECTING TO I2C BUS USING PULL-UP RESISTORS The SCL and SDA pins of the MCP3421 are open-drain configurations. These pins require a pull-up resistor as shown in Figure 6-1. The value of these pull-up resistors depends on the operating speed (standard, fast, and high speed) and loading capacitance of the I2C bus line. Higher value of pull-up resistor consumes less power, but increases the signal transition time (higher RC time constant) on the bus. Therefore, it can limit the bus operating speed. The lower value of resistor, on the other hand, consumes higher power, but allows higher operating speed. If the bus line has higher capacitance due to long bus line or high number of devices connected to the bus, a smaller pull-up resistor is needed to compensate the long RC time constant. The pull-up resistor is typically chosen between 1 kΩ and 10 kΩ ranges for standard and fast modes, and less than 1 kΩ for high speed mode in high loading capacitance environments. FIGURE 6-1: Typical Connection Example for Differential Inputs. FIGURE 6-2: Typical Connection Example for Single-Ended Input. The number of devices connected to the bus is limited only by the maximum bus capacitance of 400 pF. The bus loading capacitance affects on the bus operating speed. For example, the highest bus operating speed for the 400 pF bus capacitance is 1.7 MHz, and 3.4 MHz for 100 pF. Figure 6-3 shows an example of multiple device connections. MCP3421 VIN+ VINVSS VDD 1 2 3 4 5 6 SCL SDL 0.1 μF 10 μF R R Input Signals VDD VDD TO MCU Note: R is the pull-up resistor. (MASTER) MCP3421 VIN+ VINVSS VDD 1 2 3 4 5 6 SCL SDL 0.1 μF 10 μF R R Input Signals VDD VDD TO MCU Note: R is the pull-up resistor. (MASTER) MCP3421 DS22003B-page 22 © 2006 Microchip Technology Inc. FIGURE 6-3: Example of Multiple Device Connection on I2C Bus. 6.2 Device Connection Test The user can test the presence of the MCP3421 on the I2C bus line without performing an input data conversion. This test can be achieved by checking an acknowledge response from the MCP3421 after sending a read or write command. Here is an example using Figure 6-4: (a) Set the R/W bit “HIGH” in the address byte. (b) The MCP3421 will then acknowledge by pulling SDA bus LOW during the ACK clock and then release the bus back to the I2C Master. (c) A STOP or repeated START bit can then be issued from the Master and I2C communication can continue. FIGURE 6-4: I2C Bus Connection Test. 6.3 Application Examples The MCP3421 device can be used in a broad range of sensor and data acquisition applications. Figure 6-5, shows an example of interfacing with a bridge sensor for pressure measurement. FIGURE 6-5: Example of Pressure Measurement. In this circuit example, the sensor full scale range is ±7.5 mV with a common mode input voltage of VDD / 2. This configuration will provide a full 14-bit resolution across the sensor output range. The alternative circuit for this amount of accuracy would involve an analog gain stage prior to a 16-bit ADC. Figure 6-6 shows an example of temperature measurement using a thermistor. This example can achieve a linear response over a 50°C temperature range. This can be implemented using a standard resistor with 1% tolerance in series with the thermistor. The value of the resistor is selected to be equal to the thermistor value at the mid-point of the desired temperature range. FIGURE 6-6: Example of Temperature Measurement. SDASCL (24LC01) Microcontroller EEPROM MCP3421 (TC74) Temperature Sensor (PIC16F876) SCL 1 2 3 4 5 6 7 8 9 SDA 1 1 0 1 A2 A1 A0 1 Start Bit Address Byte Device bits Address bits R/W Start Bit MCP3421 ACK Response NPP301 MCP3421 VIN+ VINVSS VDD 1 2 3 4 5 6 SCL SDL 0.1 μF 10 μF R R VDD VDD TO MCU (MASTER) VDD 10 kΩ Resistor 10 kΩ Thermistor MCP3421 VIN+ VINVSS VDD 1 2 3 4 5 6 SCL SDL 0.1 μF 10 μF R R VDD VDD TO MCU (MASTER) VDD © 2006 Microchip Technology Inc. DS22003B-page 23 MCP3421 7.0 PACKAGING INFORMATION 7.1 Package Marking Information Legend: XX...X Customer-specific information Y Year code (last digit of calendar year) YY Year code (last 2 digits of calendar year) WW Week code (week of January 1 is week ‘01’) NNN Alphanumeric traceability code Pb-free JEDEC designator for Matte Tin (Sn) * This package is Pb-free. The Pb-free JEDEC designator ( ) can be found on the outer packaging for this package. Note: In the event the full Microchip part number cannot be marked on one line, it will be carried over to the next line, thus limiting the number of available characters for customer-specific information. e3 e3 2 5 3 1 4 6 6-Lead SOT-23 XXNN 2 5 3 1 4 6 Example CA25 MCP3421 DS22003B-page 24 © 2006 Microchip Technology Inc. 6-Lead Plastic Small Outline Transistor (OT) (SOT-23) Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging 1 D B n E E1 L c β φ α A A2 A1 p1 Mold Draft Angle Bottom β 0 5 10 0 5 10 Mold Draft Angle Top α 0 5 10 0 5 10 Lead Width B .014 .017 .020 0.35 0.43 0.50 Lead Thickness c .004 .006 .008 0.09 0.15 0.20 Foot Angle φ 0 5 10 0 5 10 Foot Length L .014 .018 .022 0.35 0.45 0.55 Overall Length D .110 .116 .122 2.80 2.95 3.10 Molded Package Width E1 .059 .064 .069 1.50 1.63 1.75 Overall Width E .102 .110 .118 2.60 2.80 3.00 Standoff A1 .000 .003 .006 0.00 0.08 0.15 Molded Package Thickness A2 .035 .043 .051 0.90 1.10 1.30 Overall Height A .035 .046 .057 0.90 1.18 1.45 Outside lead pitch p1 .075 BSC 1.90 BSC Pitch p .038 BSC 0.95 BSC Number of Pins n 6 6 Dimension Limits MIN NOM MAX MIN NOM MAX Units INCHES* MILLIMETERS Dimensions D and E1 do not include mold flash or protrusions. Mold flash or protrusions shall not exceed .005" (0.127mm) per side. Notes: JEITA (formerly EIAJ) equivalent: SC-74A * Controlling Parameter Drawing No. C04-120 BSC: Basic Dimension. Theoretically exact value shown without tolerances. See ASME Y14.5M Revised 09-12-05 © 2006 Microchip Technology Inc. DS22003B-page 27 MCP3421 APPENDIX A: REVISION HISTORY Revision B (December 2006) • Changes to Electrical Characteristics tables • Added characterization data • Changes to I2C Serial Timing Specification table • Change to Figure 5-7. Revision A (August 2006) • Original Release of this Document. MCP3421 DS22003B-page 28 © 2006 Microchip Technology Inc. NOTES: © 2006 Microchip Technology Inc. DS22003B-page 29 MCP3421 PRODUCT IDENTIFICATION SYSTEM To order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office. Device: MCP3421T: Single Channel ΔΣ A/D Converter (Tape and Reel) Address Options: XX A2 A1 A0 A0 * = 0 0 0 A1 = 0 0 1 A2 = 0 1 0 A3 = 0 1 1 A4 = 1 0 0 A5 = 1 0 1 A6 = 1 1 0 A7 = 1 1 1 * Default option. Contact Microchip factory for other address options Temperature Range: E = -40°C to +125°C Package: OT = Plastic Small Outline Transistor (SOT-23-6), 6-lead Examples: a) MCP3421A0T-E/OT: Tape and Reel, Single Channel ΔΣ A/D Converter, SOT-23-6 package. PART NO. XX X Address Temperature Range Device /XX Package Options MCP3421 DS22003B-page 30 © 2006 Microchip Technology Inc. NOTES: © 2006 Microchip Technology Inc. DS22003B-page 31 Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED, WRITTEN OR ORAL, STATUTORY OR OTHERWISE, RELATED TO THE INFORMATION, INCLUDING BUT NOT LIMITED TO ITS CONDITION, QUALITY, PERFORMANCE, MERCHANTABILITY OR FITNESS FOR PURPOSE. Microchip disclaims all liability arising from this information and its use. Use of Microchip devices in life support and/or safety applications is entirely at the buyer’s risk, and the buyer agrees to defend, indemnify and hold harmless Microchip from any and all damages, claims, suits, or expenses resulting from such use. No licenses are conveyed, implicitly or otherwise, under any Microchip intellectual property rights. Trademarks The Microchip name and logo, the Microchip logo, Accuron, dsPIC, KEELOQ, microID, MPLAB, PIC, PICmicro, PICSTART, PRO MATE, PowerSmart, rfPIC, and SmartShunt are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. AmpLab, FilterLab, Migratable Memory, MXDEV, MXLAB, SEEVAL, SmartSensor and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A. Analog-for-the-Digital Age, Application Maestro, CodeGuard, dsPICDEM, dsPICDEM.net, dsPICworks, ECAN, ECONOMONITOR, FanSense, FlexROM, fuzzyLAB, In-Circuit Serial Programming, ICSP, ICEPIC, Linear Active Thermistor, Mindi, MiWi, MPASM, MPLIB, MPLINK, PICkit, PICDEM, PICDEM.net, PICLAB, PICtail, PowerCal, PowerInfo, PowerMate, PowerTool, REAL ICE, rfLAB, rfPICDEM, Select Mode, Smart Serial, SmartTel, Total Endurance, UNI/O, WiperLock and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. All other trademarks mentioned herein are property of their respective companies. © 2006, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. Printed on recycled paper. Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet. • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. • There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. • Microchip is willing to work with the customer who is concerned about the integrity of their code. • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Microchip received ISO/TS-16949:2002 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona, Gresham, Oregon and Mountain View, California. The Company’s quality system processes and procedures are for its PIC® 8-bit MCUs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. DS22003B-page 32 © 2006 Microchip Technology Inc. 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DS22003B-page 1 MCP3421 Features • 18-bit ΔΣ ADC in a SOT-23-6 package • Differential input operation • Self calibration of Internal Offset and Gain per each conversion • On-board Voltage Reference: - Accuracy: 2.048V ± 0.05% - Drift: 5 ppm/°C • On-board Programmable Gain Amplifier (PGA): - Gains of 1,2,4 or 8 • On-board Oscillator • INL: 10 ppm of FSR (FSR = 4.096V/PGA) • Programmable Data Rate Options: - 3.75 SPS (18 bits) - 15 SPS (16 bits) - 60 SPS (14 bits) - 240 SPS (12 bits) • One-Shot or Continuous Conversion Options • Low current consumption: - 145 μA typical (VDD= 3V, Continuous Conversion) - 39 μA typical (VDD= 3V, One-Shot Conversion with 1 SPS) • Supports I2C Serial Interface: - Standard, Fast and High Speed Modes • Single Supply Operation: 2.7V to 5.5V • Extended Temperature Range: -40°C to 125°C Typical Applications • Portable Instrumentation • Weigh Scales and Fuel Gauges • Temperature Sensing with RTD, Thermistor, and Thermocouple • Bridge Sensing for Pressure, Strain, and Force. Package Types Description The MCP3421 is a single channel low-noise, high accuracy ΔΣ A/D converter with differential inputs and up to 18 bits of resolution in a small SOT-23-6 package. The on-board precision 2.048V reference voltage enables an input range of ±2.048V differentially (Δ voltage = 4.096V). The device uses a two-wire I2C compatible serial interface and operates from a single 2.7V to 5.5V power supply. The MCP3421 device performs conversion at rates of 3.75, 15, 60, or 240 samples per second (SPS) depending on the user controllable configuration bit settings using the two-wire I2C serial interface. This device has an on-board programmable gain amplifier (PGA). The user can select the PGA gain of x1, x2, x4, or x8 before the analog-to-digital conversion takes place. This allows the MCP3421 device to convert a smaller input signal with high resolution. The device has two conversion modes: (a) Continuous mode and (b) One-Shot mode. In One-Shot mode, the device enters a low current standby mode automatically after one conversion. This reduces current consumption greatly during idle periods. The MCP3421 device can be used for various high accuracy analog-to-digital data conversion applications where design simplicity, low power, and small footprint are major considerations. Block Diagram 1 2 3 4 5 VIN+ 6 VSS SCL VINVDD SDA SOT-23-6 Top View VSS VDD VIN+ VINSCL SDA Voltage Reference Clock (2.048V) I2C Interface Gain = 1, 2, 4, or 8 VREF ΔΣ ADC PGA Converter Oscillator 18-Bit Analog-to-Digital Converter with I2C Interface and On-Board Reference MCP3421 DS22003B-page 2 © 2006 Microchip Technology Inc. 1.0 ELECTRICAL CHARACTERISTICS Absolute Maximum Ratings† VDD...................................................................................7.0V All inputs and outputs w.r.t VSS ............... –0.3V to VDD+0.3V Differential Input Voltage ...................................... |VDD - VSS| Output Short Circuit Current .................................Continuous Current at Input Pins ....................................................±2 mA Current at Output and Supply Pins ............................±10 mA Storage Temperature.....................................-65°C to +150°C Ambient Temp. with power applied ...............-55°C to +125°C ESD protection on all pins ................ ≥ 6 kV HBM, ≥ 400V MM Maximum Junction Temperature (TJ) ..........................+150°C †Notice: Stresses above those listed under “Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at those or any other conditions above those indicated in the operational listings of this specification is not implied. Exposure to maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS Electrical Specifications: Unless otherwise specified, all parameters apply for TA = -40°C to +85°C, VDD = +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. All ppm units use 2*VREF as full-scale range. Parameters Sym Min Typ Max Units Conditions Analog Inputs Differential Input Range — ±2.048/PGA — V VIN = VIN+ - VINCommon- Mode Voltage Range (absolute) (Note 1) VSS-0.3 — VDD+0.3 V Differential Input Impedance (Note 2) ZIND (f) — 2.25/PGA — MΩ During normal mode operation Common Mode input Impedance ZINC (f) — 25 — MΩ PGA = 1, 2, 4, 8 System Performance Resolution and No Missing Codes (Note 8) 12 — — Bits DR = 240 SPS 14 — — Bits DR = 60 SPS 16 — — Bits DR = 15 SPS 18 — — Bits DR = 3.75 SPS Data Rate (Note 3) DR 176 240 328 SPS S1,S0 = ‘00’, (12 bits mode) 44 60 82 SPS S1,S0 = ‘01’, (14 bits mode) 11 15 20.5 SPS S1,S0 = ‘10’, (16 bits mode) 2.75 3.75 5.1 SPS S1,S0 = ‘11’, (18 bits mode) Output Noise — 1.5 — μVRMS TA = 25°C, DR = 3.75 SPS, PGA = 1, VIN = 0 Note 1: Any input voltage below or greater than this voltage causes leakage current through the ESD diodes at the input pins. This parameter is ensured by characterization and not 100% tested. 2: This input impedance is due to 3.2 pF internal input sampling capacitor. 3: The total conversion speed includes auto-calibration of offset and gain. 4: INL is the difference between the endpoints line and the measured code at the center of the quantization band. 5: Includes all errors from on-board PGA and VREF. 6: Full Scale Range (FSR) = 2 x 2.048/PGA = 4.096/PGA. 7: This parameter is ensured by characterization and not 100% tested. 8: This parameter is ensured by design and not 100% tested. © 2006 Microchip Technology Inc. DS22003B-page 3 MCP3421 Integral Nonlinearity (Note 4) INL — 10 35 ppm of FSR DR = 3.75 SPS (Note 6) Internal Reference Voltage VREF — 2.048 — V Gain Error (Note 5) — 0.05 0.35 % PGA = 1, DR = 3.75 SPS PGA Gain Error Match (Note 5) — 0.1 — % Between any 2 PGA gains Gain Error Drift (Note 5) — 5 40 ppm/°C PGA=1, DR=3.75 SPS Offset Error VOS — 15 40 μV Tested at PGA = 1 VDD = 5.0V and DR = 3.75 SPS Offset Drift vs. Temperature — 50 — nV/°C VDD = 5.0V Common-Mode Rejection — 105 — dB at DC and PGA =1, — 110 — dB at DC and PGA =8, TA = +25°C Gain vs. VDD — 5 — ppm/V TA = +25°C, VDD = 2.7V to 5.5V, PGA = 1 Power Supply Rejection at DC — 100 — dB TA = +25°C, VDD = 2.7V to 5.5V, PGA = 1 Power Requirements Voltage Range VDD 2.7 — 5.5 V Supply Current during Conversion IDDA — 155 190 μA VDD = 5.0V — 145 — μA VDD = 3.0V Supply Current during Standby Mode IDDS — 0.1 0.5 μA I2C Digital Inputs and Digital Outputs High level input voltage VIH 0.7 VDD — VDD V Low level input voltage VIL — — 0.3VDD V Low level output voltage VOL — — 0.4 V IOL = 3 mA, VDD = +5.0V Hysteresis of Schmitt Trigger for inputs (Note 7) VHYST 0.05VDD — — V fSCL = 100 kHz Supply Current when I2C bus line is active IDDB — — 10 μA Input Leakage Current IILH — — 1 μA VIH = 5.5V IILL -1 — — μA VIL = GND Pin Capacitance and I2C Bus Capacitance Pin capacitance CPIN — — 10 pF I2C Bus Capacitance Cb — — 400 pF Thermal Characteristics Specified Temperature Range TA -40 — +85 °C Operating Temperature Range TA -40 — +125 °C Storage Temperature Range TA -65 — +150 °C ELECTRICAL CHARACTERISTICS (CONTINUED) Electrical Specifications: Unless otherwise specified, all parameters apply for TA = -40°C to +85°C, VDD = +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. All ppm units use 2*VREF as full-scale range. Parameters Sym Min Typ Max Units Conditions Note 1: Any input voltage below or greater than this voltage causes leakage current through the ESD diodes at the input pins. This parameter is ensured by characterization and not 100% tested. 2: This input impedance is due to 3.2 pF internal input sampling capacitor. 3: The total conversion speed includes auto-calibration of offset and gain. 4: INL is the difference between the endpoints line and the measured code at the center of the quantization band. 5: Includes all errors from on-board PGA and VREF. 6: Full Scale Range (FSR) = 2 x 2.048/PGA = 4.096/PGA. 7: This parameter is ensured by characterization and not 100% tested. 8: This parameter is ensured by design and not 100% tested. MCP3421 DS22003B-page 4 © 2006 Microchip Technology Inc. 2.0 TYPICAL PERFORMANCE CURVES Note: Unless otherwise indicated, TA = -40°C to +85°C, VDD = +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. FIGURE 2-1: INL vs. Supply Voltage (VDD). FIGURE 2-2: INL vs. Temperature. FIGURE 2-3: Offset Error vs. Temperature. FIGURE 2-4: Noise vs. Input Voltage. FIGURE 2-5: Total Error vs. Input Voltage. FIGURE 2-6: Gain Error vs. Temperature. Note: The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. .000 .001 .002 .003 .004 .005 2.5 3 3.5 4 4.5 5 5.5 VDD (V) PGA = 1 PGA = 2 PGA = 8 PGA = 4 Integral Nonlinearity (% of FSR) 0 0.001 0.002 0.003 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (oC) Integral Nonlinearity (% of FSR) VDD = 5 V VDD = 2.7V PGA = 1 -20 -15 -10 -5 0 5 10 15 20 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (°C) Offset Error (μV) VDD = 5V PGA = 1 PGA = 2 PGA = 4 PGA = 8 0.0 2.5 5.0 7.5 10.0 -100 -75 -50 -25 0 25 50 75 100 Input Voltage (% of Full-Scale) Noise (μV, rms) PGA = 1 PGA = 2 PGA = 8 PGA = 4 TA = +25°C VDD = 5V -3.0 -2.0 -1.0 0.0 1.0 2.0 3.0 -100 -75 -50 -25 0 25 50 75 100 Input Voltage (% of Full-Scale) Total Error (mV) PGA = 1 PGA = 2 PGA = 8 PGA = 4 -0.4 -0.3 -0.2 -0.1 0 0.1 0.2 0.3 0.4 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (°C) Gain Error (% of FSR) VDD = 5.0V PGA = 1 PGA = 2 PGA = 8 PGA = 4 © 2006 Microchip Technology Inc. DS22003B-page 5 MCP3421 Note: Unless otherwise indicated, TA = -40°C to +85°C, VDD = +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. FIGURE 2-7: IDDA vs. Temperature. FIGURE 2-8: IDDS vs. Temperature. FIGURE 2-9: IDDB vs. Temperature. FIGURE 2-10: OSC Drift vs. Temperature. FIGURE 2-11: Frequency Response. 100 120 140 160 180 200 220 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (oC) IDDA (μA) VDD = 5V VDD = 2.7V 0 100 200 300 400 500 600 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (oC) IDDS (nA) VDD = 2.7V VDD = 5V 0 1 2 3 4 5 6 7 8 9 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (oC) IDDB (A) VDD = 5V VDD = 4.5V VDD = 3.3V VDD = 2.7V -1 0 1 2 3 4 5 -60 -40 -20 0 20 40 60 80 100 120 140 Temperature (°C) Oscillator Drift (%) VDD = 5.0V VDD = 2.7V Data Rate = 3.75 SPS -120 -110 -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 0 0.1 1 10 100 1000 10000 Input Signal Frequency (Hz) Magnitude (dB) 0.1 1 10 100 1k 10k MCP3421 DS22003B-page 6 © 2006 Microchip Technology Inc. 3.0 PIN DESCRIPTIONS TABLE 3-1: PIN FUNCTION TABLE 3.1 Analog Inputs (VIN+, VIN-) VIN+ and VIN- are differential signal input pins. The MCP3421 device accepts a fully differential analog input signal which is connected on the VIN+ and VINinput pins. The differential voltage that is converted is defined by VIN = (VIN+ - VIN-) where VIN+ is the voltage applied at the VIN+ pin and VIN- is the voltage applied at the VIN- pin. The input signal level is amplified by the programmable gain amplifier (PGA) before the conversion. The differential input voltage should not exceed an absolute of (2* VREF/PGA) for accurate measurement, where VREF is the internal reference voltage (2.048V) and PGA is the PGA gain setting. The converter output code will saturate if the input range exceeds (2* VREF/PGA). The absolute voltage range on each of the differential input pins is from VSS-0.3V to VDD+0.3V. Any voltage above or below this range will cause leakage currents through the Electrostatic Discharge (ESD) diodes at the input pins. This ESD current can cause unexpected performance of the device. The common mode of the analog inputs should be chosen such that both the differential analog input range and the absolute voltage range on each pin are within the specified operating range defined in Section 1.0 “Electrical Characteristics” and Section 4.0 “Description of Device Operation”. 3.2 Supply Voltage (VDD, VSS) VDD is the power supply pin for the device. This pin requires an appropriate bypass capacitor of about 0.1 μF (ceramic) to ground. An additional 10 μF capacitor (tantalum) in parallel is also recommended to further attenuate high frequency noise present in some application boards. The supply voltage (VDD) must be maintained in the 2.7V to 5.5V range for specified operation. VSS is the ground pin and the current return path of the device. The user must connect the VSS pin to a ground plane through a low impedance connection. If an analog ground path is available in the application PCB (printed circuit board), it is highly recommended that the VSS pin be tied to the analog ground path or isolated within an analog ground plane of the circuit board. 3.3 Serial Clock Pin (SCL) SCL is the serial clock pin of the I2C interface. The MCP3421 acts only as a slave and the SCL pin accepts only external serial clocks. The input data from the Master device is shifted into the SDA pin on the rising edges of the SCL clock and output from the MCP3421 occurs at the falling edges of the SCL clock. The SCL pin is an open-drain N-channel driver. Therefore, it needs a pull-up resistor from the VDD line to the SCL pin. Refer to Section 5.3 “I2C Serial Communications” for more details of I2C Serial Interface communication. 3.4 Serial Data Pin (SDA) SDA is the serial data pin of the I2C interface. The SDA pin is used for input and output data. In read mode, the conversion result is read from the SDA pin (output). In write mode, the device configuration bits are written (input) though the SDA pin. The SDA pin is an opendrain N-channel driver. Therefore, it needs a pull-up resistor from the VDD line to the SDA pin. Except for start and stop conditions, the data on the SDA pin must be stable during the high period of the clock. The high or low state of the SDA pin can only change when the clock signal on the SCL pin is low. Refer to Section 5.3 “I2C Serial Communications” for more details of I2C Serial Interface communication. Pin No Sym Function 1 VIN+ Non-Inverting Analog Input Pin 2 VSS Ground Pin 3 SCL Serial Clock Input Pin of the I2C Interface 4 SDA Bidirectional Serial Data Pin of the I2C Interface 5 VDD Positive Supply Voltage Pin 6 VIN- Inverting Analog Input Pin © 2006 Microchip Technology Inc. DS22003B-page 7 MCP3421 4.0 DESCRIPTION OF DEVICE OPERATION 4.1 General Overview The MCP3421 is a low-power, 18-Bit Delta-Sigma A/D converter with an I2C serial interface. The device contains an on-board voltage reference (2.048V), programmable gain amplifier (PGA), and internal oscillator. The user can select 12, 14, 16, or 18 bit conversion by setting the configuration register bits. The device can be operated in Continuous Conversion or One-Shot Conversion mode. In the Continuous Conversion mode, the device converts the inputs continuously. While in the One-Shot Conversion mode, the device converts the input one time and stays in the low-power standby mode until it receives another command for a new conversion. During the standby mode, the device consumes less than 0.1 μA typical. 4.2 Power-On-Reset (POR) The device contains an internal Power-On-Reset (POR) circuit that monitors power supply voltage (VDD) during operation. This circuit ensures correct device start-up at system power-up and power-down events. The POR has built-in hysteresis and a timer to give a high degree of immunity to potential ripples and noises on the power supply. A 0.1 μF decoupling capacitor should be mounted as close as possible to the VDD pin for additional transient immunity. The threshold voltage is set at 2.2V with a tolerance of approximately ±5%. If the supply voltage falls below this threshold, the device will be held in a reset condition. The typical hysteresis value is approximately 200 mV. The POR circuit is shut-down during the low-power standby mode. Once a power-up event has occurred, the device requires additional delay time (approximately 300 μs) before a conversion can take place. During this time, all internal analog circuitries are settled before the first conversion occurs. Figure 4-1 illustrates the conditions for power-up and power-down events under typical start-up conditions. When the device powers up, it automatically resets and sets the configuration bits to default settings. The default configuration bit conditions are a PGA gain of 1 V/V and a conversion speed of 240 SPS in Continuous Conversion mode. When the device receives an I2C General Call Reset command, it performs an internal reset similar to a Power-On-Reset event. FIGURE 4-1: POR Operation. 4.3 Internal Voltage Reference The device contains an on-board 2.048V voltage reference. This reference voltage is for internal use only and not directly measurable. The specifications of the reference voltage are part of the device’s gain and drift specifications. Therefore, there is no separate specification for the on-board reference. 4.4 Analog Input Channel The differential analog input channel has a switched capacitor structure. The internal sampling capacitor (3.2 pF) is charged and discharged to process a conversion. The charging and discharging of the input sampling capacitor creates dynamic input currents at the VIN+ and VIN- input pins, which is inversely proportional to the internal sampling capacitor and internal frequency. The current is also a function of the differential input voltages. Care must be taken in setting the common-mode voltage and input voltage ranges so that the input limits do not exceed the ranges specified in Section 1.0 “Electrical Characteristics”. 4.5 Digital Output Code The digital output code produced by the MCP3421 is a function of PGA gain, input signal, and internal reference voltage. In a fixed setting, the digital output code is proportional to the voltage difference between the two analog inputs. The output data format is a binary two’s complement. With this code scheme, the MSB can be considered a sign indicator. When the MSB is a logic ‘0’, it indicates a positive value. When the MSB is a logic ‘1’, it indicates a negative value. The following is an example of the output code: (a) for a negative full-scale input voltage: 100...000 (b) for a zero differential input voltage: 000...000 (c) for a positive full-scale input voltage: 011...111. The MSB is always transmitted first through the serial port. The number of data bits for each conversion is 18, 16, 14, or 12 bits depending on the conversion mode selection. VDD 2.2V 2.0V 300 μS Reset Start-up Normal Operation Reset Time MCP3421 DS22003B-page 8 © 2006 Microchip Technology Inc. The output codes will not roll-over if the input voltage exceeds the maximum input range. In this case, the code will be locked at 0111...11 for all voltages greater than +(VREF - 1 LSB) and 1000...00 for voltages less than -VREF. Table 4-2 shows an example of output codes of various input levels using 18 bit conversion mode. Table 4-3 shows an example of minimum and maximum codes for each data rate option. The output code is given by: EQUATION 4-1: The LSB of the code is given by: EQUATION 4-2: TABLE 4-1: LSB SIZE OF VARIOUS BIT CONVERSION MODES TABLE 4-2: EXAMPLE OF OUTPUT CODE FOR 18 BITS TABLE 4-3: MINIMUM AND MAXIMUM CODES 4.6 Self-Calibration The device performs a self-calibration of offset and gain for each conversion. This provides reliable conversion results from conversion-to-conversion over variations in temperature as well as power supply fluctuations. 4.7 Input Impedance The MCP3421 uses a switched-capacitor input stage using a 3.2 pF sampling capacitor. This capacitor is switched (charged and discharged) at a rate of the sampling frequency that is generated by the on-board clock. The differential mode impedance varies with the PGA settings. The typical differential input impedance during a normal mode operation is given by: Since the sampling capacitor is only switching to the input pins during a conversion process, the above input impedance is only valid during conversion periods. In a low power standby mode, the above impedance is not presented at the input pins. Therefore, only a leakage current due to ESD diode is presented at the input pins. The conversion accuracy can be affected by the input signal source impedance when any external circuit is connected to the input pins. The source impedance adds to the internal impedance and directly affects the time required to charge the internal sampling capacitor. Therefore, a large input source impedance connected to the input pins can increase the system performance errors such as offset, gain, and integral nonlinearity (INL) errors. Ideally, the input source impedance should be zero. This can be achievable by using an operational amplifier with a closed-loop output impedance of tens of ohms. Bit Resolutions LSB (V) 12 bits 1 mV 14 bits 250 μV 16 bits 62.5 μV 18 bits 15.625 μV Input Voltage (V) Digital Code ≥ VREF 011111111111111111 VREF - 1 LSB 011111111111111111 2 LSB 000000000000000010 1 LSB 000000000000000001 0 000000000000000000 -1 LSB 111111111111111111 -2 LSB 111111111111111110 - VREF 100000000000000000 < -VREF 100000000000000000 Output Code (Max Code + 1) (VIN+ – VIN-) 2.048V = × -------------------------------------- LSB 2 × 2.048V 2N = -------------------------- Where: N = the number of bits Number of Bits Data Rate Minimum Code Maximum Code 12 240 SPS -2048 2047 14 60 SPS -8192 8191 16 15 SPS -32768 32767 18 3.75 SPS -131072 131071 Note: Maximum n-bit code = 2n-1 - 1 Minimum n-bit code = -1 x 2n-1 ZIN(f) = 2.25 MΩ/PGA © 2006 Microchip Technology Inc. DS22003B-page 9 MCP3421 4.8 Aliasing and Anti-aliasing Filter Aliasing occurs when the input signal contains timevarying signal components with frequency greater than half the sample rate. In the aliasing conditions, the device can output unexpected output codes. For applications that are operating in electrical noise environments, the time-varying signal noise or high frequency interference components can be easily added to the input signals and cause aliasing. Although the MCP3421 device has an internal first order sinc filter, its’ filter response may not give enough attenuation to all aliasing signal components. To avoid the aliasing, an external anti-aliasing filter, which can be accomplished with a simple RC low-pass filter, is typically used at the input pins. The low-pass filter cuts off the high frequency noise components and provides a band-limited input signal to the MCP3421 input pins. MCP3421 DS22003B-page 10 © 2006 Microchip Technology Inc. 5.0 USING THE MCP3421 DEVICE 5.1 Operating Modes The user operates the device by setting up the device configuration register and reads the conversion data using serial I2C interface commands. The MCP3421 operates in two modes: (a) Continuous Conversion Mode or (b) One-Shot Conversion Mode (single conversion). The selection is made by setting the O/C bit in the Configuration Register. Refer to Section 5.2 “Configuration Register” for more information. 5.1.1 CONTINUOUS CONVERSION MODE (O/C BIT = 1) The MCP3421 device performs a Continuous Conversion if the O/C bit is set to logic “high”. Once the conversion is completed, the result is placed at the output data register. The device immediately begins another conversion and overwrites the output data register with the most recent data. The device also clears the data ready flag (RDY bit = 0) when the conversion is completed. The device sets the ready flag bit (RDY bit = 1), if the latest conversion result has been read by the Master. 5.1.2 ONE-SHOT CONVERSION MODE (O/C BIT = 0) Once the One-Shot Conversion (single conversion) Mode is selected, the device performs a conversion, updates the Output Data register, clears the data ready flag (RDY = 0), and then enters a low power standby mode. A new One-Shot Conversion is started again when the device receives a new write command with RDY = 1. This One-Shot Conversion Mode is recommended for low power operating applications. During the low current standby mode, the device consumes less than 1 μA typical. For example, if user collects 18 bit conversion data once a second in One-Shot Conversion mode, the device draws only about one fourth of its total operating current. In this example, the device consumes approximately 39 μA (= ~145 μA/3.75 SPS), if the device performs only one conversion per second (1 SPS) in 18-bit conversion mode with 3V power supply. © 2006 Microchip Technology Inc. DS22003B-page 11 MCP3421 5.2 Configuration Register The MCP3421 has an 8-bit wide configuration register to select for: PGA gain, conversion rate, and conversion mode. This register allows the user to change the operating condition of the device and check the status of the device operation. The user can rewrite the configuration byte any time during the device operation. Register 5-1 shows the configuration register bits. REGISTER 5-1: CONFIGURATION REGISTER R/W-1 R/W-0 R/W-0 R/W-1 R/W-0 R/W-0 R/W-0 R/W-0 RDY C1 C0 O/C S1 S0 G1 G0 1 * 0 * 0 * 1 * 0 * 0 * 0 * 0 * bit 7 bit 0 * Default Configuration after Power-On Reset Legend: R = Readable bit W = Writable bit U = Unimplemented bit, read as ‘0’ -n = Value at POR ‘1’ = Bit is set ‘0’ = Bit is cleared x = Bit is unknown bit 7 RDY: Ready Bit This bit is the data ready flag. In read mode, this bit indicates if the output register has been updated with a new conversion. In One-Shot Conversion mode, writing this bit to “1” initiates a new conversion. Reading RDY bit with the read command: 1 = Output register has not been updated. 0 = Output register has been updated with the latest conversion data. Writing RDY bit with the write command: Continuous Conversion mode: No effect One-Shot Conversion mode: 1 = Initiate a new conversion. 0 = No effect. bit 6-5 C1-C0: Channel Selection Bits These are the Channel Selection bits, but not used in the MCP3421 device. bit 4 O/C: Conversion Mode Bit 1 = Continuous Conversion Mode. Once this bit is selected, the device performs data conversions continuously. 0 = One-Shot Conversion Mode. The device performs a single conversion and enters a low power standby mode until it receives another write/read command. bit 3-2 S1-S0: Sample Rate Selection Bit 00 = 240 SPS (12 bits), 01 = 60 SPS (14 bits), 10 = 15 SPS (16 bits), 11 = 3.75 SPS (18 bits) bit 1-0 G1-G0: PGA Gain Selector Bits 00 = 1 V/V, 01 = 2 V/V, 10 = 4 V/V, 11 = 8 V/V MCP3421 DS22003B-page 12 © 2006 Microchip Technology Inc. In read mode, the RDY bit in the configuration byte indicates the state of the conversion: (a) RDY = 1 indicates that the data bytes that have just been read were not updated from the previous conversion. (b) RDY = 0 indicates that the data bytes that have just been read were updated. If the configuration byte is read repeatedly by clocking continuously after the first read (i.e., after the 5th byte in the 18-bit conversion mode), the state of the RDY bit indicates whether the device is ready with new conversion data. See Figure 5-2. For example, RDY = 0 means new conversion data is ready for reading. In this case, the user can send a stop bit to exit the current read operation and send a new read command to read out updated conversion data. See Figures 5-2 and 5-3 for reading conversion data. The user can rewrite the configuration byte any time for a new setting. Tables 5-1 and 5-2 show the examples of the configuration bit operation. 5.3 I2C Serial Communications The MCP3421 device communicates with Master (microcontroller) through a serial I2C (Inter-Integrated Circuit) interface and supports standard (100 kbits/sec), fast (400 kbits/sec) and high-speed (3.4 Mbits/sec) modes. The serial I2C is a bidirectional 2-wire data bus communication protocol using opendrain SCL and SDA lines. The MCP3421 can only be addressed as a slave. Once addressed, it can receive configuration bits or transmit the latest conversion results. The serial clock pin (SCL) is an input only and the serial data pin (SDA) is bidirectional. An example of a hardware connection diagram is shown in Figure 6-1. The Master starts communication by sending a START bit and terminates the communication by sending a STOP bit. The first byte after the START bit is always the address byte of the device, which includes the device code, the address bits, and the R/W bit. The device code for the MCP3421 device is 1101. The address bits (A2, A1, A0) are pre-programmed at the factory. In general, the address bits are specified by the customer when they order the device. The three address bits are programmed to “000” at the factory, if they are not specified by the customer. Figure 5-1 shows the details of the MCP3421 address byte. During a low power standby mode, SDA and SCL pins remain at a floating condition. More details of the I2C bus characteristic is described in Section 5.6 “I2C Bus Characteristics”. 5.3.1 DEVICE ADDRESSING The address byte is the first byte received following the START condition from the Master device. The MCP3421 device code is 1101. The device code is followed by three address bits (A2, A1, A0) which are programmed at the factory. The three address bits allow up to eight MCP3421 devices on the same data bus line. The (R/W) bit determines if the Master device wants to read the conversion data or write to the Configuration register. If the (R/W) bit is set (read mode), the MCP3421 outputs the conversion data in the following clocks. If the (R/W) bit is cleared (write mode), the MCP3421 expects a configuration byte in the following clocks. When the MCP3421 receives the correct address byte, it outputs an acknowledge bit after the R/W bit. Figure 5-1 shows the MCP3421 address byte. See Figures 5-2 and 5-3 for the read and write operations of the device. TABLE 5-1: CONFIGURATION BITS FOR WRITING R/W O/C RDY Operation 0 0 0 No effect if all other bits remain the same - operation continues with the previous settings 0 0 1 Initiate One-Shot Conversion 0 1 0 Initiate Continuous Conversion 0 1 1 Initiate Continuous Conversion TABLE 5-2: CONFIGURATION BITS FOR READING R/W O/C RDY Operation 1 0 0 New conversion data in One- Shot conversion mode has been just read. The RDY bit remains low until set by a new write command. 1 0 1 One-Shot Conversion is in progress, The conversion data is not updated yet. The RDY bit stays high. 1 1 0 New conversion data in Continuous Conversion mode has been just read. The RDY bit changes to high after this read. 1 1 1 The conversion data in Continuous Conversion mode was already read. The latest conversion data is not ready. The RDY bit stays high until a new conversion is completed. © 2006 Microchip Technology Inc. DS22003B-page 13 MCP3421 FIGURE 5-1: MCP3421 Address Byte. 5.3.2 READING DATA FROM THE DEVICE When the Master sends a read command (R/W = 1), the MCP3421 outputs the conversion data bytes and configuration byte. Each byte consists of 8 bits with one acknowledge (ACK) bit. The ACK bit after the address byte is issued by the MCP3421 and the ACK bits after each conversion data bytes are issued by the Master. When the device is configured for 18-bit conversion mode, the device outputs three data bytes followed by a configuration byte. The first 7 data bits in the first data byte are the MSB of the conversion data. The user can ignore the first 6 data bits, and take the 7th data bit (D17) as the MSB of the conversion data. The LSB of the 3rd data byte is the LSB of the conversion data (D0). If the device is configured for 12, 14, or 16 bit-mode, the device outputs two data bytes followed by a configuration byte. In 16 bit-conversion mode, the MSB of the first data byte is the MSB (D15) of the conversion data. In 14-bit conversion mode, the first two bits in the first data byte can be ignored (they are the MSB of the conversion data), and the 3rd bit (D13) is the MSB of the conversion data. In 12-bit conversion mode, the first four bits can be ignored (they are the MSB of the conversion data), and the 5th bit (D11) of the byte represents the MSB of the conversion data. Table 5-3 shows an example of the conversion data output of each conversion mode. The configuration byte follows the output data byte. The device outputs the configuration byte as long as the SCL pulses are received. The device terminates the current outputs when it receives a Not-Acknowledge (NAK), a repeated start or a stop bit at any time during the output bit stream. It is not required to read the configuration byte. However, the user may read the configuration byte to check the RDY bit condition to confirm whether the just received data bytes are updated conversion data. The user may continuously send clock (SCL) to repeatedly read the configuration bytes to check the RDY bit status. Figures 5-2 and 5-3 show the timing diagrams of the reading. 5.3.3 WRITING A CONFIGURATION BYTE TO THE DEVICE When the Master sends an address byte with the R/W bit low (R/W = 0), the MCP3421 expects one configuration byte following the address. Any byte sent after this second byte will be ignored. The user can change the operating mode of the device by writing the configuration register bits. If the device receives a write command with a new configuration setting, the device immediately begins a new conversion and updates the conversion data. Start bit Read/Write bit Address Byte R/W ACK 1 1 0 1 X X X Device Code Address Bits (Note 1) Address Acknowledge bit Address Note 1: Specified by customer and programmed at the factory. If not specified by the customer, programmed to ‘000’. TABLE 5-3: EXAMPLE OF CONVERSION DATA OUTPUT OF EACH CONVERSION MODE Conversion Mode Conversion Data Output 18-bits MMMMMMMD16 (1st data byte) - D15 ~ D8 (2nd data byte) - D7 ~ D0 (3rd data byte) - Configuration byte 16-bits MD14~D8 (1st data byte) - D7 ~ D0 (2nd data byte) - Configuration byte 14-bits MMMD12~D8 (1st data byte) - D7 ~ D0 (2nd data byte) - Configuration byte 12-bits MMMMMD10D9D8 (1st data byte) - D7 ~ D0 (2nd data byte) - Configuration byte Note: M is MSB of the data byte. MCP3421 DS22003B-page 14 © 2006 Microchip Technology Inc. FIGURE 5-2: Timing Diagram For Reading From The MCP3421 With 18-Bit Mode. 1 9 1 9 1 9 1 9 1 9 1 9 1 1 0 1 A2 A1 A0 D ACK by RDY O/C MCP3421 7 Start Bit by R/W Master Repeat of D17 (MSB) 2nd Byte Upper Data Byte (Data on Clocks 1-6th can be ignored) ACK by Master ACK by Master ACK by Master ACK by Master 17 D 16 D 15 D 14 D 13 D 12 D 11 D 10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0 C1 C0 S1 S0 G1 G0 1st Byte MCP3421 Address Byte 3rd Byte Middle Data Byte 4th Byte Lower Data Byte 5th Byte Configuration Byte (Optional) C1 C0 S1 S0 G1 G0 NAK by Master Stop Bit by Master (Optional) Nth Repeated Byte: Configuration Byte Note: – MCP3421 device code is 1101. – Address Bits A2- A0 = 000 are programmed at the factory unless customer requests specific codes. – Stop bit or NAK bit can be issued any time during reading. – Data bits on clocks 1 - 6th in 2nd byte are repeated MSB and can be ignored. SCL SDA RDY O/C © 2006 Microchip Technology Inc. DS22003B-page 15 MCP3421 FIGURE 5-3: Timing Diagram For Reading From The MCP3421 With 12-Bit to 16-Bit Modes. 1 1 0 1 A2 A1 A0 ACK by MCP3421 Start Bit by Master 2nd Byte Middle Data Byte ACK by Master ACK by Master ACK by Master D 15 D 14 D 13 D 12 D 11 D 10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0 C1 C0 S1 S0 G1 G0 1st Byte MCP3421 Address Byte 3rd Byte Lower Data Byte 4th Byte Configuration Byte (Optional) C1 C0 S1 S0 G1 G0 NAK by Master Stop Bit by Master (Optional) Nth Repeated Byte: Configuration Byte Note: – MCP3421 device code is 1101. – Address Bits A2- A0 = 000 are programmed at the factory unless customer requests specific codes. – Stop bit or NAK bit can be issued any time during reading. – In 14 - bit mode: D15 and D14 are repeated MSB and can be ignored. – In 12 - bit mode: D15 - D12 are repeated MSB and can be ignored. 1 9 1 9 1 9 1 9 SCL SDA 1 9 R/W RDY O/C RDY O/C MCP3421 DS22003B-page 16 © 2006 Microchip Technology Inc. FIGURE 5-4: Timing Diagram For Writing To The MCP3421. 5.4 General Call The MCP3421 acknowledges the general call address (0x00 in the first byte). The meaning of the general call address is always specified in the second byte. Refer to Figure 5-5. The MCP3421 supports the following general calls: 5.4.1 GENERAL CALL RESET The general call reset occurs if the second byte is ‘00000110’ (06h). At the acknowledgement of this byte, the device will abort current conversion and perform an internal reset similar to a power-on-reset (POR). 5.4.2 GENERAL CALL CONVERSION The general call conversion occurs if the second byte is ‘00001000’ (08h). All devices on the bus initiate a conversion simultaneously. For the MCP3421 device, the configuration will be set to the One-Shot Conversion mode and a single conversion will be performed. The PGA and data rate settings are unchanged with this general call. FIGURE 5-5: General Call Address Format. For more information on the general call, or other I2C modes, please refer to the Phillips I2C specification. 1 9 1 9 Stop Bit by 1 1 0 1 A2 A1 A0 R/W ACK by MCP3421 RDY C1 C0 O/C S1 S0 G1 G0 1st Byte: 2nd Byte: Master ACK by MCP3421 MCP3421 Address Byte Configuration Byte Start Bit by Master with Write command Note: – Stop bit can be issued any time during writing. – MCP3421 device code is 1101. – Address Bits A2- A0 = 000 are programmed at factory unless customer requests different codes. SCL SDA Note: The I2C specification does not allow to use “00000000” (00h) in the second byte. LSB First Byte ACK 0 0 0 0 0 0 0 0 A x x x x x x x x A (General Call Address) Second Byte ACK © 2006 Microchip Technology Inc. DS22003B-page 17 MCP3421 5.5 High-Speed (HS) Mode The I2C specification requires that a high-speed mode device must be ‘activated’ to operate in high-speed mode. This is done by sending a special address byte of 00001XXX following the START bit. The XXX bits are unique to the High-Speed (HS) mode Master. This byte is referred to as the High-Speed (HS) Master Mode Code (HSMMC). The MCP3421 device does not acknowledge this byte. However, upon receiving this code, the MCP3421 switches on its HS mode filters and communicates up to 3.4 MHz on SDA and SCL. The device will switch out of the HS mode on the next STOP condition. For more information on the HS mode, or other I2C modes, please refer to the Phillips I2C specification. 5.6 I2C Bus Characteristics The I2C specification defines the following bus protocol: • Data transfer may be initiated only when the bus is not busy. • During data transfer, the data line must remain stable whenever the clock line is HIGH. Changes in the data line while the clock line is HIGH will be interpreted as a START or STOP condition. Accordingly, the following bus conditions have been defined using Figure 5-6. 5.6.1 BUS NOT BUSY (A) Both data and clock lines remain HIGH. 5.6.2 START DATA TRANSFER (B) A HIGH to LOW transition of the SDA line while the clock (SCL) is HIGH determines a START condition. All commands must be preceded by a START condition. 5.6.3 STOP DATA TRANSFER (C) A LOW to HIGH transition of the SDA line while the clock (SCL) is HIGH determines a STOP condition. All operations can be ended with a STOP condition. 5.6.4 DATA VALID (D) The state of the data line represents valid data when, after a START condition, the data line is stable for the duration of the HIGH period of the clock signal. The data on the line must be changed during the LOW period of the clock signal. There is one clock pulse per bit of data. Each data transfer is initiated with a START condition and terminated with a STOP condition. 5.6.5 ACKNOWLEDGE The Master (microcontroller) and the slave (MCP3421) use an acknowledge pulse as a hand shake of communication for each byte. The ninth clock pulse of each byte is used for the acknowledgement. The acknowledgement is achieved by pulling-down the SDA line “LOW” during the 9th clock pulse. The clock pulse is always provided by the Master (microcontroller) and the acknowledgement is issued by the receiving device of the byte (Note: The transmitting device must release the SDA line (“HIGH”) during the acknowledge pulse.). For example, the slave (MCP3421) issues the acknowledgement (bring down the SDA line “LOW”) after the end of each receiving byte, and the master (microcontroller) issues the acknowledgement when it reads data from the Slave (MCP3421). When the MCP3421 is addressed, it generates an acknowledge after receiving each byte successfully. The Master device (microcontroller) must provide an extra clock pulse (9th pulse of each byte) for the acknowledgement from the MCP3421 (slave). The MCP3421 (slave) pulls-down the SDA line during the acknowledge clock pulse in such a way that the SDA line is stable low during the high period of the acknowledge clock pulse. During reads, the Master (microcontroller) can terminate the current read operation by not providing an acknowledge bit on the last byte that has been clocked out from the MCP3421. In this case, the MCP3421 releases the SDA line to allow the master (microcontroller) to generate a STOP or repeated START condition. FIGURE 5-6: Data Transfer Sequence on the Serial Bus. SCL SDA (A) (B) (D) (D) (C) (A) START CONDITION ADDRESS OR ACKNOWLEDGE VALID DATA ALLOWED TO CHANGE STOP CONDITION MCP3421 DS22003B-page 18 © 2006 Microchip Technology Inc. TABLE 5-4: I2C SERIAL TIMING SPECIFICATIONS Electrical Specifications: Unless otherwise specified, all limits are specified for TA = -40 to +85°C, VDD = +2.7V, +3.3V or +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. Parameters Sym Min Typ Max Units Conditions Standard Mode Clock frequency fSCL 0 — 100 kHz Clock high time THIGH 4000 — — ns Clock low time TLOW 4700 — — ns SDA and SCL rise time (Note 1) TR — — 1000 ns From VIL to VIH SDA and SCL fall time (Note 1) TF — — 300 ns From VIH to VIL START condition hold time THD:STA 4000 — — ns After this period, the first clock pulse is generated. Repeated START condition setup time TSU:STA 4700 — — ns Only relevant for repeated Start condition Data hold time (Note 3) THD:DAT 0 — 3450 ns Data input setup time TSU:DAT 250 — — ns STOP condition setup time TSU:STO 4000 — — ns STOP condition hold time THD:STD 4000 — — ns Output valid from clock (Notes 2 and 3) TAA 0 — 3750 ns Bus free time TBUF 4700 — — ns Time between START and STOP conditions. Fast Mode Clock frequency TSCL 0 — 400 kHz Clock high time THIGH 600 — — ns Clock low time TLOW 1300 — — ns SDA and SCL rise time (Note 1) TR 20 + 0.1Cb — 300 ns From VIL to VIH SDA and SCL fall time (Note 1) TF 20 + 0.1Cb — 300 ns From VIH to VIL START condition hold time THD:STA 600 — — ns After this period, the first clock pulse is generated Repeated START condition setup time TSU:STA 600 — — ns Only relevant for repeated Start condition Data hold time (Note 4) THD:DAT 0 — 900 ns Data input setup time TSU:DAT 100 — — ns STOP condition setup time TSU:STO 600 — — ns STOP condition hold time THD:STD 600 — — ns Output valid from clock (Notes 2 and 3) TAA 0 — 1200 ns Bus free time TBUF 1300 — — ns Time between START and STOP conditions. Input filter spike suppression (Note 5) TSP 0 — 50 ns SDA and SCL pins Note 1: This parameter is ensured by characterization and not 100% tested. 2: This specification is not a part of the I2C specification. This specification is equivalent to the Data Hold Time (THD:DAT) plus SDA Fall (or rise) time: TAA = THD:DAT + TF (OR TR). 3: If this parameter is too short, it can create an unintended Start or Stop condition to other devices on the bus line. If this parameter is too long, Clock Low time (TLOW) can be affected. 4: For Data Input: This parameter must be longer than tSP. If this parameter is too long, the Data Input Setup (TSU:DAT) or Clock Low time (TLOW) can be affected. For Data Output: This parameter is characterized, and tested indirectly by testing TAA parameter. 5: This parameter is ensured by characterization and not 100% tested. This parameter is not available for Standard Mode. © 2006 Microchip Technology Inc. DS22003B-page 19 MCP3421 High Speed Mode Clock frequency fSCL 0 — 3.4 1.7 MHz MHz Cb = 100 pF Cb = 400 pF Clock high time THIGH 60 120 — — ns ns Cb = 100 pF Cb = 400 pF Clock low time TLOW 160 320 — — ns Cb = 100 pF Cb = 400 pF SCL rise time (Note 1) TR — — 40 80 ns From VIL to VIH,Cb = 100 pF Cb = 400 pF SCL fall time (Note 1) TF — — 40 80 ns From VIH to VIL,Cb = 100 pF Cb = 400 pF SDA rise time (Note 1) TR: DAT — — 80 160 ns From VIL to VIH,Cb = 100 pF Cb = 400 pF SDA fall time (Note 1) TF: DATA — — 80 160 ns From VIH to VIL,Cb = 100 pF Cb = 400 pF START condition hold time THD:STA 160 — — ns After this period, the first clock pulse is generated Repeated START condition setup time TSU:STA 160 — — ns Only relevant for repeated Start condition Data hold time (Note 4) THD:DAT 00 — 70 150 ns Cb = 100 pF Cb = 400 pF Data input setup time TSU:DAT 10 — — ns STOP condition setup time TSU:STO 160 — — ns STOP condition hold time THD:STD 160 — — ns Output valid from clock (Notes 2 and 3) TAA — — 150 310 ns Cb = 100 pF Cb = 400 pF Bus free time TBUF 160 — — ns Time between START and STOP conditions. Input filter spike suppression (Note 5) TSP 0 — 10 ns SDA and SCL pins TABLE 5-4: I2C SERIAL TIMING SPECIFICATIONS (CONTINUED) Electrical Specifications: Unless otherwise specified, all limits are specified for TA = -40 to +85°C, VDD = +2.7V, +3.3V or +5.0V, VSS = 0V, VIN+ = VIN- = VREF/2. Parameters Sym Min Typ Max Units Conditions Note 1: This parameter is ensured by characterization and not 100% tested. 2: This specification is not a part of the I2C specification. This specification is equivalent to the Data Hold Time (THD:DAT) plus SDA Fall (or rise) time: TAA = THD:DAT + TF (OR TR). 3: If this parameter is too short, it can create an unintended Start or Stop condition to other devices on the bus line. If this parameter is too long, Clock Low time (TLOW) can be affected. 4: For Data Input: This parameter must be longer than tSP. If this parameter is too long, the Data Input Setup (TSU:DAT) or Clock Low time (TLOW) can be affected. For Data Output: This parameter is characterized, and tested indirectly by testing TAA parameter. 5: This parameter is ensured by characterization and not 100% tested. This parameter is not available for Standard Mode. MCP3421 DS22003B-page 20 © 2006 Microchip Technology Inc. FIGURE 5-7: I2C Bus Timing Data. TF SCL SDA TSU:STA TSP THD:STA TLOW THIGH THD:DAT TAA TSU:DAT TR TSU:STO TBUF © 2006 Microchip Technology Inc. DS22003B-page 21 MCP3421 6.0 BASIC APPLICATION CONFIGURATION The MCP3421 device can be used for various precision analog-to-digital converter applications. The device operates with very simple connections to the application circuit. The following sections discuss the examples of the device connections and applications. 6.1 Connecting to the Application Circuits 6.1.1 INPUT VOLTAGE RANGE The fully differential input signals can be connected to the VIN+ and VIN- input pins. The input range should be within absolute common mode input voltage range: VSS - 0.3V to VDD + 0.3V. Outside this limit, the ESD protection diode at the input pin begins to conduct and the error due to input leakage current increases rapidly. Within this limit, the differential input VIN (= VIN+ - VIN-) is boosted by the PGA before a conversion takes place. The MCP3421 can not accept negative input voltages on the input pins. Figures 6-1 and 6-2 show typical connection examples for differential inputs and a singleended input, respectively. For the single-ended input, the input signal is applied to one of the input pins (typically connected to the VIN+ pin) while the other input pin (typically VIN- pin) is grounded. The input signal range of the single-ended configuration is from 0V to 2.048V. All device characteristics hold for the single-ended configuration, but this configuration loses one bit resolution because the input can only stand in positive half scale. Refer to Section 1.0 “Electrical Characteristics”. 6.1.2 BYPASS CAPACITORS ON VDD PIN For accurate measurement, the application circuit needs a clean supply voltage and must block any noise signal to the MCP3421 device. Figure 6-1 shows an example of using two bypass capacitors (a 10 μF tantalum capacitor and a 0.1 μF ceramic capacitor) in parallel on the VDD line. These capacitors are helpful to filter out any high frequency noises on the VDD line and also provide the momentary bursts of extra currents when the device needs from the supply. These capacitors should be placed as close to the VDD pin as possible (within one inch). If the application circuit has separate digital and analog power supplies, the VDD and VSS of the MCP3421 should reside on the analog plane. 6.1.3 CONNECTING TO I2C BUS USING PULL-UP RESISTORS The SCL and SDA pins of the MCP3421 are open-drain configurations. These pins require a pull-up resistor as shown in Figure 6-1. The value of these pull-up resistors depends on the operating speed (standard, fast, and high speed) and loading capacitance of the I2C bus line. Higher value of pull-up resistor consumes less power, but increases the signal transition time (higher RC time constant) on the bus. Therefore, it can limit the bus operating speed. The lower value of resistor, on the other hand, consumes higher power, but allows higher operating speed. If the bus line has higher capacitance due to long bus line or high number of devices connected to the bus, a smaller pull-up resistor is needed to compensate the long RC time constant. The pull-up resistor is typically chosen between 1 kΩ and 10 kΩ ranges for standard and fast modes, and less than 1 kΩ for high speed mode in high loading capacitance environments. FIGURE 6-1: Typical Connection Example for Differential Inputs. FIGURE 6-2: Typical Connection Example for Single-Ended Input. The number of devices connected to the bus is limited only by the maximum bus capacitance of 400 pF. The bus loading capacitance affects on the bus operating speed. For example, the highest bus operating speed for the 400 pF bus capacitance is 1.7 MHz, and 3.4 MHz for 100 pF. Figure 6-3 shows an example of multiple device connections. MCP3421 VIN+ VINVSS VDD 1 2 3 4 5 6 SCL SDL 0.1 μF 10 μF R R Input Signals VDD VDD TO MCU Note: R is the pull-up resistor. (MASTER) MCP3421 VIN+ VINVSS VDD 1 2 3 4 5 6 SCL SDL 0.1 μF 10 μF R R Input Signals VDD VDD TO MCU Note: R is the pull-up resistor. (MASTER) MCP3421 DS22003B-page 22 © 2006 Microchip Technology Inc. FIGURE 6-3: Example of Multiple Device Connection on I2C Bus. 6.2 Device Connection Test The user can test the presence of the MCP3421 on the I2C bus line without performing an input data conversion. This test can be achieved by checking an acknowledge response from the MCP3421 after sending a read or write command. Here is an example using Figure 6-4: (a) Set the R/W bit “HIGH” in the address byte. (b) The MCP3421 will then acknowledge by pulling SDA bus LOW during the ACK clock and then release the bus back to the I2C Master. (c) A STOP or repeated START bit can then be issued from the Master and I2C communication can continue. FIGURE 6-4: I2C Bus Connection Test. 6.3 Application Examples The MCP3421 device can be used in a broad range of sensor and data acquisition applications. Figure 6-5, shows an example of interfacing with a bridge sensor for pressure measurement. FIGURE 6-5: Example of Pressure Measurement. In this circuit example, the sensor full scale range is ±7.5 mV with a common mode input voltage of VDD / 2. This configuration will provide a full 14-bit resolution across the sensor output range. The alternative circuit for this amount of accuracy would involve an analog gain stage prior to a 16-bit ADC. Figure 6-6 shows an example of temperature measurement using a thermistor. This example can achieve a linear response over a 50°C temperature range. This can be implemented using a standard resistor with 1% tolerance in series with the thermistor. The value of the resistor is selected to be equal to the thermistor value at the mid-point of the desired temperature range. FIGURE 6-6: Example of Temperature Measurement. SDASCL (24LC01) Microcontroller EEPROM MCP3421 (TC74) Temperature Sensor (PIC16F876) SCL 1 2 3 4 5 6 7 8 9 SDA 1 1 0 1 A2 A1 A0 1 Start Bit Address Byte Device bits Address bits R/W Start Bit MCP3421 ACK Response NPP301 MCP3421 VIN+ VINVSS VDD 1 2 3 4 5 6 SCL SDL 0.1 μF 10 μF R R VDD VDD TO MCU (MASTER) VDD 10 kΩ Resistor 10 kΩ Thermistor MCP3421 VIN+ VINVSS VDD 1 2 3 4 5 6 SCL SDL 0.1 μF 10 μF R R VDD VDD TO MCU (MASTER) VDD © 2006 Microchip Technology Inc. DS22003B-page 23 MCP3421 7.0 PACKAGING INFORMATION 7.1 Package Marking Information Legend: XX...X Customer-specific information Y Year code (last digit of calendar year) YY Year code (last 2 digits of calendar year) WW Week code (week of January 1 is week ‘01’) NNN Alphanumeric traceability code Pb-free JEDEC designator for Matte Tin (Sn) * This package is Pb-free. The Pb-free JEDEC designator ( ) can be found on the outer packaging for this package. Note: In the event the full Microchip part number cannot be marked on one line, it will be carried over to the next line, thus limiting the number of available characters for customer-specific information. e3 e3 2 5 3 1 4 6 6-Lead SOT-23 XXNN 2 5 3 1 4 6 Example CA25 MCP3421 DS22003B-page 24 © 2006 Microchip Technology Inc. 6-Lead Plastic Small Outline Transistor (OT) (SOT-23) Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging 1 D B n E E1 L c β φ α A A2 A1 p1 Mold Draft Angle Bottom β 0 5 10 0 5 10 Mold Draft Angle Top α 0 5 10 0 5 10 Lead Width B .014 .017 .020 0.35 0.43 0.50 Lead Thickness c .004 .006 .008 0.09 0.15 0.20 Foot Angle φ 0 5 10 0 5 10 Foot Length L .014 .018 .022 0.35 0.45 0.55 Overall Length D .110 .116 .122 2.80 2.95 3.10 Molded Package Width E1 .059 .064 .069 1.50 1.63 1.75 Overall Width E .102 .110 .118 2.60 2.80 3.00 Standoff A1 .000 .003 .006 0.00 0.08 0.15 Molded Package Thickness A2 .035 .043 .051 0.90 1.10 1.30 Overall Height A .035 .046 .057 0.90 1.18 1.45 Outside lead pitch p1 .075 BSC 1.90 BSC Pitch p .038 BSC 0.95 BSC Number of Pins n 6 6 Dimension Limits MIN NOM MAX MIN NOM MAX Units INCHES* MILLIMETERS Dimensions D and E1 do not include mold flash or protrusions. Mold flash or protrusions shall not exceed .005" (0.127mm) per side. Notes: JEITA (formerly EIAJ) equivalent: SC-74A * Controlling Parameter Drawing No. C04-120 BSC: Basic Dimension. Theoretically exact value shown without tolerances. See ASME Y14.5M Revised 09-12-05 © 2006 Microchip Technology Inc. DS22003B-page 27 MCP3421 APPENDIX A: REVISION HISTORY Revision B (December 2006) • Changes to Electrical Characteristics tables • Added characterization data • Changes to I2C Serial Timing Specification table • Change to Figure 5-7. Revision A (August 2006) • Original Release of this Document. MCP3421 DS22003B-page 28 © 2006 Microchip Technology Inc. NOTES: © 2006 Microchip Technology Inc. DS22003B-page 29 MCP3421 PRODUCT IDENTIFICATION SYSTEM To order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office. Device: MCP3421T: Single Channel ΔΣ A/D Converter (Tape and Reel) Address Options: XX A2 A1 A0 A0 * = 0 0 0 A1 = 0 0 1 A2 = 0 1 0 A3 = 0 1 1 A4 = 1 0 0 A5 = 1 0 1 A6 = 1 1 0 A7 = 1 1 1 * Default option. Contact Microchip factory for other address options Temperature Range: E = -40°C to +125°C Package: OT = Plastic Small Outline Transistor (SOT-23-6), 6-lead Examples: a) MCP3421A0T-E/OT: Tape and Reel, Single Channel ΔΣ A/D Converter, SOT-23-6 package. PART NO. XX X Address Temperature Range Device /XX Package Options MCP3421 DS22003B-page 30 © 2006 Microchip Technology Inc. NOTES: © 2006 Microchip Technology Inc. DS22003B-page 31 Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED, WRITTEN OR ORAL, STATUTORY OR OTHERWISE, RELATED TO THE INFORMATION, INCLUDING BUT NOT LIMITED TO ITS CONDITION, QUALITY, PERFORMANCE, MERCHANTABILITY OR FITNESS FOR PURPOSE. Microchip disclaims all liability arising from this information and its use. Use of Microchip devices in life support and/or safety applications is entirely at the buyer’s risk, and the buyer agrees to defend, indemnify and hold harmless Microchip from any and all damages, claims, suits, or expenses resulting from such use. No licenses are conveyed, implicitly or otherwise, under any Microchip intellectual property rights. Trademarks The Microchip name and logo, the Microchip logo, Accuron, dsPIC, KEELOQ, microID, MPLAB, PIC, PICmicro, PICSTART, PRO MATE, PowerSmart, rfPIC, and SmartShunt are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. AmpLab, FilterLab, Migratable Memory, MXDEV, MXLAB, SEEVAL, SmartSensor and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A. Analog-for-the-Digital Age, Application Maestro, CodeGuard, dsPICDEM, dsPICDEM.net, dsPICworks, ECAN, ECONOMONITOR, FanSense, FlexROM, fuzzyLAB, In-Circuit Serial Programming, ICSP, ICEPIC, Linear Active Thermistor, Mindi, MiWi, MPASM, MPLIB, MPLINK, PICkit, PICDEM, PICDEM.net, PICLAB, PICtail, PowerCal, PowerInfo, PowerMate, PowerTool, REAL ICE, rfLAB, rfPICDEM, Select Mode, Smart Serial, SmartTel, Total Endurance, UNI/O, WiperLock and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. All other trademarks mentioned herein are property of their respective companies. © 2006, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. Printed on recycled paper. Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet. • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. • There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. • Microchip is willing to work with the customer who is concerned about the integrity of their code. • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Microchip received ISO/TS-16949:2002 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona, Gresham, Oregon and Mountain View, California. The Company’s quality system processes and procedures are for its PIC® 8-bit MCUs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. DS22003B-page 32 © 2006 Microchip Technology Inc. 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Manila Tel: 63-2-634-9065 Fax: 63-2-634-9069 Singapore Tel: 65-6334-8870 Fax: 65-6334-8850 Taiwan - Hsin Chu Tel: 886-3-572-9526 Fax: 886-3-572-6459 Taiwan - Kaohsiung Tel: 886-7-536-4818 Fax: 886-7-536-4803 Taiwan - Taipei Tel: 886-2-2500-6610 Fax: 886-2-2508-0102 Thailand - Bangkok Tel: 66-2-694-1351 Fax: 66-2-694-1350 EUROPE Austria - Wels Tel: 43-7242-2244-39 Fax: 43-7242-2244-393 Denmark - Copenhagen Tel: 45-4450-2828 Fax: 45-4485-2829 France - Paris Tel: 33-1-69-53-63-20 Fax: 33-1-69-30-90-79 Germany - Munich Tel: 49-89-627-144-0 Fax: 49-89-627-144-44 Italy - Milan Tel: 39-0331-742611 Fax: 39-0331-466781 Netherlands - Drunen Tel: 31-416-690399 Fax: 31-416-690340 Spain - Madrid Tel: 34-91-708-08-90 Fax: 34-91-708-08-91 UK - Wokingham Tel: 44-118-921-5869 Fax: 44-118-921-5820 WORLDWIDE SALES AND SERVICE 10/19/06 Features • High-performance, Low-power AVR® 8-bit Microcontroller • RISC Architecture – 118 Powerful Instructions – Most Single Clock Cycle Execution – 32 x 8 General Purpose Working Registers – Fully Static Operation – Up to 16 MIPS Throughput at 16 MHz • Data and Non-volatile Program Memory – 2K Bytes of In-System Programmable Program Memory Flash Endurance: 10,000 Write/Erase Cycles – 128 Bytes of In-System Programmable EEPROM Endurance: 100,000 Write/Erase Cycles – 128 Bytes Internal SRAM – Programming Lock for Flash Program and EEPROM Data Security • Peripheral Features – 8-bit Timer/Counter with Separate Prescaler – 8-bit High-speed Timer with Separate Prescaler 2 High Frequency PWM Outputs with Separate Output Compare Registers Non-overlapping Inverted PWM Output Pins – Universal Serial Interface with Start Condition Detector – 10-bit ADC 11 Single Ended Channels 8 Differential ADC Channels 7 Differential ADC Channel Pairs with Programmable Gain (1x, 20x) – On-chip Analog Comparator – External Interrupt – Pin Change Interrupt on 11 Pins – Programmable Watchdog Timer with Separate On-chip Oscillator • Special Microcontroller Features – Low Power Idle, Noise Reduction, and Power-down Modes – Power-on Reset and Programmable Brown-out Detection – External and Internal Interrupt Sources – In-System Programmable via SPI Port – Internal Calibrated RC Oscillator • I/O and Packages – 20-lead PDIP/SOIC: 16 Programmable I/O Lines – 32-lead QFN/MLF: 16 programmable I/O Lines • Operating Voltages – 2.7V - 5.5V for ATtiny26L – 4.5V - 5.5V for ATtiny26 • Speed Grades – 0 - 8 MHz for ATtiny26L – 0 - 16 MHz for ATtiny26 • Power Consumption at 1 MHz, 3V and 25°C for ATtiny26L – Active 16 MHz, 5V and 25°C: Typ 15 mA – Active 1 MHz, 3V and 25°C: 0.70 mA – Idle Mode 1 MHz, 3V and 25°C: 0.18 mA – Power-down Mode: < 1 μA 8-bit Microcontroller with 2K Bytes Flash ATtiny26 ATtiny26L Summary 1477KS–AVR–08/10 2 1477KS–AVR–08/10 ATtiny26(L) Pin Configuration Note: The bottom pad under the QFN/MLF package should be soldered to ground. 1 2 3 4 5 6 7 8 9 10 20 19 18 17 16 15 14 13 12 11 (MOSI/DI/SDA/OC1A) PB0 (MISO/DO/OC1A) PB1 (SCK/SCL/OC1B) PB2 (OC1B) PB3 VCC GND (ADC7/XTAL1) PB4 (ADC8/XTAL2) PB5 (ADC9/INT0/T0) PB6 (ADC10/RESET) PB7 PA0 (ADC0) PA1 (ADC1) PA2 (ADC2) PA3 (AREF) GND AVCC PA4 (ADC3) PA5 (ADC4) PA6 (ADC5/AIN0) PA7 (ADC6/AIN1) PDIP/SOIC 1 2 3 4 5 6 7 8 24 23 22 21 20 19 18 17 32 31 30 29 28 27 26 25 9 10 11 12 13 14 15 16 MLF Top View NC (OC1B) PB3 NC VCC GND NC (ADC7/XTAL1) PB4 (ADC8/XTAL2) PB5 NC PA2 (ADC2) PA3 (AREF) GND NC NC AVCC PA4 (ADC3) NC (ADC9/INT0/T0) PB6 (ADC10/RESET) PB7 NC (ADC6/AIN1) PA7 (ADC5/AIN0) PA6 (ADC4) PA5 NC PB2 (SCK/SCL/OC1B) PB1 (MISO/DO/OC1A) PB0 (MOSI/DI/SDA/OC1A) NC NC NC PA0 (ADC0) PA1 (ADC1) 3 1477KS–AVR–08/10 ATtiny26(L) Description The ATtiny26(L) is a low-power CMOS 8-bit microcontroller based on the AVR enhanced RISC architecture. By executing powerful instructions in a single clock cycle, the ATtiny26(L) achieves throughputs approaching 1 MIPS per MHz allowing the system designer to optimize power consumption versus processing speed. The AVR core combines a rich instruction set with 32 general purpose working registers. All the 32 registers are directly connected to the Arithmetic Logic Unit (ALU), allowing two independent registers to be accessed in one single instruction executed in one clock cycle. The resulting architecture is more code efficient while achieving throughputs up to ten times faster than conventional CISC microcontrollers. The ATtiny26(L) has a high precision ADC with up to 11 single ended channels and 8 differential channels. Seven differential channels have an optional gain of 20x. Four out of the seven differential channels, which have the optional gain, can be used at the same time. The ATtiny26(L) also has a high frequency 8-bit PWM module with two independent outputs. Two of the PWM outputs have inverted non-overlapping output pins ideal for synchronous rectification. The Universal Serial Interface of the ATtiny26(L) allows efficient software implementation of TWI (Two-wire Serial Interface) or SM-bus interface. These features allow for highly integrated battery charger and lighting ballast applications, low-end thermostats, and firedetectors, among other applications. The ATtiny26(L) provides 2K bytes of Flash, 128 bytes EEPROM, 128 bytes SRAM, up to 16 general purpose I/O lines, 32 general purpose working registers, two 8-bit Timer/Counters, one with PWM outputs, internal and external Oscillators, internal and external interrupts, programmable Watchdog Timer, 11-channel, 10-bit Analog to Digital Converter with two differential voltage input gain stages, and four software selectable power saving modes. The Idle mode stops the CPU while allowing the Timer/Counters and interrupt system to continue functioning. The ATtiny26(L) also has a dedicated ADC Noise Reduction mode for reducing the noise in ADC conversion. In this sleep mode, only the ADC is functioning. The Power-down mode saves the register contents but freezes the oscillators, disabling all other chip functions until the next interrupt or hardware reset. The Standby mode is the same as the Power-down mode, but external oscillators are enabled. The wakeup or interrupt on pin change features enable the ATtiny26(L) to be highly responsive to external events, still featuring the lowest power consumption while in the Power-down mode. The device is manufactured using Atmel’s high density non-volatile memory technology. By combining an enhanced RISC 8-bit CPU with Flash on a monolithic chip, the ATtiny26(L) is a powerful microcontroller that provides a highly flexible and cost effective solution to many embedded control applications. The ATtiny26(L) AVR is supported with a full suite of program and system development tools including: Macro assemblers, program debugger/simulators, In-circuit emulators, and evaluation kits. 4 1477KS–AVR–08/10 ATtiny26(L) Block Diagram Figure 1. The ATtiny26(L) Block Diagram WATCHDOG TIMER MCU CONTROL REGISTER UNIVERSAL SERIAL INTERFACE TIMER/ COUNTER0 DATA DIR. REG.PORT A DATA REGISTER PORT A PROGRAMMING LOGIC TIMING AND CONTROL TIMER/ COUNTER1 MCU STATUS REGISTER PORT A DRIVERS PA0-PA7 VCC GND + - ANALOG COMPARATOR 8-BIT DATA BUS ADC ISP INTERFACE INTERRUPT UNIT EEPROM INTERNAL OSCILLATOR OSCILLATORS CALIBRATED OSCILLATOR INTERNAL DATA DIR. REG.PORT B DATA REGISTER PORT B PORT B DRIVERS PB0-PB7 PROGRAM COUNTER STACK POINTER PROGRAM FLASH SRAM GENERAL PURPOSE REGISTERS INSTRUCTION REGISTER INSTRUCTION DECODER STATUS REGISTER Z Y X ALU CONTROL LINES AVCC 5 1477KS–AVR–08/10 ATtiny26(L) Pin Descriptions VCC Digital supply voltage pin. GND Digital ground pin. AVCC AVCC is the supply voltage pin for Port A and the A/D Converter (ADC). It should be externally connected to VCC, even if the ADC is not used. If the ADC is used, it should be connected to VCC through a low-pass filter. See page 94 for details on operating of the ADC. Port A (PA7..PA0) Port A is an 8-bit general purpose I/O port. PA7..PA0 are all I/O pins that can provide internal pull-ups (selected for each bit). Port A has alternate functions as analog inputs for the ADC and analog comparator and pin change interrupt as described in “Alternate Port Functions” on page 46. Port B (PB7..PB0) Port B is an 8-bit general purpose I/O port. PB6..0 are all I/O pins that can provide internal pullups (selected for each bit). PB7 is an I/O pin if not used as the reset. To use pin PB7 as an I/O pin, instead of RESET pin, program (“0”) RSTDISBL Fuse. Port B has alternate functions for the ADC, clocking, timer counters, USI, SPI programming, and pin change interrupt as described in “Alternate Port Functions” on page 46. An External Reset is generated by a low level on the PB7/RESET pin. Reset pulses longer than 50 ns will generate a reset, even if the clock is not running. Shorter pulses are not guaranteed to generate a reset. XTAL1 Input to the inverting oscillator amplifier and input to the internal clock operating circuit. XTAL2 Output from the inverting oscillator amplifier. 6 1477KS–AVR–08/10 ATtiny26(L) General Information Resources A comprehensive set of development tools, application notes and datasheets are available for download on http://www.atmel.com/avr. Code Examples This datasheet contains simple code examples that briefly show how to use various parts of the device. These code examples assume that the part specific header file is included before compilation. Be aware that not all C compiler vendors include bit definitions in the header files and interrupt handling in C is compiler dependent. Please confirm with the C compiler documentation for more details. 7 1477KS–AVR–08/10 ATtiny26(L) Register Summary Address Name Bit 7 Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1 Bit 0 Page $3F ($5F) SREG I T H S V N Z C 10 $3E ($5E) Reserved $3D ($5D) SP SP7 SP6 SP5 SP4 SP3 SP2 SP1 SP0 11 $3C ($5C) Reserved $3B ($5B) GIMSK - INT0 PCIE1 PCIE0 - - - - 58 $3A ($5A) GIFR - INTF0 PCIF - - - - - 59 $39 ($59) TIMSK - OCIE1A OCIE1B - - TOIE1 TOIE0 - 59 $38 ($58) TIFR - OCF1A OCF1B - - TOV1 TOV0 - 60 $37 ($57) Reserved $36 ($56) Reserved $35 ($55) MCUCR - PUD SE SM1 SM0 - ISC01 ISC00 37 $34 ($54) MCUSR - - - - WDRF BORF EXTRF PORF 36 $33 ($53) TCCR0 - - - - PSR0 CS02 CS01 CS00 66 $32 ($52) TCNT0 Timer/Counter0 (8-Bit) 67 $31 ($51) OSCCAL Oscillator Calibration Register 29 $30 ($50) TCCR1A COM1A1 COM1A0 COM1B1 COM1B0 FOC1A FOC1B PWM1A PWM1B 70 $2F ($4F) TCCR1B CTC1 PSR1 - - CS13 CS12 CS11 CS10 71 $2E ($4E) TCNT1 Timer/Counter1 (8-Bit) 72 $2D ($4D) OCR1A Timer/Counter1 Output Compare Register A (8-Bit) 72 $2C ($4C) OCR1B Timer/Counter1 Output Compare Register B (8-Bit) 73 $2B ($4B) OCR1C Timer/Counter1 Output Compare Register C (8-Bit) 73 $2A ($4A) Reserved $29 ($49) PLLCSR - - - - - PCKE PLLE PLOCK $28 ($48) Reserved $27 ($47) Reserved $26 ($46) Reserved $25 ($45) Reserved $24 ($44) Reserved $23 ($43) Reserved $22 ($42) Reserved $21 ($41) WDTCR - - - WDCE WDE WDP2 WDP1 WDP0 78 $20 ($40) Reserved $1F ($3F) Reserved $1E ($3E) EEAR - EEAR6 EEAR5 EEAR4 EEAR3 EEAR2 EEAR1 EEAR0 18 $1D ($3D) EEDR EEPROM Data Register (8-Bit) 19 $1C ($3C) EECR - - - - EERIE EEMWE EEWE EERE 19 $1B ($3B) PORTA PORTA7 PORTA6 PORTA5 PORTA4 PORTA3 PORTA2 PORTA1 PORTA0 $1A ($3A) DDRA DDA7 DDA6 DDA5 DDA4 DDA3 DDA2 DDA1 DDA0 $19 ($39) PINA PINA7 PINA6 PINA5 PINA4 PINA3 PINA2 PINA1 PINA0 $18 ($38) PORTB PORTB7 PORTB6 PORTB5 PORTB4 PORTB3 PORTB2 PORTB1 PORTB0 $17 ($37) DDRB DDB7 DDB6 DDB5 DDB4 DDB3 DDB2 DDB1 DDB0 $16 ($36) PINB PINB7 PINB6 PINB5 PINB4 PINB3 PINB2 PINB1 PINB0 $15 ($35) Reserved $14 ($34) Reserved $13 ($33) Reserved $12 ($32) Reserved $11 ($31) Reserved $10 ($30) Reserved $0F ($2F) USIDR Universal Serial Interface Data Register (8-Bit) 81 $0E ($2E) USISR USISIF USIOIF USIPF USIDC USICNT3 USICNT2 USICNT1 USICNT0 81 $0D ($2D) USICR USISIE USIOIE USIWM1 USIWM0 USICS1 USICS0 USICLK USITC 82 $0C ($2C) Reserved $0B ($2)B Reserved $0A ($2A) Reserved $09 ($29) Reserved $08 ($28) ACSR ACD ACBG ACO ACI ACIE ACME ACIS1 ACIS0 91 $07 ($27) ADMUX REFS1 REFS0 ADLAR MUX4 MUX3 MUX2 MUX1 MUX0 101 $06 ($26) ADCSR ADEN ADSC ADFR ADIF ADIE ADPS2 ADPS1 ADPS0 103 $05 ($25) ADCH ADC Data Register High Byte 104 $04 ($24) ADCL ADC Data Register Low Byte 104 … Reserved $00 ($20) Reserved 8 1477KS–AVR–08/10 ATtiny26(L) Instruction Set Summary Mnemonic Operands Description Operation Flags # Clocks ARITHMETIC AND LOGIC INSTRUCTIONS ADD Rd, Rr Add Two Registers Rd ← Rd + Rr Z,C,N,V,H 1 ADC Rd, Rr Add with Carry Two Registers Rd ← Rd + Rr + C Z,C,N,V,H 1 ADIW Rdl, K Add Immediate to Word Rdh:Rdl ← Rdh:Rdl + K Z,C,N,V,S 2 SUB Rd, Rr Subtract Two Registers Rd ← Rd - Rr Z,C,N,V,H 1 SUBI Rd, K Subtract Constant from Register Rd ← Rd - K Z,C,N,V,H 1 SBC Rd, Rr Subtract with Carry Two Registers Rd ← Rd - Rr - C Z,C,N,V,H 1 SBCI Rd, K Subtract with Carry Constant from Reg. Rd ← Rd - K - C Z,C,N,V,H 1 SBIW Rdl, K Subtract Immediate from Word Rdh:Rdl ← Rdh:Rdl - K Z,C,N,V,S 2 AND Rd, Rr Logical AND Registers Rd ← Rd • Rr Z,N,V 1 ANDI Rd, K Logical AND Register and Constant Rd ← Rd • K Z,N,V 1 OR Rd, Rr Logical OR Registers Rd ← Rd v Rr Z,N,V 1 ORI Rd, K Logical OR Register and Constant Rd ← Rd v K Z,N,V 1 EOR Rd, Rr Exclusive OR Registers Rd ← Rd ⊕ Rr Z,N,V 1 COM Rd One’s Complement Rd ← $FF - Rd Z,C,N,V 1 NEG Rd Two’s Complement Rd ← $00 - Rd Z,C,N,V,H 1 SBR Rd, K Set Bit(s) in Register Rd ← Rd v K Z,N,V 1 CBR Rd, K Clear Bit(s) in Register Rd ← Rd • ($FF - K) Z,N,V 1 INC Rd Increment Rd ← Rd + 1 Z,N,V 1 DEC Rd Decrement Rd ← Rd - 1 Z,N,V 1 TST Rd Test for Zero or Minus Rd ← Rd • Rd Z,N,V 1 CLR Rd Clear Register Rd ← Rd ⊕ Rd Z,N,V 1 SER Rd Set Register Rd ← $FF None 1 BRANCH INSTRUCTIONS RJMP k Relative Jump PC ← PC + k + 1 None 2 IJMP Indirect Jump to (Z) PC ← Z None 2 RCALL k Relative Subroutine Call PC ← PC + k + 1 None 3 ICALL Indirect Call to (Z) PC ← Z None 3 RET Subroutine Return PC ← STACK None 4 RETI Interrupt Return PC ← STACK I 4 CPSE Rd, Rr Compare, Skip if Equal if (Rd = Rr) PC ← PC + 2 or 3 None 1/2/3 CP Rd, Rr Compare Rd - Rr Z,N,V,C,H 1 CPC Rd, Rr Compare with Carry Rd - Rr - C Z,N,V,C,H 1 CPI Rd, K Compare Register with Immediate Rd - K Z,N,V,C,H 1 SBRC Rr, b Skip if Bit in Register Cleared if (Rr(b) = 0) PC ← PC + 2 or 3 None 1/2/3 SBRS Rr, b Skip if Bit in Register is Set if (Rr(b) = 1) PC ← PC + 2 or 3 None 1/2/3 SBIC P, b Skip if Bit in I/O Register Cleared if (P(b) = 0) PC ← PC + 2 or 3 None 1/2/3 SBIS P, b Skip if Bit in I/O Register is Set if (P(b) = 1) PC ← PC + 2 or 3 None 1/2/3 BRBS s, k Branch if Status Flag Set if (SREG(s) = 1) then PC ← PC + k + 1 None 1/2 BRBC s, k Branch if Status Flag Cleared if (SREG(s) = 0) then PC ← PC + k + 1 None 1/2 BREQ k Branch if Equal if (Z = 1) then PC ← PC + k + 1 None 1/2 BRNE k Branch if Not Equal if (Z = 0) then PC ← PC + k + 1 None 1/2 BRCS k Branch if Carry Set if (C = 1) then PC ← PC + k + 1 None 1/2 BRCC k Branch if Carry Cleared if (C = 0) then PC ← PC + k + 1 None 1/2 BRSH k Branch if Same or Higher if (C = 0) then PC ← PC + k + 1 None 1/2 BRLO k Branch if Lower if (C = 1) then PC ← PC + k + 1 None 1/2 BRMI k Branch if Minus if (N = 1) then PC ← PC + k + 1 None 1/2 BRPL k Branch if Plus if (N = 0) then PC ← PC + k + 1 None 1/2 BRGE k Branch if Greater or Equal, Signed if (N ⊕ V = 0) then PC ← PC + k + 1 None 1/2 BRLT k Branch if Less than Zero, Signed if (N ⊕ V = 1) then PC ← PC + k + 1 None 1/2 BRHS k Branch if Half-carry Flag Set if (H = 1) then PC ← PC + k + 1 None 1/2 BRHC k Branch if Half-carry Flag Cleared if (H = 0) then PC ← PC + k + 1 None 1/2 BRTS k Branch if T-flag Set if (T = 1) then PC ← PC + k + 1 None 1/2 BRTC k Branch if T-flag Cleared if (T = 0) then PC ← PC + k + 1 None 1/2 BRVS k Branch if Overflow Flag is Set if (V = 1) then PC ← PC + k + 1 None 1/2 BRVC k Branch if Overflow Flag is Cleared if (V = 0) then PC ← PC + k + 1 None 1/2 BRIE k Branch if Interrupt Enabled if (I = 1) then PC ← PC + k + 1 None 1/2 BRID k Branch if Interrupt Disabled if (I = 0) then PC ← PC + k + 1 None 1/2 DATA TRANSFER INSTRUCTIONS MOV Rd, Rr Move between Registers Rd ← Rr None 1 LDI Rd, K Load Immediate Rd ← K None 1 LD Rd, X Load Indirect Rd ← (X) None 2 LD Rd, X+ Load Indirect and Post-inc. Rd ← (X), X ← X + 1 None 2 LD Rd, -X Load Indirect and Pre-dec. X ← X - 1, Rd ← (X) None 2 9 1477KS–AVR–08/10 ATtiny26(L) LD Rd, Y Load Indirect Rd ← (Y) None 2 LD Rd, Y+ Load Indirect and Post-inc. Rd ← (Y), Y ← Y + 1 None 2 LD Rd, -Y Load Indirect and Pre-dec. Y ← Y - 1, Rd ← (Y) None 2 LDD Rd,Y+q Load Indirect with Displacement Rd ← (Y + q) None 2 LD Rd, Z Load Indirect Rd ← (Z) None 2 LD Rd, Z+ Load Indirect and Post-inc. Rd ← (Z), Z ← Z + 1 None 2 LD Rd, -Z Load Indirect and Pre-dec. Z ← Z - 1, Rd ← (Z) None 2 LDD Rd, Z+q Load Indirect with Displacement Rd ← (Z + q) None 2 LDS Rd, k Load Direct from SRAM Rd ← (k) None 2 ST X, Rr Store Indirect (X) ← Rr None 2 ST X+, Rr Store Indirect and Post-inc. (X) ← Rr, X ← X + 1 None 2 ST -X, Rr Store Indirect and Pre-dec. X ← X - 1, (X) ← Rr None 2 ST Y, Rr Store Indirect (Y) ← Rr None 2 ST Y+, Rr Store Indirect and Post-inc. (Y) ← Rr, Y ← Y + 1 None 2 ST -Y, Rr Store Indirect and Pre-dec. Y ← Y - 1, (Y) ← Rr None 2 STD Y+q, Rr Store Indirect with Displacement (Y + q) ← Rr None 2 ST Z, Rr Store Indirect (Z) ← Rr None 2 ST Z+, Rr Store Indirect and Post-inc. (Z) ← Rr, Z ← Z + 1 None 2 ST -Z, Rr Store Indirect and Pre-dec. Z ← Z - 1, (Z) ← Rr None 2 STD Z+q, Rr Store Indirect with Displacement (Z + q) ← Rr None 2 STS k, Rr Store Direct to SRAM (k) ← Rr None 2 LPM Load Program Memory R0 ← (Z) None 3 LPM Rd, Z Load Program Memory Rd ← (Z) None 3 IN Rd, P In Port Rd ← P None 1 OUT P, Rr Out Port P ← Rr None 1 PUSH Rr Push Register on Stack STACK ← Rr None 2 POP Rd Pop Register from Stack Rd ← STACK None 2 BIT AND BIT-TEST INSTRUCTIONS SBI P, b Set Bit in I/O Register I/O(P,b) ← 1 None 2 CBI P, b Clear Bit in I/O Register I/O(P,b) ← 0 None 2 LSL Rd Logical Shift Left Rd(n+1) ← Rd(n), Rd(0) ← 0 Z,C,N,V 1 LSR Rd Logical Shift Right Rd(n) ← Rd(n+1), Rd(7) ← 0 Z,C,N,V 1 ROL Rd Rotate Left through Carry Rd(0) ← C, Rd(n+1) ← Rd(n), C ← Rd(7) Z,C,N,V 1 ROR Rd Rotate Right through Carry Rd(7) ← C, Rd(n) ← Rd(n+1), C ← Rd(0) Z,C,N,V 1 ASR Rd Arithmetic Shift Right Rd(n) ← Rd(n+1), n = 0..6 Z,C,N,V 1 SWAP Rd Swap Nibbles Rd(3..0) ← Rd(7..4), Rd(7..4) ← Rd(3..0) None 1 BSET s Flag Set SREG(s) ← 1 SREG(s) 1 BCLR s Flag Clear SREG(s) ← 0 SREG(s) 1 BST Rr, b Bit Store from Register to T T ← Rr(b) T 1 BLD Rd, b Bit Load from T to Register Rd(b) ← T None 1 SEC Set Carry C ← 1 C 1 CLC Clear Carry C ← 0 C 1 SEN Set Negative Flag N ← 1 N 1 CLN Clear Negative Flag N ← 0 N 1 SEZ Set Zero Flag Z ← 1 Z 1 CLZ Clear Zero Flag Z ← 0 Z 1 SEI Global Interrupt Enable I ← 1 I 1 CLI Global Interrupt Disable I ← 0 I 1 SES Set Signed Test Flag S ← 1 S 1 CLS Clear Signed Test Flag S ← 0 S 1 SEV Set Two’s Complement Overflow V ← 1 V 1 CLV Clear Two’s Complement Overflow V ← 0 V 1 SET Set T in SREG T ← 1 T 1 CLT Clear T in SREG T ← 0 T 1 SEH Set Half-carry Flag in SREG H ← 1 H 1 CLH Clear Half-carry Flag in SREG H ← 0 H 1 NOP No Operation None 1 SLEEP Sleep (see specific descr. for Sleep function) None 1 WDR Watchdog Reset (see specific descr. for WDR/timer) None 1 Instruction Set Summary (Continued) Mnemonic Operands Description Operation Flags # Clocks 10 1477KS–AVR–08/10 ATtiny26(L) Notes: 1. This device can also be supplied in wafer form. Please contact your local Atmel sales office for detailed ordering information and minimum quantities. 2. Pb-free packaging alternative, complies to the European Directive for Restriction of Hazardous Substances (RoHS directive). Also Halide free and fully Green. 3. Code Indicators: – U: matte tin – R: tape & reel Ordering Information Speed (MHz) Power Supply (V) Ordering Code(2) Package(2) Operational Range 8 2.7 - 5.5 ATtiny26L-8PU ATtiny26L-8SU ATtiny26L-8SUR ATtiny26L-8MU ATtiny26L-8MUR 20P3 20S 20S 32M1-A 32M1-A Industrial (-40°C to +85°C)(1) 16 4.5 - 5.5 ATtiny26-16PU ATtiny26-16SU ATtiny26-16SUR ATtiny26-16MU ATtiny26-16MUR 20P3 20S 20S 32M1-A 32M1-A Industrial (-40°C to +85°C)(1) Package Type 20P3 20-lead, 0.300" Wide, Plastic Dual Inline Package (PDIP) 20S 20-lead, 0.300" Wide, Plastic Gull Wing Small Outline (SOIC) 32M1-A 32-pad, 5 x 5 x 1.0 body, Lead Pitch 0.50 mm Quad Flat No-Lead/Micro Lead Frame Package (QFN/MLF) 11 1477KS–AVR–08/10 ATtiny26(L) Packaging Information 20P3 2325 Orchard Parkway San Jose, CA 95131 TITLE DRAWING NO. R REV. 20P3, 20-lead (0.300"/7.62 mm Wide) Plastic Dual Inline Package (PDIP) 20P3 C 1/12/04 PIN 1 E1 A1 B E B1 C L SEATING PLANE A D e eB eC COMMON DIMENSIONS (Unit of Measure = mm) SYMBOL MIN NOM MAX NOTE A – – 5.334 A1 0.381 – – D 25.493 – 25.984 Note 2 E 7.620 – 8.255 E1 6.096 – 7.112 Note 2 B 0.356 – 0.559 B1 1.270 – 1.551 L 2.921 – 3.810 C 0.203 – 0.356 eB – – 10.922 eC 0.000 – 1.524 e 2.540 TYP Notes: 1. This package conforms to JEDEC reference MS-001, Variation AD. 2. Dimensions D and E1 do not include mold Flash or Protrusion. Mold Flash or Protrusion shall not exceed 0.25 mm (0.010"). 12 1477KS–AVR–08/10 ATtiny26(L) 20S 13 1477KS–AVR–08/10 ATtiny26(L) 32M1-A 2325 Orchard Parkway San Jose, CA 95131 TITLE DRAWING NO. R REV. 32M1-A, 32-pad, 5 x 5 x 1.0 mm Body, Lead Pitch 0.50 mm, 32M1-A E 5/25/06 3.10 mm Exposed Pad, Micro Lead Frame Package (MLF) COMMON DIMENSIONS (Unit of Measure = mm) SYMBOL MIN NOM MAX NOTE D1 D E1 E b e A3 A2 A1 A D2 E2 0.08 C L 1 2 3 P P 0 1 2 3 A 0.80 0.90 1.00 A1 – 0.02 0.05 A2 – 0.65 1.00 A3 0.20 REF b 0.18 0.23 0.30 D D1 D2 2.95 3.10 3.25 4.90 5.00 5.10 4.70 4.75 4.80 4.70 4.75 4.80 4.90 5.00 5.10 E E1 E2 2.95 3.10 3.25 e 0.50 BSC L 0.30 0.40 0.50 P – – 0.60 – – 12o Note: JEDEC Standard MO-220, Fig. 2 (Anvil Singulation), VHHD-2. TOP VIEW SIDE VIEW BOTTOM VIEW 0 Pin 1 ID Pin #1 Notch (0.20 R) K 0.20 – – K K 14 1477KS–AVR–08/10 ATtiny26(L) Errata The revision letter refers to the revision of the device. ATtiny26 Rev. B/C/D • First Analog Comparator conversion may be delayed 1. First Analog Comparator conversion may be delayed If the device is powered by a slow rising VCC, the first Analog Comparator conversion will take longer than expected on some devices. Problem Fix/Workaround When the device has been powered or reset, disable then enable the Analog Comparator before the first conversion. 15 1477KS–AVR–08/10 ATtiny26(L) Datasheet Revision History Please note that the referring page numbers in this section refer to the complete document. Rev. 1477K-08/10 Added tape and reel part numbers in “Ordering Information” on page 171. Removed text “Not recommended for new design” from cover page. Updated last page. Rev. 1477J-06/07 1. “Not recommended for new design” Rev. 1477I-05/06 1. Updated “Errata” on page 175 Rev. 1477H-04/06 1. Updated typos. 2. Added “Resources” on page 6. 3. Updated features in “System Control and Reset” on page 32. 4. Updated “Prescaling and Conversion Timing” on page 96. 5. Updated algorithm for “Enter Programming Mode” on page 112. Rev. 1477G-03/05 1. MLF-package alternative changed to “Quad Flat No-Lead/Micro Lead Frame Package QFN/MLF”. 2. Updated “Electrical Characteristics” on page 126 3. Updated “Ordering Information” on page 171 Rev. 1477F-12/04 1. Updated Table 16 on page 33, Table 9 on page 28, and Table 29 on page 57. 2. Added Table 20 on page 40. 3. Added “Changing Channel or Reference Selection” on page 98. 4. Updated “Offset Compensation Schemes” on page 105. 5. Updated “Electrical Characteristics” on page 126. 6. Updated package information for “20P3” on page 172. 7. Rearranged some sections in the datasheet. Rev. 1477E-10/03 1. Removed Preliminary references. 2. Updated “Features” on page 1. 3. Removed SSOP package reference from “Pin Configuration” on page 2. 4. Updated VRST and tRST in Table 16 on page 33. 5. Updated “Calibrated Internal RC Oscillator” on page 29. 16 1477KS–AVR–08/10 ATtiny26(L) 6. Updated DC Characteristics for VOL, IIL, IIH, ICC Power Down and VACIO in “Electrical Characteristics” on page 126. 7. Updated VINT, INL and Gain Error in “ADC Characteristics” on page 129 and page 130. Fixed typo in “Absolute Accuracy” on page 130. 8. Added Figure 106 in “Pin Driver Strength” on page 146, Figure 120, Figure 121 and Figure 122 in “BOD Thresholds and Analog Comparator Offset” on page 155. Updated Figure 117 and Figure 118. 9. Removed LPM Rd, Z+ from “Instruction Set Summary” on page 169. This instruction is not supported in ATtiny26. Rev. 1477D-05/03 1. Updated “Packaging Information” on page 172. 2. Removed ADHSM from “ADC Characteristics” on page 129. 3. Added section “EEPROM Write During Power-down Sleep Mode” on page 20. 4. Added section “Default Clock Source” on page 26. 5. Corrected PLL Lock value in the “Bit 0 – PLOCK: PLL Lock Detector” on page 73. 6. Added information about conversion time when selecting differential channels on page 97. 7. Corrected {DDxn, PORTxn} value on page 42. 8. Added section “Unconnected Pins” on page 46. 9. Added note for RSTDISBL Fuse in Table 50 on page 108. 10. Corrected DATA value in Figure 61 on page 116. 11. Added WD_FUSE period in Table 60 on page 123. 12. Updated “ADC Characteristics” on page 129 and added Table 66, “ADC Characteristics, Differential Channels, TA = -40°C to +85°C,” on page 130. 13. Updated “ATtiny26 Typical Characteristics” on page 131. 14. Added LPM Rd, Z and LPM Rd, Z+ in “Instruction Set Summary” on page 169. Rev. 1477C-09/02 1. Changed the Endurance on the Flash to 10,000 Write/Erase Cycles. Rev. 1477B-04/02 1. Removed all references to Power Save sleep mode in the section “System Clock and Clock Options” on page 23. 2. Updated the section “Analog to Digital Converter” on page 94 with more details on how to read the conversion result for both differential and single-ended conversion. 3. Updated “Ordering Information” on page 171 and added QFN/MLF package information. Rev. 1477A-03/02 1. Initial version. 17 1477KS–AVR–08/10 ATtiny26(L) 1477KS–AVR–08/10 Headquarters International Atmel Corporation 2325 Orchard Parkway San Jose, CA 95131 USA Tel: 1(408) 441-0311 Fax: 1(408) 487-2600 Atmel Asia Unit 1-5 & 16, 19/F BEA Tower, Millennium City 5 418 Kwun Tong Road Kwun Tong, Kowloon Hong Kong Tel: (852) 2245-6100 Fax: (852) 2722-1369 Atmel Europe Le Krebs 8, Rue Jean-Pierre Timbaud BP 309 78054 Saint-Quentin-en- Yvelines Cedex France Tel: (33) 1-30-60-70-00 Fax: (33) 1-30-60-71-11 Atmel Japan 9F, Tonetsu Shinkawa Bldg. 1-24-8 Shinkawa Chuo-ku, Tokyo 104-0033 Japan Tel: (81) 3-3523-3551 Fax: (81) 3-3523-7581 Product Contact Web Site www.atmel.com Technical Support Enter Product Line E-mail Sales Contact www.atmel.com/contacts Literature Requests www.atmel.com/literature Disclaimer: The information in this document is provided in connection with Atmel products. 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Un design innovant pour une utilisation efficace et confortable Un boîtier léger et compact et une interface utilisateur conviviale Un afficheur d’une dimension et d’une lisibilité inégalées Une calibration 100% numérique pour une précision et un coût maîtrisés Des versions entièrement programmables respectant la norme SCPI Nouvelle technologie d’affichage intelligent (Smart Persistence Oscilloscope) Des atouts qui séduiront l’Enseignement et l’Industrie Si intelligents que vous pouvez les choisir pour leurs courbes ! Série MTX COMPACT « Une gamme complète et performante d’instruments de laboratoire » MTX 3252 MTX 3352 MTX 3354 MTX 3240 MTX 3250 MTX 3250 Multimètre-Analyseur MTX 3240 Générateur BF-Mesureur MTX 3252 - MTX 3352 - MTX 3354 Oscilloscopes-Analyseurs Tout comme le générateur de la même famille, le multimètre MTX est un appareil surdoué, multifonction. Grâce à son analyse du signal il évite à l’utilisateur la mise en oeuvre d’autres instruments (oscilloscope par exemple), pour contrôler la validité des mesures réalisées. Impossible de commettre les, si fréquentes et souvent ignorées, erreurs dues à un facteur de crête trop élevé. En effet, le MTX 3250 mesure en permanence les crêtes rapides à 500 μs et vous alerte en cas d’anomalie. Mieux encore, en validant alors le “Mode AUTO PEAK”, le multimètre commutera automatiquement sur une gamme adaptée à la nature du signal mesuré. L’affichage du facteur de crête vous permettra aussi d’établir un premier diagnostic qualitatif sur vos signaux. Le MTX 3250, c’est encore la rationalisation de l’investissement réalisé, puisqu’il est également fréquencemètre, thermomètre et même enregistreur, vous évitant ainsi l’achat d’un instrument d’usage ponctuel. Ainsi, pour les enregistrements en laboratoire jusqu’à 4 voies et 12 paramètres, la version “acquisition de données” et son logiciel PC associé rendront un service performant à partir d’un instrument polyvalent. La température est directement mesurée à partir de Pt 100 ou Pt 1000, de même que la fréquence, jusqu’à 1 MHz, avec période et rapport cyclique. Et pour mieux répondre à vos attentes dans le cadre de systèmes automatisés, cet instrument existe en version programmable à 100 % via une liaison optique RS232 à 57600 bauds, compatible SCPI. Un multimètre à la pointe des modes 50 000 points 500 mV - 500 mV - 600 V 500 μA - 500 mA & 10 A 5 μA – 500 mA & 10A 500 Ω - 50 MΩ 170 x 270 x 190 mm optique RS232 LCD 50 x 140 mm 500 V & 0,5%L + 3D 0,2%L + 3D 0,5%L + 3D 0,1%L + 3D Masse : 2,3 kg 57 600 bauds Rétro-éclairage 1000 V (50 000 pts) 10 kHz Aff. triple 0,08%L+3D 100 kHz CARACTÉRISTIQUES : Affichage Gammes et Gammes et Gammes et Gammes et Gammes et Dimensions Interface précision précision VAC de base précision IDC de base précision IAC précision de base H x L x P MTX 3250-P de base VDC Bande passante Bande passante Ohm MTX 3250-A Autres mesures : test continuité, test diode capacités 50 nF – 50 mF, fréquence 1 Hz - 1 MHz, rapport cyclique 0,01% à 100%, température - 200 à + 800°C, pt 100 et pt 1000. Fonction PEAK HOLD : Pk+/ -500 μs sur I & V, facteur de crête Fonctions complémentaires : SURV = MIN/MAX datés / MATH = dB, dBm, ax+b / OFFSET (Offset, nul, delta%) / Data HOLD & Auto HOLD Fonction supplémentaire sur MTX 3250-P : PRINT, cadence 0,5 s à 10 h, horloge et calendrier, pilotage RS232 optique Fonction supplémentaire sur MTX 3250-A : DATA LOGGER avec 1500 mesures stockées, 1 ou 3 valeurs simultanées. Normes : sécurité selon IEC 61010-1, 2001 et CEM selon NF 61326-1, 1998 Garantie : 3 ans Le multimètre MTX 3250 est fourni avec 1 câble d’alimentation secteur, 1 jeu de cordons de mesure, une notice de fonctionnement et une présentation interactive de l’instrument sur CD-Rom. Pour commander MTX3250 Multimètre de Table 50 000 pts MTX3250-P Multimètre de Table 50 000 pts + RS232 Fourni avec un cordon de liaison optique RS232, un manuel de programmation et les drivers Labwindows / Labview sur CD-Rom. MTX3250-A Multimètre de Table 50 000 pts + Acquisition Fourni avec un cordon de liaison optique RS232, un manuel de programmation, les drivers Labwindows / Labview et le logiciel d’acquisition de données SX-DMM sur CD-Rom. Tout commence avec une connexion réduite à 3 bornes qui limite erreurs et manipulations, permettant un “AUTORANGING” courant complet de 50 μA à 20 A. Le MTX 3250 offre ensuite, grâce à son affichage triple, les combinaisons de mesures qui répondent simplement et efficacement à vos applications courantes, comme, par exemple, la mesure de bande passante (affichage de l’atténuation en dB et de la fréquence). Accessoires et informations pour commander Pour la maîtrise métrologique, le “Mode SPEC” calcule et affiche les incertitudes de l’instrument en fonction des gammes et de la valeur mesurée Le Mode offre la lecture directe de la grandeur mesurée, ainsi que l’unité physique correspondante Le “Mode Surveillance” enregistre les minima et les maxima afin de piéger et de dater vos défauts L’association fonctionnelle Le “Mode RELATIF” exprimé en absolu, pourcentage ou dB (ratio), permet une exploitation directe MTX 3250 : multimètre-analyseur intégré Asservissement et affichage de la fréquence Contrôle et affichage de l’AMPLITUDE VCC (crête/crête) et de l’OFFSET VDC Contrôle et affichage du rapport cyclique Le MTX 3240 c’est aussi la rationalisation de l’investissement réalisé, puisqu’il est aussi un fréquencemètre 100 MHz (Cat. I, 300 V), vous évitant ainsi l’achat d’un instrument d’usage souvent ponctuel. Et pour répondre enfin de manière économique à vos attentes dans le cadre de systèmes automatisés, ce générateur existe en version programmable à 100 % via une liaison rapide, compatible SCPI. Sa technologie permet à chacun de bénéficier de fonctions nouvelles, indispensables : Réglage de la fréquence, garantie stable au digit près, et accélérateur intelligent avec changement de gammes automatique pour la fréquence Changement de gammes automatique optimisé pour l’amplitude “LEVEL et OFFSET” Rapport cyclique réglable sans variation ni division de la fréquence Fonction “LOGIC” pour une réponse simple et rapide à la génération de signaux logiques à seuils directement ajustables Un générateur robuste, avec des sorties protégées 60 VDC / 40 VAC L’association fonctionnelle MTX 3240 : générateur-mesureur autonome Un générateur doté de caractéristiques innovantes LCD 50 x 140 mm 0,1 Hz à 5,1 MHz Sinus, carré, 1) Principale : LIN ou LOG 0,1 Hz à 100 MHz 115 V - 230 V - 170x270x190 mm optique Afficheur principal 7 gammes + triangle, jusqu’à 20 VCC CONTINU, Précision : 0,05 % 240 V Masse : 2,8 kg RS232 20 mm réglage fin au digit impulsion, circuit ouvert, 1 : 50 Min Entrée 300 V, Cat. I 50 / 60 Hz 4 grandeurs près + accélérateur rampe, TTL, gamme automatique De 10 ms à 10 s Sensibilité 300 V, Cat.II simultanées Précision : 0,05 % LOGIC 2) TTL Interne ou externe automatique Distorsion Protection : overload < 0,5 % 60 VDC / 40 VAC CARACTÉRISTIQUES : Affichage Gamme Formes Sorties Balayage Fréquencemètre Alimentation Dimensions Interface de fréquence de signaux externe H x L x P (MTX 3240-P) Normes : sécurité selon IEC 61010-1, 2001 et CEM selon NF 61326-1, 1998 Garantie : 3 ans Le générateur MTX 3240 est fourni avec un câble d’alimentation secteur, une notice de fonctionnement et une présentation interactive de l’instrument sur CD-Rom. Pour commander MTX3240 Générateur de Fonctions 5,1 MHz MTX3240-P Générateur de Fonctions 5,1 MHz + RS232 Fourni avec un cordon de liaison optique RS232, un manuel de programmation et les drivers Labwindows / Labview sur CD-Rom. Accessoires et informations pour commander Autre apport de l’innovation pour l’utilisateur : une fonctionnalité complète pour l’investissement réalisé. En effet, l’association fonctionnelle du MX 3240 permet sa mise en oeuvre autonome, ce qui évite, par exemple, l’utilisation systématique d’un oscilloscope ou d‘un multimètre, simplement pour en contrôler les réglages. MTX 3240 avec fréquencemètre intégré Interface Homme-Machine, la simplicité au service de la performance L’instrument peut être piloté par la souris ou le clavier Les réglages sont simplifiés et conviviaux grâce aux 21 touches d’accès direct de la face avant et à l’environnement « Windows-Like » Une aide en ligne détaillée et en 5 langues est disponible à tout moment grâce à la touche Léger, compact et pourvu d'une poignée, il dispose d’un « Pack Terrain » qui permet la mise en oeuvre de l ‘oscilloscope sans le sortir de sa sacoche Affichage Des curseurs peuvent être placés à tout moment sur les signaux pour réaliser des mesures précises Grâce à une profondeur mémoire de 50.000 points, le zoom de trace horizontal, « Winzoom », peut aller jusqu’à un facteur x100 en affichant des vrais points acquis . Une dynamique verticale exceptionnelle de 2,5 mV à 100 V par div. La fonction « full trace » permet de diviser l’écran en deux afin d’optimiser la lisibilité des courbes Affichage jusqu’à 4 courbes à l’écran et possibilité d’établir des comparaisons entre deux courbes. Pour plus de simplicité et un gain de temps conséquent, l’utilisateur peut sélectionner et afficher 2 mesures automatiques parmi 19. MTX 3252, MTX 3352 et MTX 3354 : Oscilloscopes – analyseurs 2 ou 4 voies, de 60 à 150 MHz ! Experts en communication Équipés d’une liaison RS232, de l’interface Centronics et d’une liaison USB indispensables à la communication vers un PC ou une imprimante Gestion à distance grâce à la liaison Ethernet et au serveur HTML présent dans chaque instrument Des fonctions complexes d’analyse sont accessibles en mode « avancé » , elles sont masquées en mode « standard » dans un soucis de simplification. Enregistrement, une mémoire infinie …, Les oscilloscopes de la gamme MTX COMPACT disposent d’un écran LCD couleur 5"7 avec rétro éclairage pour une excellente précision de lecture Le choix est large puisque MTX COMPACT propose trois appareils de 2 et 4 voies de bande passante 60 MHz, 100 MHz ou 150 MHz qui séduiront aussi bien l'Enseignement Technique que l’Industrie Le déclenchement dispose de 5 modes différents : Pulse, déclenchement sur largeur d'impulsion, Retard, déclenchement sur fronts avec retardateur, TV, déclenchement sur un signal TV, Comptage, déclenchement sur font avec comptage d'événements et Secteur, déclenchement sur le front ascendant ou descendant de la tension réseau 50/60 Hz Outre ces multiples modes de paramétrage, le Holdoff est disponible sur la majorité de ces fonctions de déclenchement , technologie d’affichage de type « analogique » qui permet de faire apparaître les évolutions du signal (modulations, jitters etc.) et les phénomènes uniques (transitoires, glitchs, etc.) La profondeur mémoire de 50.000 points est une référence dans cette catégorie d’oscilloscopes Durée d’enregistrement et fréquence d’échantillonnage 20 fois plus élevées qu’un oscilloscope traditionnel Les oscilloscopes de la gamme MTX COMPACT dispose d’une résolution exceptionnelle de 100 Gé/s en mode répétitif et 200 Mé/s en mode monocoup, ce qui permet d’avoir des calibres de base de temps allant de 200 s/div à 1ns/div. Enregistrement de courbes et rappel à l’écran Possibilité de sauvegarder des fichiers dans l’instrument, de les imprimer ou de les exporter vers un PC en vue d’une exploitation ultérieure dans les applications « Windows » (rapports, tableurs, impression, images …) Les traces et les fichiers enregistrés sont horodatés Les fichiers sont générés dans des formats standards : .gif, .pcl, .txt, .bmp, .eps, .prn, etc. Des instruments intégrés pour un « outil global » L’ensemble des oscilloscopes sont pourvus de la fonction analyse FFT temps réel et multivoie du signal Pour les utilisateurs du domaine de l’Electrotechnique, nous proposons en option l’analyse d’harmoniques multivoies 31 rangs Enfin, pour tous ceux qui doivent surveiller dans le temps les variations de phénomènes physiques ou mécaniques , un enregistreur numérique rapide est intégrable dans l’instrument, sous forme d’un module software MTX 3252, MTX 3352 et MTX 3354 : Oscilloscopes – analyseurs 2 ou 4 voies, de 60 à 150 MHz ! Ergonomie La technologie " " permet de faire persister les acquisitions pendant une durée paramétrée pour observer un cumul de traces. L’intensité lumineuse ou la couleur, affectée au point à l’écran, va décroître si celui-ci n’est pas renouvelé lors d’une nouvelle acquisition. L’acquisition se fait donc en trois dimensions : - le temps - l’amplitude - l’occurrence Grâce à sa profondeur mémoire de 50.000 points, l’oscilloscope acquiert et traite l’information en parallèle. Le nombre d’acquisitions à la seconde peut-être multiplié par un facteur supérieur à 1000, ainsi le temps mort entre deux acquisitions est considérablement réduit. Représentation à l’écran des 50.000 points acquis par un système de compression intelligente. L’occurrence apporte une dimension statistique à la répartition des échantillons. La couleur ou l’intensité lumineuse met en évidence les irrégularités du signal. Durées d’affichage des points acquis : 100ms, 200ms, 500ms, 1s, 2s, 5s, 10s et infini. , Smart Persistence Oscilloscope : L’outil indispensable de visualisation intelligente ! La technologie SPO La nouvelle génération d’oscilloscopes MTX COMPACT est dotée de l'affichage " " (Smart Persistence Oscilloscope) qui permet, comme en analogique, de faire apparaître les évolutions du signal dans le temps, les jitters, les modulations et les phénomènes instables. Par ailleurs, ce mode d’affichage permet aussi la mise en évidence des phénomènes uniques tels que les transitoires ou les glitchs. Acquisition Traitement rapide Affichage Parallèle N acquisitions = 1 affichage Modulation AM ou FM Évènements uniques ou transitoires Caractérisation du bruit Signal vidéo Applications - MTX3354E-C : oscilloscope numérique 4x150MHz, couleur, Ethernet - MTX3354E-CK : MTX3354E-C + SX-METRO/P - MTX3252BE-C : oscilloscope numérique 2x60MHz, couleur, Ethernet - MTX3352BE-C : oscilloscope numérique 2x100MHz, couleur, Ethernet - MTX3252BED : MTX3252BE-C + sonde différentielle MTX1032-B - MTX3352BED : MTX3352BE-C + sonde différentielle MTX1032-C Pour commander MTX 3252, MTX 3352 & MTX 3354 : Oscilloscopes – analyseurs 60, 100 et 150 MHz Interface Homme-Machine Affichage LCD couleur 5” 7 (115x86mm) - LCD monochrome ou couleur 5” 7 (115x86mm) 320 x 240 + rétro éclairage CCFL - 320 x 240 + rétro éclairage CCFL Nombre de courbes à l’écran 4 courbes + 4 références Commandes 21 touches de raccourcis directs + 1 encodeur + 1 touche « aide » Menu « Windows-like » - 100% des commandes accessibles via la souris Choix des langues par le menu (FRA/ANG/ESP/ITA/ALL) Déviation Verticale Bande passante 150 MHz 100 MHz 60 MHz (limiteur de BP 15 MHz (limiteur de BP (limiteur de BP 1,5 MHz et 5 kHz) 15 MHz) 15 MHz) Nombre voies 4 voies classe 1 – Cat. II 300 V 2 voies classe 1 – Cat. II 300 V Sensibilité 2,5 mV – 100 V/div + expansion verticale 2,5 mV – 100 V/div "Winzoom" jusqu'à un facteur 10 + expansion verticale « Winzoom » (sensibilité maximum 250 μV/div) jusqu’à un facteur 10 Temps de montée < 3 ns < 3,5 ns Déviation Horizontale Vitesse de balayage 1 ns à 200 s/div - winzoom graphique jusqu'à un facteur 100 Déclenchement Modes Auto, Normal, Monocoup, Auto 50% Types Front, Largeur d’impulsion, Retard, Comptage d’évènements, Compteur de lignes TV, Hold-off Sources CH1, CH2, CH3, CH4, Secteur CH1, CH2, EXT et Secteur Mémoire Numérique Échantillonnage maxi. Répétitif = 100 Gé/s Monocoup = 200 Mé/s (2 voies), convertisseur 9 bits Profondeur mémoire 50.000 points – 4 références + 16 courbes de 50 kpts Modes d’affichage Glitch, Enveloppe, Moyennage, XY Numérique SPO (Smart Persistence Oscilloscope) Durée 100 ms, 200 ms, 500 ms, 1 s, 2 s, 5 s 100 s et Infini Représentation Monochrome ou couleur Vitesse d’acquisition 50 kwaveform/s max. par voie – 19 Mé traités par s et par voie Mode Enregistreur Cadence d’acquisition De 40 μs à 54 μs d’intervalle d’échantillonnage (2 s à 31 jours d’enregistrement) Exploitation Horodatage direct, conversion et unités des grandeurs physiques, mesures par curseurs et recherche d’événement, fichiers exploitables sur tableur standard Mode Analyseur d’Harmoniques Étendue d’analyse 31 rangs simultanément sur 1 à 4 voies Exploitation Affichage permanent : valeur RMS totale THD – Rang sélectionné : %F, phase, freq, Vrms Interface RS232, Centronics, USB, Ethernet avec serveur HTML Caractéristiques Générales Boîtier 210 x 177 x 200 – 2,5 Kg – IP30 Alimentation 100 à 240 VAC – 47 à 63 Hz Sécurité IEC 1010-1 (2001) – Surtension de l’alimentation CAT II 240 V – Surtension des entrées de mesure CAT II 300V CEM NF EN 61326-1 07/97 + A1 10/98 CARACTÉRISTIQUES : MTX 3354 MTX 3352 MTX 3252 AD.COM- Code : 906210055 - Ed.5 - 09/2007 - Caractéristiques sous réserve de modifications liées à l’évolution de la technologie L’efficacité s’affiche avec élégance Le design moderne et séduisant des appareils de la famille MTX permet, grâce à une forte compacité, leur parfaite intégration dans votre cadre de travail. Posés directement sur la paillasse, l’espace libéré devant eux est déjà très appréciable, de plus, leur hauteur est calculée pour pouvoir les glisser sans peine sous les demi-étagères. Leur faible profondeur et leur largeur standard vous permettent aussi de les placer sur ces mêmes demi-étagères ou de les poser sur un autre instrument. Le déplacement et le transport sont très aisés grâce à leur poignée intégrée et à leur légèreté. Une architecture et des formes avantageuses qui vous font gagner de la place Même à distance ou dans des conditions d’éclairage difficiles (soleil, néons), les mesures sont parfaitement lisibles, grâce notamment à un afficheur négatif de grandes dimensions (50 x 140 mm), à un rétro-éclairage à matrice de leds ajustable, ainsi qu’à une hauteur exceptionnelle de 20 mm de l’afficheur principal (MTX 3240 et MTX 3250). Monochrome ou couleur, l’écran LCD réglable et orientable des MTX 3252 et MTX 3352 vous garantira, lui aussi, une lisibilité en toutes circonstances. Les zones fonctionnelles de l’ensemble des MTX Compact sont vastes, cohérentes et hiérarchisées, et les connexions mesure facilement accessibles, puisque situées en façade. Sur le générateur et le multimètre, la sélection des fonctions primaires s’établit directement au moyen de touches avec leds de validation intégrées. Un encodeur performant permet de réaliser efficacement les réglages et des touches contextuelles en bord d’écran indiquent clairement la configuration. Pour les oscilloscopes, en plus des accès directs aux fonctions essentielles par clavier et aux réglages par encodeur, le pilotage via la souris, sous environnement “Windows-like”, est totalement innovant dans cette catégorie d’appareil. Une technologie de leader, l’innovation jusqu’au bout des doigts Une lisibilité privilégiée, une Interface Homme-Machine conviviale (I.H.M.) Les qualités des MTX ne se limitent pas à leur physique. Ces instruments ont une tête bien faite, grâce notamment à leurs micro-processeurs 16 ou 32 bits de dernière génération, aux logiciels téléchargeables et à la calibration 100% numérique. Du point de vue de la sécurité, une protection électronique réarmable a permis la suppression du fusible secteur sur certains modèles. Tous les modèles de la famille MTX Compact peuvent disposer d’interfaces de communication performantes et du langage standard SCPI. Même le clavier de sélection est à la pointe de la technologie grâce à ses contacts à microswitches qui assurent une durabilité exceptionnelle de plus de 100 000 manoeuvres. Les touches bénéficient, quant à elles, d’une gravure laser inaltérable. Avec la famille MTX Compact, Metrix permet à chaque professionnel d’accéder à des Instruments “de haute couture”, dont vous ne pourrez plus vous passer Pour informations et commandes FRANCE Chauvin-Arnoux 190, rue Championnet 75876 PARIS Cedex 18 Tél : (33) 01 44 85 44 58 Fax : (33) 01 46 27 07 48 info@metrix.fr www.metrix.fr SUISSE Chauvin Arnoux AG Einsiedlerstraße 535 - 8810 HORGEN Tél : 01/727 75 55 Fax : 01/727 75 56 info@chauvin-arnoux.ch www.chauvin-arnoux.ch MOYEN-ORIENT Chauvin Arnoux Middle East Ain El Zalka, Immeuble Zalka 686 ZALKA (Beyrouth) Tél : +961 1 890 425 Fax : +961 1 890 424 camie@chauvin-arnoux.com www.chauvin-arnoux.com Cube 3D Printer 2nd generation ® User Guide See inside for use and safety information. Table of Contents Introduction 2 Important safety information EN 3 DE 4 ES 5 FR 6 IT 7 DU 8 DA 9 JP . 10 Cube 3D Printer features . 11 MTX 3250 Mull tt iimètt rre de Tablle 50 000 ptts mull tt ii ffonctt iion Multimètre-Analyseur MTX : la précision et le contrôle... ! • Très vaste affichage triple 50 000 pts pour une meilleure efficacité • Précision de base 0,08%, bande-passante 100 kHz • 3 bornes de mesure et «AUTORANGING» complet en courant • Fonction « SPEC » permettant de visualiser directement les incertitudes • Avec le mode « AUTOPEAK » plus de limitation du facteur de crête • Mesures fréquentielles jusqu’à 1 MHz, de la période et du rapport cyclique • Mesures de température à partir de sondes Pt 100 / Pt 1000 • Mode «MATH» pour une lecture directe (conversion & unité physique) • Un modèle programmable via RS232 selon protocole SCPI • Un modèle « Acquisition de données » avec enregistreur horodaté MTX Compactt Caractéristiques techniques MTX 3250 Tensions DC, AC et AC+DC Gammes 500 mV 5 V 50 V 500 V 600 V Résolution 10 μV 100 μV 1 mV 10 mV 100 mV Précision DC 0,08% L+3D 0,08% L+3D 0,08% L+3D 0,1% L+3D 0,1% L+3D Bande passante AC / AC+DC 40 Hz à 100 kHz/DC à 100 kHz Précision de base AC et AC+DC 0,5% L+30D (0,5% L+40D/gamme 500 mV) pour DC à 1 kHz Impédance d’entrée 10 MΩ/1 GΩ 11 MΩ 10 MΩ 10 MΩ 10 MΩ Protection / Surcharge admissible Protection réarmable 600 Vrms permanents/1000 VDC ou 700 VAC (1min max.) Courants DC, AC et AC+DC Gammes 500 μA 5 mA 50 mA 500 mA 10A Résolution 10 nA 100 nA 1 μA 10 μA 1mA Précision DC 0,2% L+5D 0,2% L+3D 0,2% L+3D 0,2% L+5D 0,5% L+5D Bande passante AC / AC+DC 40 Hz à 10 kHz/DC à 10 kHz Précision de base AC et AC+DC 0,5% L+30D (2,5% L+30D/gamme 10A) pour DC à 1 kHz Protection / Surcharge admissible Fusible HPC 10 A, 600 V/50 kA/20 ADC ou 20 Arms (30 s max.) Mesures Fréquentielles Fréquence (tension ou courant) Plage de mesure 1 Hz à 1 MHz - 7 gammes de 5,0000 Hz à 1,0000 MHz - Précision 0,03% L+2D jusqu’à 50 kHz Rapport Cyclique Résolution 0,01% - Période du signal de 10 μs à 0,8 s Résistances & Continuité Gammes 500 Ω 5 kΩ 50 kΩ 500 kΩ 5 MΩ 50 MΩ Résolution 10 mΩ 100 mΩ 1 Ω 10 Ω 100 Ω 1 kΩ Précision de base 0,1% L+5 D 0,1% L+3 D 0,1% L+3 D 0,1% L+3 D 0,5% L+3 D 1% L+5 D Protection Protection réarmable 600 Vrms Détection en continuité sonore Gamme 500 Ω - Seuil 10 Ω à 15 Ω - Temps de réponse 1ms Test de diode Mesures de tension de diode De 0 à 4,5 V - Précision 0,2% L+3D - courant de mesure 1 mA env. - Protection réarmable 600 Vrms Capacités Performances 7 Gammes de 50,00 nF à 50,00 mF - Précision de base 1%L+ 3D - Temps de mesure 1 s jusqu’à 50 μF Protection Protection réarmable 600 Vrms Température ( sondes Pt 100 / Pt 1000 ) Performances Plage de mesure –125,0 °C à +800,0 °C - Précision 0,5 °C (de –125 °C à 75 °C) - Protection réarmable 600 Vrms Autres Mesures Capture de pics rapides >500 μs Valide sur toutes les gammes - Erreur additionnelle en Tension 3% L+10D, en Courant 4% L+10D Mesure en dBm Résolution 0,01 dBm - Référence ajustable de 1 Ω à 9999 Ω - Protection réarmable 600 Vrms Puissance Résistive U2/R ou R I2 Résolution 100 μW - Référence ajustable de 1 Ω à 9999 Ω - Protection réarmable 600 Vrms Autres fonctions Fonction AUTOPEAK Surveillance permanente du facteur de crête (FC) et gestion automatique des gammes pour respecter le FC spécifié Fonction SPEC Calcul de la tolérance de mesure sous forme x% L+x D ou Min et Max sur les afficheurs 2 et 3 Fonction HOLD & AUTOHOLD Maintien manuel de l’affichage (HOLD) ou automatique sur mesure stable (AUTOHOLD) Fonction REL / fonction dB Mode triple affichage : valeur relative, valeur absolue, écart en % / écart en dB Fonction SURV Surveillance et mémorisation automatique des «MIN» et «MAX» avec horodatage des événements Fonction MATH Mise à l’échelle et à l’unité pour les grandeurs physiques (fonction y = Ax+B et unité définissables) Fonction STORE Acquisition de données (jusqu’à 3 mesures à la fois) - Cadence 0,5 s à 10 h - 10 mémoires & env. 1500 mesures Fonction PRINT Envoi direct sur imprimante série des mesures horodatées - Cadence 0,5 s à 10 h * choix de l’utilisateur à la mise sous tension Caractéristiques générales MTX 3250 Affichage Affichage triple 50 000 points - Dimensions utiles : 130 x 50 mm - Hauteur des chiffres 20 mm et 10 mm Cadence de Mesure 2 mesures par seconde (mode 50 000 pts) Rétroéclairage Affichage positif rétroéclairé par matrice de LED’s - Contraste réglable Température Référence 23°C ±5° (après 1 heure de chauffe) - Fonctionnement 0°C à +45°C - Stockage -20°C à +70°C Humidité Relative < 80 % à 40°C CEM / Sécurité Emission et immunité selon NF EN 61326-1, 1998 / IEC 61010, 2001 Alimentation Secteur 230 V ±10% ou 110 V ±10% (50 Hz – 60 Hz) - Cat.II/300V Alimentation Batterie (Option) Accumulateur 9,6 V - Autonomie 10 à 12 H environ (suivant les fonctions) Interface RS232 Optique (Option) Trame : 8 bits de données, 1 bit de stop, pas de parité - Vitesses sélectionnables 9600 & 57600 bauds Boîtier ABS V0 - Dimensions H/L/P : 170 x 270 x 195 mm - Masse : 2,8 kg - Indice de protection IP30 Caractéristiques sous réserve de modifications liées à l’évolution de la technologie 190, rue Championnet 75876 PARIS cedex 18 Tél. : 01 44 85 44 58 Fax : 01 46 27 07 48 Filiale suisse : Einsiedlerstrasse 535 8810 HORGEN Tél. : 01 / 727 75 55 - Fax : 01 / 727 75 56 Agences : Lille 03 20 55 96 41 Lyon 04 72 65 77 60 Nancy 03 83 92 19 21 Nantes 02 40 84 01 16 Paris 01 44 85 45 70 Rennes 02 99 22 80 80 Toulouse 05 62 74 50 30 Code : 906210045 - Ed.1-03/02 MTX 3250 : Multimètre 50 000 pts multifonction Pour commander : Etat de livraison : 1multimètre de table MTX 3250, 1 câble d’alimentation secteur, 1 jeu de cordons de mesure et 1 notice de fonctionnement. MTX3250 : Multimètre de table 50 000 pts MTX 3250 MTX3250-P : Multimètre de table 50 000 pts MTX 3250 + RS232 MTX3250-A : Multimètre de table 50 000 pts MTX 3250 + acquisitionAt a glance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 Requirements for your Cube . 13 Unpacking and setting up your Cube 14 Link your Cube to your Cubify account 15 Unlock your Cube 16 Download and install Cube Software for Windows 16 Download and install Cube Software for Mac OSX . 17 Download your free creations 17 Cubify Software overview 18 WI-FI set-up . 20 Non-wireless computer set-up (without WI-FI options) 20 Update Cube firmware . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 Setting Print Jet gap 21 Material Cartridge installation 22 Printing preparation . 23 Printing your first creation 24-25 Replacing Material Cartridge . 26 Cloud printing from Cubify.com . 27 Finishing your creation 27 Maintaining your Cube . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 Print Pad leveling instructions . 29 Thank you for purchasing the “Cube®” 3D Printer. This printer is portable with a plug and print design that enables everybody in the family to express their creativity like never before. With ten different material colors to choose from, enjoy the freedom to print in your true colors or to mix it up.Cube 3D Printers ready-to-print technology provides a new dimension to your imagination and helps you share your creations with others in the Cubify community at Cubify.com. At Cubify.com you can: • Upload your creations for sale • Purchase creations from others • Get your creations 3D printed and shipped to you • Buy the Cube 3D Printer and Cube Cartridges • Engage with other creative partners COPYRIGHT INFORMATION © 2013 by 3D Systems, Inc. All rights reserved. This document is subject to change without notice. This document is copyrighted and contains proprietary information that is the property of 3D Systems, Inc. Cubify, Cube, and the 3D Systems logo are registered trademarks of 3D Systems, Inc. Use of the Cubify.com website constitutes acceptance of its Terms of Service and Privacy Policy. FCC NOTICE This equipment has been tested and found to comply with the limits for a class “B” digital device, pursuant to Part 15 of the FCC Rules. These limits are designed to provide reasonable protection against harmful interference. This equipment generates, uses, and can radiate radio frequency energy and, if not installed and used in accordance with the instruction manual, may cause harmful interference to radio communications. Operation of this equipment in a residential area is likely to cause harmful interference in which case the user will be required to correct the interference at their expense. WARRANTY 3D Systems warrants that the Cube 3D Printer will be free from defects in materials and workmanship, during the applicable warranty period, when used under the normal conditions described in the documentation provided to you, including this user’s guide. 3D Systems will promptly repair or replace the Cube 3D Printer, if required, to make it free of defects during the warranty period. This warranty excludes (i) normal consumable or expendable parts (such as Material Cartridges, Print Pads, and CubeStick), (ii) repairs required during the warranty period because of abnormal use or conditions (such as riots, floods, misuse, neglect or improper service by anyone except 3D Systems or its authorized service provider), and (iii) repairs required during the warranty period because of the use of non-integrated, non-approved or non-licensed materials in the Cube 3D Printer. The warranty period for the Cube 3D Printer is ninety (90) days and shall start on the date your Cube 3D printer is activated. For consumers who are covered by consumer protection laws or regulations in their country of residence, the benefits conferred by our ninety (90) day warranty are in addition to, and operate concurrently with, all rights and remedies conveyed by such consumers protection laws and regulations, including but not limited to these additional rights. THIS WARRANTY IS THE ONLY WARRANTY PROVIDED FOR THE CUBE 3D PRINTER. TO THE MAXIMUM EXTENT PERMITTED BY LAW, 3D SYSTEMS EXPRESSLY DISCLAIMS ALL OTHER WARRANTIES FOR THE CUBE 3D PRINTER AND EACH OF ITS COMPONENTS, WHETHER THOSE WARRANTIES ARE EXPRESS, IMPLIED OR STATUTORY INCLUDING WARRANTIES OF MERCHANTABILITY AND FITNESS FOR INTENDED OR PARTICULAR PURPOSES. LIMITATION OF LIABILITY 3D SYSTEMS WILL NOT BE RESPONSIBLE FOR CONSEQUENTIAL, EXEMPLARY OR INCIDENTAL DAMAGES (SUCH AS LOSS OF PROFIT OR EMPLOYEE’S TIME) REGARDLESS OF THE REASON. IN NO EVENT SHALL THE LIABILITY AND/OR OBLIGATIONS OF 3D SYSTEMS ARISING OUT OF THE PURCHASE, LEASE, LICENSE AND/OR USE OF THE EQUIPMENT BY YOU OR OTHERS EXCEED THE PURCHASE PRICE OF THE CUBE 3D PRINTER. 2 1 INTRODUCTION SAFETY SYMBOLS AND DEFINITIONS SAFETY GUIDELINES • Follow all safety rules in this section and observe all cautions and warnings in this guide. • Do not modify any safety features or make modifications to the Cube. Doing so is prohibited and voids warranty. • Use of print materials, or 3D prints other than 3D Systems print materials and genuine 3D Systems components may void warranty. • Adult supervision is required; observe children closely and intervene as necessary to prevent potential safety problems and ensure the Cube’s appropriate use. Ensure small 3D prints are not accessible to young children. These 3D prints are potential choking hazards for young children. • When the Cube is operating, the print tip on the Print Jet becomes hot; avoid touching this area until it has cooled down. • Do not change color of material during printing; doing so may damage the Cube. ! Hot Surface Hazard: A hot surface is accessible in the vicinity of this sign or at the Print Jet; avoid contact. Hot surfaces can cause severe burns. Caution: Indicates something may happen that could cause loss of data, damage to equipment, or could cause personal injury. Caution: Indicates a pinch point hazard that could cause person injury. 3 2 IMPORTANT SAFETY INFORMATION (EN) SICHERHEITSSYMBOLE UND DEFINITIONEN SICHERHEITSHINWEISE • Befolgen Sie alle Sicherheitsvorschriften in diesem Abschnitt und beachten Sie alle Vorsichtsmaßnahmen und Warnhinweise in diesem Handbuch. • Modifizieren Sie keine Sicherheitsmerkmale und nehmen Sie keine Änderungen am Cube-Drucker vor. Dies ist verboten und kann zum Erlöschen der Gewährleistung führen. • Die Verwendung von Druckmaterialien oder 3D-Drucken, die keine Originalteile von 3D Systems sind, kann zum Erlöschen der Gewährleistung führen. • Die Beaufsichtigung durch Erwachsene ist erforderlich; beobachten Sie Kinder genau und greifen Sie gegebenenfalls ein, um mögliche Sicherheitsprobleme zu vermeiden und um sicherzustellen, dass der Cube-Drucker ordnungsgemäß verwendet wird. Stellen Sie sicher, dass kleine 3D-Drucke für kleine Kinder nicht zugänglich sind. Diese 3D-Drucke bergen eine mögliche Erstickungsgefahr für kleine Kinder. • Wenn der Cube-Drucker in Betrieb ist, wird die Druckdüse am Print Jet heiß; vermeiden Sie es, diesen Bereich zu berühren, bis er abgekühlt ist. • Verändern Sie während des Druckens nicht die Farbe des Materials; dadurch kann der Cube-Drucker beschädigt werden. ! Gefahr durch heiße Oberflächen: Eine heiße Oberfläche ist in der Nähe dieses Zeichens oder am Print Jet zugänglich; vermeiden Sie jeglichen Kontakt. Heiße Oberflächen können schwere Verbrennungen verursachen. Achtung: Weist darauf hin, dass etwas passieren kann, was zum Verlust von Daten, zu Schäden an den Geräten oder zu Körperverletzungen führen könnte. Achtung: Weist auf eine Einklemmgefahr hin, die zu Körperverletzungen führen könnte. 4 2A WICHTIGE SICHERHEITSINFORMATIONEN (DE) SÍMBOLOS Y DEFINICIONES DE SEGURIDAD PAUTAS DE SEGURIDAD • Siga todas las normas de seguridad de esta sección y esté atento a todas las precauciones y advertencias en esta guía. • No modifique ninguna medida de seguridad ni realice modificaciones a la impresora Cube. Hacerlo está prohibido y anula la garantía. • El uso de materiales de impresión o de piezas impresas en 3D que no sean componentes auténticos de 3D Systems puede anular la garantía. • Se requiere la supervisión de un adulto; observe a los niños de cerca e intervenga cada vez que sea necesario para prevenir cualquier posible problema de seguridad y asegurar el uso adecuado de la impresora Cube. Asegúrese de que las piezas pequeñas impresas en 3D no estén al alcance de niños pequeños. Estas piezas impresas en 3D representan un posible peligro de asfixia para los niños pequeños. • Cuando la impresora Cube está funcionando, la punta de impresión se calienta; evite tocar esta área hasta que se haya enfriado. • No cambie el color de los materiales durante la impresión; esto puede dañar la impresora Cube. ! Peligro de superficie caliente: Una superficie caliente se encuentra cerca de esta señal o en la impresora; evite el contacto con esta. Las superficies calientes pueden causar quemaduras graves. Precaución: Indica que algo puede ocurrir y causar una pérdida de datos, daños al equipo o lesiones. Precaución: Indica un peligro de punto de pellizco que podría causar lesiones a una persona. 5 2B INFORMACIÓN IMPORTANTE SOBRE SEGURIDAD (ES) SYMBOLES DE SÉCURITÉ ET DÉFINITIONS CONSIGNES DE SÉCURITÉ • Suivez toutes les consignes de sécurité de cette section et respectez tous les messages d’avertissement du présent manuel. • Ne pas modifier les dispositifs de sécurité ou apporter des modifications au Cube. Cela est interdit et annule la garantie. • L’utilisation de matériaux d’impression ou d’imprimés en 3D autres que les composants 3D Systems d’origine peut entraîner l’annulation de la garantie. • La supervision d’un adulte est requise ; surveillez les enfants en restant à proximité et intervenez si nécessaire pour éviter tout problème de sécurité potentiel et garantir la bonne utilisation du Cube. Assurez-vous que les petits imprimés en 3D restent hors de portée des jeunes enfants. Ces imprimés en 3D présentent un risque d’étouffement pour les jeunes enfants. • Lorsque le Cube fonctionne, l’extrémité de l’imprimante peut chauffer ; évitez de toucher cette zone jusqu’à ce qu’elle est refroidie. • Ne pas modifier la couleur du matériau pendant l’impression ; cela peut endommager le Cube. ! Surface chaude: Une surface chaude est présente à proximité de ce symbole ou sur l’imprimante ; évitez de la toucher. Les surfaces chaudes peuvent causer de graves brûlures. Attention: Indique que quelque chose pourrait occasionner une perte des données, des dommages sur l’équipement ou des blessures. Attention: Indique un risque de pincement qui pourrait occasionner une blessure. 6 2C INFORMATIONS IMPORTANTES RELATIVES À LA SÉCURITÉ (FR) SIMBOLI E DEFINIZIONI DI SICUREZZA LINEE GUIDA DI SICUREZZA • Attenersi a tutte le norme di sicurezza contenute in questa sezione e rispettare le avvertenze e le precauzioni indicate nella guida. • Non alterare le caratteristiche di sicurezza né apportare modifiche a Cube. Queste operazioni sono vietate e determinano l’annullamento della garanzia. • L’utilizzo di materiali di stampa o di componenti di stampa 3D non originali di 3D Systems può comportare l’annullamento della garanzia. • È richiesta la supervisione di un adulto; tenere sotto stretta sorveglianza i bambini e, in caso di necessità, intervenire per evitare eventuali problemi di sicurezza e per garantire l’uso appropriato di Cube. Assicurarsi che i bambini non possano raggiungere i componenti più piccoli delle stampe 3D, poiché questi potrebbero costituire un pericolo di soffocamento. • Durante l’attività di Cube, la puntina di stampa del getto d’inchiostro si riscalda; evitare il contatto con quest’area fino al completo raffreddamento. • Non modificare il colore del materiale durante la stampa; quest’operazione potrebbe causare danni a Cube. ! Pericolo di superficie calda: In prossimità di questo simbolo o del getto d’inchiostro è presente una superficie calda; evitare il contatto. Le superfici calde possono causare ustioni gravi. Attenzione: Segnala che un determinato evento potrebbe causare la perdita di dati, danni all’impianto o lesioni personali. Attenzione: Segnala la presenza di una zona ad alto rischio che potrebbe causare lesioni personali. 7 2D INFORMAZIONI IMPORTANTI SULLA SICUREZZA (IT) VEILIGHEIDSSYMBOLEN EN DEFINITIES VEILIGHEIDSVOORSCHRIFTEN • Volg alle veiligheidsvoorschriften in dit gedeelte en neem alle opmerkingen en waarschuwingen in deze handleiding in acht. • Breng geen wijzigingen aan de veiligheidsvoorzieningen of de Cube aan. Dit is verboden en de garantie komt hierdoor te vervallen. • Door gebruik van printmaterialen of 3D-prints anders dan originele 3D Systems-onderdelen kan de garantie komen te vervallen. • Oezicht door een volwassene is vereist; houd kinderen nauwlettend in het oog en grijp zo nodig in om mogelijke veiligheidsproblemen te voorkomen en het juiste gebruik van de Cube te verzekeren. Zorg dat jonge kinderen niet bij kleine 3D-prints kunnen komen. Deze 3D-prints houden potentieel verstikkingsgevaar in voor jonge kinderen. • Wanneer de Cube in werking is, wordt de printtip op de Print Jet heet. Raak dit oppervlak niet aan totdat het is afgekoeld. • Wijzig de kleur van het materiaal niet tijdens het printen, anders kan de Cube worden beschadigd. ! Gevaar voor hete oppervlakken: Er is een toegankelijk heet oppervlak in de nabijheid van dit waarschuwingsteken of bij de Print Jet. Niet aanraken. Hete oppervlakken kunnen ernstige brandwonden veroorzaken. Let op: Duidt op iets wat kan gebeuren dat verlies van gegevens, beschadiging van de apparatuur of lichamelijk letsel kan veroorzaken. Let op: Duidt op een gevaarlijk knelpunt dat lichamelijk letsel kan veroorzaken. 8 2E BELANGRIJKE VEILIGHEIDSINFORMATIE (DU) SIKKERHEDSSYMBOLER OG -DEFINITIONER SIKKERHEDSRETNINGSLINIER • Følg alle sikkerhedsreglerne i denne sektion og overhold forsigtighedsregler og advarsler i denne vejledning. • Sikkerhedsfeatures må ikke modificeres og der må ikke foretages andre modifikationer til Cube. Det er forbudt at gøre det og det vil ugyldiggøre garantien. • Brugen af andre printmaterialer eller 3D prints end de ægte 3D Systems komponenter kan ugyldiggøre garantien. • Det er påkrævet at en voksen holder opsyn; hold nøje øje med børn og træd ind hvis det er nødvendigt for at undgå potentielle sikkerhedsproblemer og at sikre at Cube bruges rigtigt. Sørg for at små børn ikke har adgang til små 3D prints. Disse 3D prints kan udgøre en kvælningsfare for små børn. • Når Cube er i drift bliver spidsen på printaggregatet varm, så dette område må ikke røres før det er kølet af. • Materialets farve må ikke ændres under trykning; det kan beskadige Cube. ! Fare for varm overflade: Der findes en varm overflade i nærheden af dette skilt eller ved printaggregatet; undgå kontakt. Varme overflader kan forårsage alvorlige forbrændinger. Forsigtig: Viser at der kunne ske noget som kan resultere i datatab, beskadigelse af udstyr eller personskade. Forsigtig: Viser at der findes en knusefare, som kan forårsage personskade. 9 2F VIGTIG SIKKERHEDSINFORMATION (DA) 安全関連シンボルと定義 SIK 安全ガイドライン • 本項の安全ルールのすべてに従い、また本書のすべての要注意および警告事項を守ってください。 • 安全機能を修正したり、Cube に改変を加えたりしないでください。そうすることは禁じられており、保証は無効になります。 • 純正 3D Systems コンポーネント以外のプリント材料や 3D プリントの使用は、保証を無効にする場合があります。 • 大人の監視が必要です。子供が使用している時には身近で見守り、安全上の問題を未然に防ぎ、Cube が適切に使用されるよう、必要に応じて介入するようにしてください。小型の 3D プリントが幼児の手に届くことのないようにしてください。これらの 3D プリントは幼児にとっては、のどを詰まらせる危険物となります。 • Cube が動作中は、プリントジェット上のプリントチップが熱くなります。その部分が冷めるまでは、触れないようにしてください。 • プリント中に材料の色を変更しないでください。そうすると Cube を損傷させることがあります。 ! 高熱面の危険: このシンボルサインの近くまたはプリントジェット部には表面が高熱となっている部分があります。触れないように注意してください。高熱面で火傷することになります。 要注意: データの喪失、機器の損傷、または人身傷害を引き起こすような何かが起こる場合もあることを示します。 要注意: 人身傷害を引き起こす可能性のある危険個所を示します。 10 2G 安全に関する重要情報 (JP) The Print Jet print tip heats the material and produces a thin flowing material of plastic creating layers that adhere to the Print Pad. After each layer is produced, the Print Pad lowers so that a new layer can be drawn on top of the last. This process continues until the last layer on the top of the creation is jetted. CUBE 3D PRINTER FEATURES • Material Cartridge • Durable, ABS & PLA Plastic • 25 free 3D print creations • USB & WI-FI connectivity CUBE 3D PRINTER PROPERTIES Technology: Plastic Jet Printing (PJP) Print Jets: Single jet Max. Creation Size: 5.5” x 5.5” x 5.5” (140 x 140 x 140 mm) Material: Tough recyclable plastic Layer Thickness: 10 mil | 0.01 inches 250 microns | 0.25 mm Supports: Fully automated; easy to peel off Cartridge: 1 Cartridge prints 13 to 14 mid-sized creations Material Colors: See Cubify.com for color choices 11 3 CUBE 3D PRINTER FEATURES A Cube Tube B Print Jet C Print Pad D Operator’s Touchscreen F Material Cartridge G Fuse H Power Cord Connection I USB Port* J USB Connection** E ON/OFF Push-Button Power Switch & Menu Function RIGHT SIDE *For firmware update only. **For loading Cube build files only. A B C D E F BACK G H I J 12 4 AT A GLANCE WEIGHT & DIMENSIONS • Weight (without cartridge): 4.3 kg (9.5 lbs.) SOFTWARE • Complimentary software for Windows and Mac OSX. This application converts your 3D model into layered slices (G-code), ready for printing on your machine. MINIMUM HARDWARE REQUIREMENTS • A PC with these minimum requirements will be required to run the Cubify Software • Processor: Multi-core processor - 2 GHz or faster per core • System RAM: 2 GB • Screen Resolution: 1024x768 WINDOWS REQUIREMENTS • Cubify Software runs on 32 and 64-bit Operating Systems • Windows XP Professional or Home Edition with Service Pack 3 • Windows 7 • Windows is required for ad-hoc WiFi Print Job submission. NOTE: Ad-Hoc WI-FI connection will not connect to Windows XP operating systems • If not already installed, the Cubify Software installer will automatically install the Microsoft .NET 4.0 Framework WIRELESS OPTIONS • 802.11b/g with: WLAN Infrastructure or Ad hoc Mode. NON-WIRELESS OPTION • USB Memory Stick, to transfer print files (supplied with the Cube) MAC OSX REQUIREMENTS • Cubify Software runs on Mac OSX 10.8 ELECTRICAL REQUIREMENTS • Outlet requirements: 100-240 Volts, at 50/60 Hz. • Cube electrical rating: 24V DC, 3.75 amp. MATERIAL STORAGE • All polymers degrade with time. The following conditions ensure the material remains high quality: • Do not unpack until material is needed. • Store at room temperature: 16-29° C (60 - 85° F) OPERATING ENVIRONMENT • Room Temperature: 16-29° C (60 - 85° F) • Nozzle- 280°C (536°F) • Print Pad- 66-77° C (150-170°F) 13 11.25” (28.6 cm) 8” (20.3 cm) 8” (20.3 cm) 8” (20.3 cm) 8” (20.3 cm) 10” (25.4 cm) 13” (33 cm) 5 REQUIREMENTS FOR YOUR CUBE WALL Print Pad USB Cable** Cube Tube USB Stick Quick Start Guide CubeStick™ Unclog Tool*** The Cube Power Supply Material Cartridge* Power Cord cubify.com *Neon green included **For downloading firmware ***For unclogging filament in Print Jet SETTING UP YOUR CUBE 1. Remove the white plastic inserts from each side of the box by pinching the tabs in and bending the inserts out. Pull to remove both from the box. Lift top of carton up and remove. 2. Lift foam from the top of Cube. Remove foam from the midsection and below the plate by gently pulling out from the front. Remove Cube from box. 3. Lift foam that Cube was sitting on to access power supply and other accessories. 4. Remove the Print Pad from the foam under the Cube and set it aside. 5. Plug round connector of power supply into the back of the Cube. 6. Plug power cord into power supply, then into outlet or power strip. 14 6 UNPACKING AND SETTING UP YOUR CUBE Creating an account on Cubify.com is easy and gives you access to all of the great designs and collections that will drive and inspire your creativity. Once you have set up an account you will be able to activate your Cube by entering your Cube’s individual serial number in the designated bar under the Activate My Cube tab. An activation code will then be sent to the email account you used to register. This code will be used to unlock your Cube so that you can get your 3D printing underway. NOTE: If you do not receive an activation code email, please check your spam filter settings. 1. Go to Cubify.com and log in to your account or click Sign Up to create an account. 2. Click on Activate My Cube from My Cubify drop down menu. 3. Enter your 12-character serial number (digits and letters found on the back of your Cube), then click Activate. 4. Your Cube’s serial number is now activated on Cubify.com and your Cube will be assigned to your Cubify account. Your activation code will be displayed, and also sent to you in an e-mail along with links to help you get started creating. Make sure the email doesn’t get caught in your spam filter! 5. Using the links from your activation code email, or the My Cubes section in your account, you can download: • The Cube Software, which turns your 3D models into a file that the Cube can print. The software must be installed on your computer. Your Cube won’t print without it! • 25 free, ready-to-print 3D files created by top artists and designers (also on your USB stick) • Cube 3D Printer User Guide (also on your USB stick) • Material Safety Data Sheets 6. Now it is time to unlock your Cube so you can begin creating. NOTE: If you use a PC, you need Microsoft® Windows® 7 or XP (SP3+). If you use a Mac, you need Mac OSX 10.8+ 15 7 LINK YOUR CUBE TO YOUR CUBIFY ACCOUNT Once you’ve linked your Cube to your Cubify account, you can activate your Cube to start printing! 1. Your activation code to unlock your Cube will appear on the “Congratulations” screen after you activate your Cube in your Cubify account. A copy of the code should also be sent to your email. 2. Make sure your Cube is plugged in. Press the button on the front panel to turn it on. It may take a few seconds for it to warm up. 3. Tap on the touchscreen to view the Unlock Cube screen. 4. Enter your activation code and then tap on the check box to unlock your Cube. Your Cube will automatically transition to the main menu touchscreen and is now unlocked and ready to create! Download the Cube Software and experiment with it to create your own designs. 1. To download Cube Software, go to My Cubify and click on My Accounts. Click on My Downloads and select Cube software on the menu page. 2. Click on Setup.exe in the Cube Client.zip. The next screen will ask you if you would like to open this file. Click Yes. 3. The Cubify Setup Wizard will guide you through the steps to install the software on your computer. Click Next to launch the Setup Wizard. • It is necessary to download the Cube software in order to be able to import .stl or .creation files and convert them to .cube files. • The .cube file is a machine-specific file type that is coded for the Cube to read and print. 16 8 UNLOCK YOUR CUBE 9 DOWNLOAD AND INSTALL CUBE SOFTWARE FOR WINDOWS Download the Cube Software and experiment with it to create your own designs. 1. In the menu, Click on Download Cube Software. Double click on the zip file and the Cubify Mac installer.dmg will appear. 2. Double click Cubify Mac Installer dmg, to make a new window open. Drag to install into the Applications folder. 3. If the “Drag to install” window did not open, locate the Cubify driver in your device panel. Click on Cubify to download software. NOTE: Safari users will not have to unzip the dmg; it unzips as it downloads. • It is necessary to download the Cube software in order to be able to import .stl or .creation files and convert them to .cube files. • The .cube file is a machine-specific file type that is coded for the Cube to read and print. Follow these simple steps to access your 25 free creations: 1. Click on Download Free Creations to start your immediate download. 2. Click Extract in the Creation zip file and extract it to your computer. There are a couple of file types involved in 3D printing with your Cube: • .stl file: A 3-dimensional solid computer-aided design (CAD) file that defines the geometry to be printed. This is the typical file type output of a 3D design program or software tool. • .creation file: These files represent the pre-made 3D models that are optimized for your Cube and available on Cubify for download. Simply import your .creation file into the Cube software and click Build to prepare your creation for print. • .cube file: This is the machine code read by the Cube 3D Printer that enables it to build your print. 17 10 DOWNLOAD AND INSTALL CUBE SOFTWARE FOR MAC OSX 11 DOWNLOAD YOUR FREE CREATIONS Cubify Software allows you manipulate your design before sending it to the Cube for printing. It simulates the Cube’s Print Pad so you can orient, scale, re-size and rotate the 3D print to get the best result when printing. Become familiar with the software and understand the functions before starting your first print. IMPORT Select your creation file. CENTER Center your creation on the Print Pad; multiple items are auto-positioned. BUILD Creates the Cube file by adding support structures; the slicing process will prepare your creation. PRINT Transfers your creation to the Cube. HEAL Fix your creation. ORIENT & SCALE Rotate the X, Y, Z axis; move creation back & forth or side to side; increase or decrease your creation. These icons permit you to view your creation in various view points. After selecting a particular view, place your cursor on the part and hold the mouse button down; rotate the Print Pad to view different angles of the part. Isometric Right Left Front Back Top Bottom COLOR* Select your background and model color. SAVE Saves your creation without printing. UNLOAD Deletes your creation from the Cube software. CONFIGURE Connect your computer and the Cube via WI-FI using the USB cable. Also, download firmware upgrades. SETTINGS Choose your print size; turn rafts & supports ON or OFF. MODEL INFO Displays your creation information. * The color you select will not affect the color of your printed creation 18 12 CUBIFY SOFTWARE OVERVIEW SET UP A COMPUTER (AD HOC) NETWORK An ad hoc network is a temporary connection to your Cube through your computer or wireless smart phone. Ad hoc networks can only be wireless, so you must have a wireless network adapter installed in your computer to setup or join an ad hoc network. NOTE: Ad hoc will not connect to operating systems running Windows XP. 1. Tap on Setup in the main menu and scroll until you come to the WI-FI Setup screen. 2. Tap WI-FI Setup. 3. Tap on Ad Hoc in the menu screen. The next screen will show Ad Hoc mode enabled meaning it is ready to connect to your computer. 4. Open your Internet Access located in your Network Settings. NOTE: Always select your Cube from your network settings before opening Cube Software. 5. Select your Cube on the menu. Click on Connect. This may take a few minutes. 7. Open your Cube software and click on “Configure” and select your Cube from the menu. Click on “Connect;” once connected, you may begin to print your creation. 19 13 WI-FI SETUP SET UP A COMPUTER (WLAN) NETWORK WLAN (wireless local area network) allow easy establishment of a secure wireless network. WLAN connection will allow you to connect to your printer through the Cube Software. 1. Tap on Setup in the main menu and scroll until you come to the WI-FI Setup. 2. Tap on WI-FI Setup. 3. Tap on WLAN in the menu screen. “Acquiring networks” screen appears, meaning it is searching for available networks to connect to the printer. 4. Tap the on the touch screen to select your network. If your network is password protected, enter your password and tap on “OK.” 5. After your WI-FI is set up, if needed, the WI-FI firmware may update to the latest version if the latest firmware is not installed. NOTE: This will only occur when you are connecting to the network for the first time and if the WI-FI firmware needs updating. Once it is updated, the network will be successfully connected. 6. If the WI-FI firmware update was necessary, you may get one of two responses: (1) Network connected update successful; (2) Network connected... Update not successful. NOTE: If WiFi firmware update reported unsuccessful, the WLAN WiFi connection should still work. The WiFi firmware will try again with next connection attempt. 7. Click the function button to take you back to the main menu before printing 8. Open your Cube software and click on “Configure” and select your Cube from the menu. Click on “Connect;” once connected, you may begin to print your creation. 9. Open your Internet Access located on your computer’s taskbar. 10. Scroll and select the network on the menu. Click on Connect. This may take a few minutes. If you are not connected to a network, you can download your creations from your computer and save them to your USB Memory Stick for printing. • When printing with the USB Memory Stick, the file must be in the main folder located on the memory stick. If it is put into another folder it cannot be accessed from Cube screen. • Install your USB Memory Stick into your computer’s USB connection and save your creation to your USB Memory Stick. • When you are ready to print your creation, install your memory stick into the Cube’s USB connection and print your creations. 1. Tap on Setup on the Cube touch screen and scroll to Update Firmware. 2. If your USB Memory Stick is connected to the Cube, please remove the memory stick, (the screen will display the directions to do this). Tap the NEXT arrow on the display; locate the USB cable and connect the cable to the Cube USB port located on the back of the Cube (again, these instructions will be displayed on the screen). Connect the other end of cable to your computer’s USB port. NOTE: Never leave the USB cable connected except when upgrading the firmware. 3. Tap the NEXT arrow. SETTINGS SAVED will be displayed at the top of the screen. You will have 6 seconds to press and hold the control button until firmware loader appears on the screen. If the button is released before the firmware screen appears, the screen will go blank. Repeat Step 1 to restart the update firmware process. 4. Continue to hold button until Settings Saved and then Cube Firmware Loader appears on the touch screen. You are now in the correct mode to update your firmware. NOTE: Even though the update is unsuccessful, WLAN WI-FI should still work, To try again, simply reconnect to the network, 5. Open Cubify Software and click the Configure icon; click on Load New Cube Firmware. Once you see the pop-up message Firmware Updated Successfully, click OK. 6. Disconnect the USB cable from your Cube and computer; unplug the Cube power cord. Plug power cord back in and press the Cube power button; a message will display on the touchscreen “Setting restored,” click on the check mark to save settings. This does not change your previous settings that you have made. Your new firmware is now installed. 14 NON-WIRELESS COMPUTER SETUP 15 UPDATE CUBE FIRMWARE SWITCHING FROM AD-HOC TO WLAN CONNECTION (OR VICE VERSA) 1. Close out Cube Software. 2. Disconnect the Cube’s wireless connection through the Internet Access menu on your operating system. 3. Now follow steps for WLAN / ad-hoc set up. 4. Re-open your software after your Cube successfully connects to WLAN / ad-hoc. There must be an appropriate distance between the print tip and the Print Pad to make sure the first printed layer sticks properly. When the Cube is operating, the print tip on the Print Jet nozzle becomes hot; avoid touching this area until it has cooled down. 1. Make sure the textured side of the Print Pad is facing upwards. Place the magnetic side of the print pad onto the magnetic print plate. 2. Press Set Up on your Cube’s touchscreen and tap Next until Set Gap appears. 3. Select Set Gap. The Print Jet and Print Pad will move into position to set the gap. Place a standard piece of paper between the print tip and Print Pad. If the paper cannot slide between the tip and pad, tap the down arrow on the touchscreen until you can slide the paper between the tip and pad. 4. Press the Z up key to move the Print Pad toward the print tip until the paper is tight between the Print Pad and print tip. Tap the down key until the paper slides back and forth with no resistance. 5. Tap the check mark to save this setting to the printer’s memory. If the setting is saved, the “Gap” value will be applied for each subsequent print. If you wish to cancel this setting, tap the X box. 6. If you have any trouble printing after the gap setting has been saved, you can use this procedure at any time to change the gap. 7. Once the gap has been set, tap PREV to start the Load Cartridge process. ! 21 16 SETTING PRINT JET GAP 1. Use scissors to remove the cartridge from its packaging. 2. Tap Setup on your Cube’s touchscreen, select Load Cartridge, and press Next. 3. Remove the thumbscrew from the side of the Material Cartridge and remove the blue tape from the material. Save the screw to reinstall into cartridge if material was not fully used during printing. This will prevent filament from unravelling during storage. Press Next to begin cartridge installation. Do not pull out plastic until the thumbscrew is removed from the cartridge. Failure to remove the thumbscrew will damage the cartridge. 7. Filament will then be drawn into the Print Jet until a small amount comes out of the heated tip at the bottom. Carefully dispose of the filament without touching the hot print tip. Press Next. (If nothing comes out of the print tip, remove the plastic from the top of the Print Jet and repeat these installation steps.) WARNING: PRINT TIP BECOMES EXTREMELY HOT DURING SET-UP AND OPERATION. DO NOT TOUCH PRINT TIP. 8. Insert Cube Tube (the clear tube that’s already around the filament at the top) into the Print Jet. Press Next and your cartridge is loaded and ready to go! 4. Tilt cartridge so that bottom of cartridge is resting on cartridge holder. 5. Slide cartridge in the holder ensuring that top of cartridge feeder is seated into the Cube feeder. Do not slide cartridge without tilting, doing so can cause damage to t he Cube Feeder. 6. Insert the filament coming from the tube into the hole at the top of the Print Jet. Press Next, and continue to push filament into the Print Jet until it feeds through on its own. ! ! 22 17 MATERIAL CARTRIDGE INSTALLATION NOTE: CubeStick must be applied to the Print Pad every time you start a new creation. CAUTION: Do not use any adhesive other than the CubeStick. Doing so can damage your print tip and the Print Pad. Make sure you use the 2nd generation CubeStick for your 2nd generation Cube. WARNING: APPLY ADHESIVE JUST PRIOR TO STARTING A PRINT. ADHESIVE WILL DRY IF YOU WAIT TO START A PRINT AFTER APPLICATION. ADHESIVE SHOULD STILL BE WET AT START OF PRINT. 1. Find the CubeStick in the original packaging. 2. Remove Print Pad from the Cube. Apply a thin, even coat of adhesive on the entire top surface of the Print Pad. (Don’t over-apply!) Check the Print Pad after applying the adhesive to make sure there aren’t any gaps where the adhesive was not applied. If you find any gaps, apply adhesive to those areas. Never put glue on side with magnet. 3. Replace CubeStick cap after use. 4. Place Print Pad on the Print Pad platform. ! ! 23 18 PRINTING PREPARATION NOTE: Your Cube will have printed test creations before leaving the factory. These test creations may have been printed in a different material color than you are using. Therefore, a small amount of material may be remaining in the Print Jet. The start of your first creation may have some of this material color until it transitions over to your material color. WARNING: PRINT JET NOZZLE TIP BECOMES EXTREMELY HOT DURING SET-UP AND OPERATION. DO NOT TOUCH NOZZLE. Do not change color of material or cartridge during printing; doing so may damage the Cube. To get started printing your first creation; the creation files that were downloaded when you activated your account or a .stl file will need to be converted to a .Cube file. This is the machine code file that the Cube printer will read to build your creation. WI-FI CONNECTION if you wished to connect to WI-FI via Ad hoc or WLAN connection, please refer to Section 14: WI-FI Set-Up and follow the step by step instructions. Then resume these instructions and follow the steps below to connect to your Cube. These steps apply to both Ad Hoc and WLAN connections. 1. Open Cube Software and click on Configure. Click on your Cube model in the dialog box and click on Connect. Your computer is now connected to your Cube. 2. Click on Import and select a file from the creation files that you downloaded from Cubify.com or a .stl file. The creation that you selected will appear on Print Pad. The software’s Print Pad is a simulation of your Cube Print Pad. 3. In the software you can orient, scale, and manipulate your creation to ensure that you obtain the optimum printing results. Refer to Section 13: Cubify Software Overview to understand the different functions your software provides. 4. To prepare your creation file for printing, click on Build and save your creation as a .cube file. PRINTING WITH WI-FI CONNECTION 1. If have a WI-FI connection, click on Print and select the Cube file saved from your creation file; click on Open to send your file to the Cube. 2. On the Cube touchscreen, finger tap check box to start the build file or tap the X box to cancel. 3. The Cube will begin the heating process; this will take a few minutes. During the heating process, the print tip and the Print Pad will reach the temperatures that have been preset by the manufacturer. Once these temperatures are reached, your creation will begin to print. NOTE: During printing operation, please do not place a cover over your Cube. The Cube generates heat during printing and if covered, it can cause damage to the Cube. ! 24 19 PRINTING YOUR FIRST CREATION PRINTING YOUR FIRST CREATION (continued) PRINTING FROM THE USB MEMORY STICK 1. If you are using your USB Memory Stick to download your Cube file, please install the USB Memory Stick into the USB port on your computer. Open the file folder where your Cube file is located and save it on the USB Memory Stick. Remove memory stick and install it into the Cube’s USB port. 2. Tap Print on the touchscreen and use left or right arrows until you see your Cube file displayed. 3. Tap on your .cube file; the print touchscreen will appear showing the total time it will take to print your creation and the time left to complete it during printing. It also shows the maximum height of your creation and the current height while printing. 4. The Cube will begin the heating process that will take a few minutes. During the heating process, the print tip and the Print Pad will reach the temperatures that have been preset by the manufacturer. Once these temperatures are reached, your creation will begin to print. Once the creation is finished printing, your touch screen will display Print Finished, press the check box to confirm. Wait until the Print Pad is completely cool before removing your print. WARNING: PRINT JET NOZZLE TIP BECOMES EXTREMELY HOT DURING SET-UP AND OPERATION. DO NOT TOUCH NOZZLE. Do not change color of material or cartridge during printing; doing so may damage the Cube. ABORTING YOUR PRINT To abort your creation during warm up or printing, finger tap the STOP button on touchscreen. The next screen will read Are you sure you want to abort the print? Finger tap the check box to abort or tap the X box to cancel. Wait until the print tip completely cools before touching. ! 25 Do not change color of material or cartridge during printing; doing so may damage the Cube. 1. After being instructed by the touchscreen, pull Cube Tube away from top of Print Jet. Do not pull the material out until the touch screen instructs you to do so. 2. Pull material out of Print Jet and press Next. NOTE: After the material is pulled out from the print jet, material debris may be visible at the print tip that could clog the tip. Using the unclog tool, insert the tool’s point into the Print Jet; if debris is present, it will protrude from the print tip. 3. Tilt cartridge down to clear Cube feeder and remove cartridge from cube; remove the Cube Tube from material. Replace thumb screw into cartridge to hold the unused material in place. Refer to Step 18: Material Cartridge Installation for instructions on how to install your new cartridge. 26 20 REPLACING MATERIAL CARTRIDGE ! 1. Click on My Cubify and select My Account. 2. Click on My Downloads. 3. If you would like to download a file to save on your computer or print using Cube software, click on DOWNLOAD. The file will download to the Cube software on your computer to save your creation. 4. Your file can download directly to your Cube by clicking on PRINT ON MY CUBE; the Select Cube Printer screen will appear and will search for available Cube printers on your network. 5. If a Cube was not discovered in your network, you can either save the .cube file to your computer or connect through an ad hoc network (refer to PRINTING WITH WI-FI CONNECTION, page 25). 6. If a Cube is discovered in your network, you have a choice of printing now or decline printing. If you select print now, your print will start printing on the Cube you selected in the network. REMOVING YOUR CREATION FROM THE PRINT PAD • Place the Print Pad with your creation in a container (not supplied) filled with regular warm tap water and let it soak approximately five minutes or until your creation eases free from the Print Pad. Clean and dry the Print Pad before reuse. • To clean your Print Pad, rinse under tap water and dry with a lint-free wipe. REMOVING RAFTS AND SUPPORTS (IF REQUIRED) • Use small pliers to remove supports and rafts. In places where the supports are inside your creation and are hard to get to, use small wire snips (not provided) to remove the supports. 27 21 CLOUD PRINTING FORM CUBIFY.COM 22 FINISHING FIRST CREATION CLEANING THE EXTERIOR • Clean the Cube’s exterior with a lint free cloth and water. Dampen the cloth with water and wipe the outer surfaces of any debris that is visible. CLEANING THE PRINT PAD • Submerge the Print Pad in a container filled with warm tap water. Please do not use well water; it contains certain minerals that may make your creation difficult to remove from the Print Pad. • Rinse pad under tap water and dry using a lint free wipe. CLEANING THE PRINT TIP • Using the small pliers, pull away any plastic debris away from the Print Jet tip (if the debris is stubborn, the Print Jet may need to be heated to make the debris soft enough to remove). CLEANING THE TOUCHSCREEN WARNING: PRINT JET NOZZLE TIP BECOMES EXTREMELY HOT DURING SET-UP AND OPERATION. DO NOT TOUCH NOZZLE. • Wipe the touchscreen with the soft, lint-free cloth. Do not spray cleaners on the touchscreen. 4 AMP FUSE REPLACEMENT CAUTION: Before replacing the fuse, switch Cube’s power off and unplug power cord. 1. 4 Amp, 250 V fast blow fuse is located on the back left side of your Cube. 2. Using a flat screwdriver, turn screwdriver counterclockwise to remove fuse from the fuse holder. 3. Using a flat screwdriver, turn screwdriver clockwise to install new fuse into the fuse holder. After new fuse is installed, plug power cord into outlet and switch power on. ! ! 28 23 MAINTAINING YOUR CUBE Your Print Pad may become un-leveled when transporting your Cube. If this occurs, please follow these instructions to level your Print Pad for optimum creation experience. The Print Pad pad can be leveled with the two adjusting bolts located underneath the front and back of the pad. 1. Press the power button on the Cube control panel. 2. Ensure Print Pad is properly installed on print plate. 3. Finger tap SETUP on the touchscreen and scroll through the menu until LEVEL PLATE appears. Press LEVEL PLATE; the Print Pad will move up to Print Jet tip. 4. Using the Up arrow, raise the Print Pad to the Print Jet tip until the tip is near the Print Pad but not touching it. 5. Touch the “clockwise” and “counter-clockwise” buttons to automatically move the Print Jet around the four corners of the Print Pad. During each movement along the sides of the Print Pad, observe any gaps between the pad and the Print Jet. If there are gaps on one side of Print Pad and the Print Jet tip is barely touching the Print Pad on the other side, the Print Pad pad is not level. NOTE: Please refer to the FRONT TO BACK or SIDE TO SIDE instructions on how to turn the adjusting knob bolts to level the pad properly. 6. Place a standard sheet of paper between the Print Jet tip and the Print Pad where the adjustment needs to be made. If the paper cannot slide between the Print Jet tip and Print Pad, tap the down arrow key on touchscreen. ROLL SIDE TO SIDE If the gap between the Print Pad and Print Jet tip is at the right or left side, adjust the pad using the rear adjusting knob bolt (B). If the gap is too low, the pad is too low and will need to be moved up. 1. If the pad needs to be moved up, turn the front adjusting knob bolt (A) clockwise to loosen and then turn the rear adjusting knob bolt (B) clockwise. To move the pad down, turn the rear adjusting knob bolt (B) counterclockwise. 2. While adjusting, slide the paper between the Print Jet tip and Print Pad where the adjustment was needed; the paper should slide back and forth with no resistance. If resistance is felt, the pad is too high. Please repeat the adjustment. 3. Once adjustments are made, tighten the adjusting knob bolts. TILT FRONT TO BACK If the gap between the Print Pad and Print Jet tip is at the front or back of pad, adjust the pad using the rear adjusting knob bolt (B). If the gap appears on the side, the pad is too low and will need to be moved up. 1. If the pad needs to be moved up, turn the front adjusting knob bolt (A) clockwise to loosen and then turn the back adjusting knob bolt (B) clockwise. To move the pad down, turn the rear adjusting knob bolt (B) counterclockwise. 2. While adjusting, slide the paper between the Print Jet tip and Print Pad where the adjustment was needed; the paper should slide back and forth with no resistance. If resistance is felt, the pad is too high. Please repeat the adjustment. 3. Once adjustments are made, tighten the adjusting knob bolts. NOTE: After the print jet is adjusted, you will need to reset the print jet gap; please refer to Section 16: Setting Print Jet Gap. A B 29 24 PRINT PAD LEVELING INSTRUCTIONS 3D Systems, Inc. 333 Three D Systems Circle | Rock Hill, SC | 29730 Cubify.com ©2013 3D Systems, Inc. All rights reserved. The 3D Systems logo, Cube and Cubify are registered trademarks of 3D Systems, Inc. pn 350341-01, Rev. C This document was generated on 01/08/2014 PLEASE CHECK WWW.MOLEX.COM FOR LATEST PART INFORMATION Part Number: 43375-1001 Status: Active Overview: Sabre™ Power Connector Description: Sabre™ Crimp Terminal, Female, Double 18 AWG, 4.57mm Max. Insulation Diameter, Reel Packaged, Tin (Sn) Plated Brass Contact with TPA Documents: Drawing (PDF) Product Literature (PDF) RoHS Certificate of Compliance (PDF) General Product Family Crimp Terminals Series 43375 Application Power Comments For double crimping of 18 AWG wire in a side-by-side orientation. Terminal mates to 3.18mm wide x 0.51mm ) thick flat blade PC tab. Allows 44441 receptacle housings to comply with the UL1977 finger proof access requirement Crimp Quality Equipment Yes MolexKits Yes Overview Sabre™ Power Connector Product Literature Order No 987650-5662 Product Name Sabre™ UPC 800754365994 Physical Durability (mating cycles max) 25 Gender Female Material - Metal Brass Material - Plating Mating Tin Material - Plating Termination Tin Net Weight 0.360/g Packaging Type Reel Plating min - Mating 0.508μm Plating min - Termination 0.508μm Termination Interface: Style Crimp or Compression Wire Insulation Diameter 4.57mm max. Wire Size AWG 14, 16, 18+18 Wire Size mm² N/A Electrical Current - Maximum per Contact 18A Voltage - Maximum 600V Material Info Reference - Drawing Numbers Product Specification RPSX-44441-001 Sales Drawing SD-43375-1001 Series image - Reference only EU RoHS China RoHS ELV and RoHS Compliant REACH SVHC Contains SVHC: No Low-Halogen Status Low-Halogen Need more information on product environmental compliance? Email productcompliance@molex.com For a multiple part number RoHS Certificate of Compliance, click here Please visit the Contact Us section for any non-product compliance questions. Search Parts in this Series 43375Series Mates With 43178 Male Crimp Terminals Use With 44441 Receptacle Housings Application Tooling | FAQ Tooling specifications and manuals are found by selecting the products below. Crimp Height Specifications are then contained in the Application Tooling Specification document. Global Description Product # Manual Extraction Tool 0638130500 Terminator Die - Doubles 0638405200 Hand Crimp Tool for Flat Blade Crimp Terminal 0638117300 Extraction Tool 0638132700 Mini-Mac™ Applicator 0638916100 Mini-Mac™ Applicator, For Narrow Insulation Crimp 0638917000 This document was generated on 01/08/2014 PLEASE CHECK WWW.MOLEX.COM FOR LATEST PART INFORMATION This document was generated on 01/22/2014 PLEASE CHECK WWW.MOLEX.COM FOR LATEST PART INFORMATION Part Number: 43031-0002 Status: Active Overview: Micro-Fit 3.0™ Connectors Description: Micro-Fit 3.0™ Crimp Terminal, Male, with Gold (Au) Plated Tin/Brass Alloy Contact, 20-24 AWG, Reel Documents: Drawing (PDF) RoHS Certificate of Compliance (PDF) Product Specification PS-43045 (PDF) Product Literature (PDF) Test Summary TS-43045-002 (PDF) General Product Family Crimp Terminals Series 43031 Application Power Crimp Quality Equipment Yes Overview Micro-Fit 3.0™ Connectors Packaging Alternative 43031-0008 (Loose) Product Literature Order No 987650-5984 Product Name Micro-Fit 3.0™ UPC 800754369411 Physical Gender Male Material - Metal Phosphor Bronze Material - Plating Mating Gold Material - Plating Termination Tin Net Weight 0.061/g Packaging Type Reel Plating min - Mating 0.381μm Plating min - Termination 2.540μm Termination Interface: Style Crimp or Compression Wire Insulation Diameter 1.85mm max. Wire Size AWG 20, 22, 24 Wire Size mm² N/A Material Info Reference - Drawing Numbers Product Specification PS-43045, RPS-43045-003, RPS-43045-004 Sales Drawing SD-43031-**** Test Summary TS-43045-002 Series image - Reference only EU RoHS China RoHS ELV and RoHS Compliant REACH SVHC Contains SVHC: No Low-Halogen Status Low-Halogen Need more information on product environmental compliance? Email productcompliance@molex.com For a multiple part number RoHS Certificate of Compliance, click here Please visit the Contact Us section for any non-product compliance questions. Search Parts in this Series 43031Series Mates With 43030 Application Tooling | FAQ Tooling specifications and manuals are found by selecting the products below. Crimp Height Specifications are then contained in the Application Tooling Specification document. Global Description Product # Extraction Tool 0011030043 Insertion Tool for Crimp Terminal 0638120800 Hand Crimp Tool 0638190000 FineAdjust™ Applicator for Insulation OD 1.30-1.85mm - 20-24 AWG 0639004500 FineAdjust™ Applicator for 0639018800 Insulation OD 1.10-1.30mm - 20-24 AWG FineAdjust™ Applicator for Insulation OD 0.91-1.09mm - 20-24 AWG 0639018900 T2 Terminator™ for insulation OD 1.30-1.85mm - 20-24 AWG 0639104500 T2 Terminator™ for insulation OD 1.10-1.30mm - 20-24 AWG 0639118800 T2 Terminator™ for insulation OD 0.91-1.09mm - 20-24 AWG 0639118900 This document was generated on 01/22/2014 PLEASE CHECK WWW.MOLEX.COM FOR LATEST PART INFORMATION This document was generated on 04/14/2014 PLEASE CHECK WWW.MOLEX.COM FOR LATEST PART INFORMATION Part Number: 39-28-8060 Status: Active Overview: Mini-Fit Jr.™ Power Connectors Description: Mini-Fit® Jr. Header, Dual Row, Vertical, without Snap-in Plastic Peg PCB Lock, 6 Circuits, PA Polyamide Nylon 6/6 94V-0, Tin (Sn) Plating, without Drain Holes Documents: 3D Model Packaging Specification PK-5566-003 (PDF) Drawing (PDF) Test Summary TS-5556-002 (PDF) Product Specification PS-5556-001 (PDF) RoHS Certificate of Compliance (PDF) Agency Certification CSA LR19980 UL E29179 General Product Family PCB Headers Series 5566 Application Power, Wire-to-Board Comments The 5566 header should be used with standard Mini- Fit® female terminals. If increased amperage of up to 13A per circuit is needed, please consider using the Mini-Fit® Plus HCS family 45750 terminals with 46015 headers; . See Molex Product specification PS-5666-001 for current de-rating information. Overview Mini-Fit Jr.™ Power Connectors Product Name Mini-Fit Jr.™ UPC 800753580732 Physical Breakaway No Circuits (Loaded) 6 Circuits (maximum) 6 Color - Resin Natural Durability (mating cycles max) 30 First Mate / Last Break No Flammability 94V-0 Glow-Wire Compliant No Guide to Mating Part No Keying to Mating Part None Lock to Mating Part Yes Material - Metal Brass Material - Plating Mating Tin Material - Plating Termination Tin Material - Resin Nylon Net Weight 1.778/g Number of Rows 2 Orientation Vertical PC Tail Length 3.50mm PCB Locator Yes PCB Retention None PCB Thickness - Recommended 1.60mm Packaging Type Bag Pitch - Mating Interface 4.20mm Pitch - Termination Interface 4.20mm Polarized to Mating Part Yes Series image - Reference only EU RoHS China RoHS ELV and RoHS Compliant REACH SVHC Contains SVHC: No Low-Halogen Status Low-Halogen Need more information on product environmental compliance? Email productcompliance@molex.com For a multiple part number RoHS Certificate of Compliance, click here Please visit the Contact Us section for any non-product compliance questions. Search Parts in this Series 5566Series Mates With 5557 Mini-Fit Jr.™ Receptacle Housing Polarized to PCB Yes Shrouded Fully Stackable No Surface Mount Compatible (SMC) No Temperature Range - Operating -40°C to +105°C Termination Interface: Style Through Hole Electrical Current - Maximum per Contact 9A Voltage - Maximum 600V Solder Process Data Duration at Max. Process Temperature (seconds) 5 Lead-free Process Capability Wave Capable (TH only) Max. Cycles at Max. Process Temperature 1 Process Temperature max. C 260 Material Info Old Part Number 5566-06A-210 Reference - Drawing Numbers Packaging Specification PK-5566-003 Product Specification PS-5556-001, RPS-5557-036, RPS-5557-058 Sales Drawing SD-5566-002 Test Summary TS-5556-002 This document was generated on 04/14/2014 PLEASE CHECK WWW.MOLEX.COM FOR LATEST PART INFORMATION 1. General description The UHF EPCglobal Generation 2 standard allows the commercialized provision of mass adoption of UHF RFID technology for passive smart tags and labels. Main fields of applications are supply chain management and logistics for worldwide use with special consideration of European, US and Chinese frequencies to ensure that operating distances of several meters can be realized. The G2X is a dedicated chip for passive, intelligent tags and labels supporting the EPCglobal Class 1 Generation 2 UHF RFID standard. It is especially suited for applications where operating distances of several meters and high anti-collision rates are required. The G2X is a product out of the NXP Semiconductors UCODE product family. The entire UCODE product family offers anti-collision and collision arbitration functionality. This allows a reader to simultaneously operate multiple labels / tags within its antenna field. A UCODE G2X based label/ tag requires no external power supply. Its contact-less interface generates the power supply via the antenna circuit by propagative energy transmission from the interrogator (reader), while the system clock is generated by an on-chip oscillator. Data transmitted from interrogator to label/tag is demodulated by the interface, and it also modulates the interrogator’s electromagnetic field for data transmission from label/tag to interrogator. A label/tag can be operated without the need for line of sight or battery, as long as it is connected to a dedicated antenna for the targeted frequency range. When the label/tag is within the interrogator’s operating range, the high-speed wireless interface allows data transmission in both directions. In addition to the EPC specifications the G2X offers an integrated EAS (Electronic Article Surveillance) feature and read protection of the memory content. On top of the specification of the G2XL the G2XM offers 512-bit of user memory. SL3ICS1002/1202 UCODE G2XM and G2XL Rev. 3.8 — 11 November 2013 139038 Product data sheet COMPANY PUBLIC 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 2 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 2. Features and benefits 2.1 Key features 512-bit user memory (G2XM only) 240-bit of EPC memory 64-bit tag identifier (TID) including 32-bit unique serial number Memory read protection EAS (Electronic Article Surveillance) command Calibrate command 32-bit kill password to permanently disable the tag 32-bit access password to allow a transition into the secured transmission state Broad international operating frequency: from 840 MHz to 960 MHz Long read/write ranges due to extremely low power design Reliable operation of multiple tags due to advanced anti-collision Forward link: 40-160 kbit/s Return link: 40-640 kbit/s 2.2 Key benefits High sensitivity provides long read range Low Q-factor for consistent performance on different materials Improved interference suppression for reliable operation in multi-reader environment Large input capacitance for ease of assembly and high assembly yield Highly advanced anti-collision resulting in highest identification speed Reliable and robust RFID technology suitable for dense reader and noisy environments 2.3 Custom commands EAS Alarm Enables the UHF RFID tag to be used as EAS tag without the need for a backend data base. Read Protect Protects all memory content including CRC16 from unauthorized reading. Calibrate Activates permanent back-scatter in order to evaluate the tag-to-reader performance. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 3 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 3. Applications Supply chain management Item level tagging Asset management Container identification Pallet and case tracking Product authentication Outside above mentioned applications, please contact NXP Semiconductors for support. 4. Ordering information Table 1. Ordering information G2XM Type number Package Name Description Version SL3ICS1002FUG/V7AF Wafer Bumped die on sawn wafer - SL3S1002FTB1 XSON3 plastic extremely thin small outline package;3 terminals; body 1 x 1.45 x 0,5 mm SOT1122 Table 2. Ordering information G2XL Type number Package Name Description Version SL3ICS1202FUG/V7AF Wafer Bumped die on sawn wafer - SL3S1202FTB1 XSON3 plastic extremely thin small outline package;3 terminals; body 1 x 1.45 x 0,5 mm SOT1122 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 4 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 5. Block diagram The SL3ICS1002/1202 IC consists of three major blocks: - Analog RF Interface - Digital Controller - EEPROM The analog part provides stable supply voltage and demodulates data received from the reader for being processed by the digital part. Further, the modulation transistor of the analog part transmits data back to the reader. The digital section includes the state machines, processes the protocol and handles communication with the EEPROM, which contains the EPC and the user data. Fig 1. Block diagram of G2X IC 001aai335 MOD DEMOD VREG VDD data in data out R/W ANALOG RF INTERFACE PAD PAD RECT DIGITAL CONTROL ANTENNA ANTICOLLISION READ/WRITE CONTROL ACCESS CONTROL EEPROM INTERFACE CONTROL RF INTERFACE CONTROL EEPROM MEMORY SEQUENCER CHARGE PUMP 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 5 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 6. Wafer layout and pinning information 6.1 Wafer layout (1) X-scribe line width: 56.4 m (2) Y-scribe line width: 56.4 m (3) Chip step, x-length: 488.0 m (4) Chip step, y-length: 470,0 m (5) Bump to bump distance X (TP1 - RFN): 351,0 m (6) Bump to bump distance Y (RFN - RFP): 333,0 m (7) Distance bump to metal sealring X: 40,3 m (8) Distance bump to metal sealring Y: 40,3 m Bump size X x Y: 60 m x 60 m Fig 2. Wafer layout and pinning information not to scale! 001aai346 (1) (7) (2) (8) (5) (6) (4) (3) Y X TP2 TP1 RFN RFP 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 6 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 7. Package outline Fig 3. Package outline SOT1122 Outline References version European projection Issue date IEC JEDEC JEITA SOT1122 MO-252 sot1122_po Unit mm max nom min 0.50 0.04 0.55 0.425 0.30 0.25 0.22 0.35 0.30 0.27 A(1) Dimensions Notes 1. Dimension A is including plating thickness. 2. Can be visible in some manufacturing processes. SOT1122 A1 D 1.50 1.45 1.40 1.05 1.00 0.95 E e e1 0.55 0.50 0.47 0.45 0.40 0.37 b b1 L L1 09-10-09 XSON3: plastic extremely thin small outline package; no leads; 3 terminals; body 1 x 1.45 x 0.5 mm D E e1 e A1 b1 L1 L e1 0 1 2 mm scale 3 1 2 b 4× (2) 4× (2) A pin 1 indication type code terminal 1 index area 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 7 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL Table 3. Pin description of SOT1122 Symbol Pin Description RFP 1 Ungrouded antenna connector RFN 2 Grounded antenna connector n.c. 3 not connected Table 4. SOT1122 Marking Type Type code (Marking) Comment SL3S1202FTB1 UL UCODE G2XL SL3S1002FTB1 UM UCODE G2XM 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 8 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 8. Mechanical specification 8.1 Wafer specification See Ref. 20 “Data sheet - Delivery type description – General specification for 8” wafer on UV-tape with electronic fail die marking, BL-ID document number: 1093**”. 8.1.1 Wafer • Designation: each wafer is scribed with batch number and wafer number • Diameter: 200 mm (8”) • Thickness: 150 m ± 15 m • Number of pads 4 • Pad location: non diagonal/ placed in chip corners • Distance pad to pad RFN-RFP 333.0 μm • Distance pad to pad TP1-RFN: 351.0 μm • Process: CMOS 0.14 μm • Batch size: 25 wafers • Dies per wafer: 120.000 8.1.2 Wafer backside • Material: Si • Treatment: ground and stress release • Roughness: Ra max. 0.5 m, Rt max. 5 m 8.1.3 Chip dimensions • Die size without scribe: 0.414 mm x 0.432 mm = 0.178 mm2 • Scribe line width: x-dimension:56.4 m (width is measured on top metal layer) y-dimension: 56.4 m (width is measured on top metal layer) 8.1.4 Passivation on front • Type Sandwich structure • Material: PE-Nitride (on top) • Thickness: 1.75 m total thickness of passivation 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 9 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 8.1.5 Au bump • Bump material: > 99.9% pure Au • Bump hardness: 35 – 80 HV 0.005 • Bump shear strength: > 70 MPa • Bump height: 18 m • Bump height uniformity: – within a die: ± 2 m – within a wafer: ± 3 m – wafer to wafer: ± 4 m • Bump flatness: ± 1.5 m • Bump size: – RFP, RFN 60 x 60 m – TP1, TP2 60 x 60 m – Bump size variation: ± 5 m • Under bump metallization: sputtered TiW 8.1.6 Fail die identification No inkdots are applied to the wafer. Electronic wafer mapping (SECS II format) covers the electrical test results and additionally the results of mechanical/visual inspection. See Ref. 20 “Data sheet - Delivery type description – General specification for 8” wafer on UV-tape with electronic fail die marking, BL-ID document number: 1093**” 8.1.7 Map file distribution See Ref. 20 “Data sheet - Delivery type description – General specification for 8” wafer on UV-tape with electronic fail die marking, BL-ID document number: 1093**” 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 10 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 9. Limiting values [1] Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any conditions other than those described in the Operating Conditions and Electrical Characteristics section of this specification is not implied. [2] This product includes circuitry specifically designed for the protection of its internal devices from the damaging effects of excessive static charge. Nonetheless, it is suggested that conventional precautions be taken to avoid applying greater than the rated maxima. [3] For ESD measurement, the die chip has been mounted into a CDIP20 package. Table 5. Limiting values[1][2] In accordance with the Absolute Maximum Rating System (IEC 60134) Voltages are referenced to RFN Symbol Parameter Conditions Min Max Unit Die Tstg storage temperature range -55 +125 C Toper operating temperature -40 +85 C VESD electrostatic discharge voltage Human body model [3] - 2 kV SOT1122 Tstg storage temperature range -55 +125 C Ptot total power dissipation - 30 mW Toper operating temperature -40 +85 C VESD electrostatic discharge voltage Human body model - 2 kV 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 11 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 10. Characteristics 10.1 Wafer characteristics [1] Power to process a Query command [2] Measured with a 50 source impedance [3] At minimum operating power [4] Values measured for a 40 kHz phase reserval command under matched conditions 10.2 Package characteristics [1] Measured with network analyzer at 915 MHz; values at 0.5 dBm after peakmax of on-set of die, measured in the center of the pads. Table 6. Wafer characteristics Symbol Parameter Conditions Min Typ Max Unit Memory characteristics tRET EEPROM data retention Tamb 55 C 50 - - year NWE EEPROM write endurance Tamb 55 C 100000 - - cycle Interface characteristics Ptot total power dissipation - 30 mW foper operating frequency 840 - 960 MHz Pmin minimum operating power supply [1][2] - -15 - dBm Ci input capacitance (parallel) [3] - 0.88 - pF Q quality factor (Im (Zchip) / Re (Zchip)) [3] - 9 - - Z impedance (915 MHz) - 22 - j195 - - modulated jammer suppression 1.0 MHz [4] - - 4 - dB - unmodulated jammer suppression 1.0 MHz [4] - - 4 - dB Table 7. Package interface characteristics Symbol Parameter Conditions Min Typ Max Unit Interface characteristics SOT1122 Ci input capacitance (parallel) [1] - 1.02 - pF Z SOT1122 impedance (915 MHz) - 18.6 - j171.2 - 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 12 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 11. Packing information 11.1 Wafer See Ref. 20 “Data sheet - Delivery type description – General specification for 8” wafer on UV-tape with electronic fail die marking, BL-ID document number: 1093**”. 11.2 SOT1122 Part orientation T1. For details please refer to http://www.standardics.nxp.com/packaging/packing/pdf/sot886.t1.t4.pdf. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 13 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 12. Functional description 12.1 Power transfer The interrogator provides an RF field that powers the tag, equipped with a UCODE G2X. The antenna transforms the impedance of free space to the chip input impedance in order to get the maximum possible power for the G2X on the tag. The RF field, which is oscillating on the operating frequency provided by the interrogator, is rectified to provide a smoothed DC voltage to the analog and digital modules of the IC. The antenna that is attached to the chip may use a DC connection between the two antenna pads. Therefore the G2X also enables loop antenna design. Possible examples of supported antenna structures can be found in the reference antenna design guide. 12.2 Data transfer 12.2.1 Reader to G2X Link An interrogator transmits information to the UCODE G2X by modulating an RF signal in the 840 MHz - 960 MHz frequency range. The G2X receives both information and operating energy from this RF signal. Tags are passive, meaning that they receive all of their operating energy from the interrogator's RF waveform. An interrogator is using a fixed modulation and data rate for the duration of at least an inventory round. It communicates to the G2X by modulating an RF carrier using DSB-ASK, SSB-ASK or PR-ASK with PIE encoding. For further details refer to Section 17, Ref. 1, section 6.3.1.2. Interrogator-to-tag (R=>T) communications. 12.2.2 G2X to reader Link An interrogator receives information from the UCODE G2X by transmitting a continuous-wave RF signal to the tag; the G2X responds by modulating the reflection coefficient of its antenna, thereby generating modulated sidebands used to backscatter an information signal to the interrogator. The system is a reader talks first (RTF) system, meaning that a G2X modulates its antenna reflection coefficient with an information signal only after being directed by the interrogator. G2X backscatter is a combination of ASK and PSK modulation depending on the tuning and bias point. The backscattered data is either modulated with FM0 baseband or Miller sub carrier. For further details refer to Section 17, Ref. 1, section 6.3.1.3. tag-to-interrogator (T=>R) communications. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 14 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 12.3 Operating distances RFID tags based on the UCODE G2X silicon may achieve maximum operating distances according the following formula: (1) (2) [1] CEPT/ETSI regulations [CEPT1], [ETSI1]. [2] New CEPT/ETSI regulations. [ETSI3]. [3] FCC 47 part 15 regulation [FCC1]. [4] These read distances are maximum values for general tags and labels. Practical usable values may be lower due to damping by object materials and environmental conditions. A special tag antenna design can help achieve higher values. The typical write range is > 50% of the read range. Table 8. Symbol description Symbol Description Unit Ptag minimum required RF power for the tag W Gtag gain of the tag antenna - EIRP transmitted RF power m wavelength m Rmax maximum achieved operating distance for a /2-dipole m loss factor assumed to be 0.5 considering matching and package losses - R distance m Table 9. Operating distances for UCODE G2X based tags and labels in released frequency bands Frequency range Region Available power Calculated read distance single antenna [4] Unit 868.4 to 868.65 MHz (UHF) Europe [1] 0.5 W ERP 3.6 m 865.5 to 867.6 MHz (UHF) Europe [2] 2 W ERP 7.1 m 902 to 928 MHz (UHF) America [3] 4 W EIRP 7.5 m Ptag EIRP Gtag 4R ---------- 2 = Rmax EIRP Gtag 2 42Ptag = --------------------------------------- 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 15 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 12.4 Air interface standards The G2X is certified according EPCglobal 1.0.9 and fully supports all parts of the "Specification for RFID Air Interface EPCglobal, EPCTM Radio-Frequency Identity Protocols, Class-1 Generation-2 UHF RFID, Protocol for Communications at 860 MHz - 960 MHz, Version 1.1.0". EPCglobal compliance and interoperability certification 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 16 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 13. Physical layer and signaling 13.1 Reader to G2X communication 13.1.1 Physical layer For interrogator-to-G2X link modulation refer to Section 17, Ref. 1, annex H.1 Baseband waveforms, modulated RF, and detected waveforms. 13.1.2 Modulation An interrogator sends information to one or more G2X by modulating an RF carrier using double-sideband amplitude shift keying (DSB-ASK), single-sideband amplitude shift keying (SSB-ASK) or phase-reversal amplitude shift keying (PR-ASK) using a pulse-interval encoding (PIE) format. The G2X receives the operating energy from this same modulated RF carrier. Section 17, Ref. 1: Annex H, as well as chapter 6.3.1.2.2. The G2X is capable of demodulating all three modulation types. 13.1.3 Data encoding The R=>T link is using PIE. For the definition of the therefore relevant reference time interval for interrogator-to-chip signaling (Tari) refer to Section 17, Ref. 1, chapter 6.3.1.2.3. The Tari is specified as the duration of a data-0. 13.1.4 Data rates Interrogators shall communicate using Tari values between 6.25 s and 25 s, inclusive. For interrogator compliance evaluation the preferred Tari values of 6.25 s, 12.5 s or 25 s should be used. For further details refer to Section 17, Ref. 1, chapter 6.3.1.2.4. 13.1.5 RF envelope for R=>T A specification of the relevant RF envelope parameters can be found in Section 17, Ref. 1, chapter 6.3.1.2.5. 13.1.6 Interrogator power-up/down waveform For a specification of the interrogator power-up and power-down RF envelope and waveform parameters refer to Section 17, Ref. 1, chapters 6.3.1.2.6 and 6.3.1.2.7. 13.1.7 Preamble and frame-sync An interrogator shall begin all R=>T signaling with either a preamble or a frame-sync. A preamble shall precede a Query command and denotes the start of an inventory round. For a definition and explanation of the relevant R=>T preamble and frame-sync refer to Section 17, Ref. 1, chapter 6.3.1.2.8. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 17 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 13.2 G2X to reader communication An interrogator receives information from a G2X by transmitting an unmodulated RF carrier and listening for a backscattered reply. The G2X backscatters by switching the reflection coefficient of its antenna between two states in accordance with the data being sent. For further details refer to Section 17, Ref. 1, chapter 6.3.1.3. 13.2.1 Modulation The UCODE G2X communicates information by backscatter-modulating the amplitude and/or phase of the RF carrier. Interrogators shall be capable of demodulating either demodulation type. 13.2.2 Data encoding The encoding format, selected in response to interrogator commands, is either FM0 baseband or Miller-modulated subaltern. The interrogator commands the encoding choice 13.2.2.1 FM0 baseband FM0 inverts the baseband phase at every symbol boundary; a data-0 has an additional mid-symbol phase inversion. For details on FM0 and generator state diagram, FM0 symbols and sequences and how FM0 transmissions should be terminated refer to Section 17, Ref. 1, chapter 6.3.1.3. 13.2.2.2 FM0 Preamble T=>R FM0 signaling begin with one of two defined preambles, depending on the value of the TRext bit specified in the Query command that initiated the inventory round. For further details refer to Section 17, Ref. 1, chapter 6.3.1.3. 13.2.2.3 Miller-modulated sub carrier Baseband Miller inverts its phase between two data-0s in sequence. Baseband Miller also places a phase inversion in the middle of a data-1 symbol. For details on Miller-modulated sub carrier, generator state diagram, sub carrier sequences and terminating sub carrier transmissions refer to Section 17, Ref. 1, chapter 6.3.1.3. 13.2.2.4 Miller sub carrier preamble T=>R sub carrier signaling begins with one of the two defined preambles. The choice depends on the value of the TRext bit specified in the Query command that initiated the inventory round. For further details refer to Section 17, Ref. 1, chapter 6.3.1.3. 13.2.3 Data rates The G2X IC supports tag to interrogator data rates and link frequencies as specified in Section 17, Ref. 1, chapter 6.3.1.3. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 18 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 13.3 Link timing For the interrogator interacting with a UCODE G2X equipped tag population exact link and response timing requirements must be fulfilled, which can be found in Section 17, Ref. 1, chapter 6.3.1.6. 13.3.1 Regeneration time The regeneration time is the time required if a G2X is to demodulate the interrogator signal, measured from the last falling edge of the last bit of the G2X response to the first falling edge of the interrogator transmission. This time is referred to as T2 and can vary between 3.0 Tpri and 20 Tpri. For a more detailed description refer to Section 17, Ref. 1, chapter 6.3.1.6. 13.3.2 Start-up time For a detailed description refer to Section 17, Ref. 1, chapter 6.3.1.3.4. 13.3.3 Persistence time An interrogator chooses one of four sessions and inventories tags within that session (denoted S0, S1, S2, and S3). The interrogator and associated UCODE G2X population operate in one and only one session for the duration of an inventory round (defined above). For each session, a corresponding inventoried flag is maintained. Sessions allow tags to keep track of their inventoried status separately for each of four possible time-interleaved inventory processes, using an independent inventoried flag for each process. Two or more interrogators can use sessions to independently inventory a common UCODE G2X chip population. A session flag indicates whether a G2X may respond to an interrogator. G2X chips maintain a separate inventoried flag for each of four sessions; each flag has symmetric A and B values. Within any given session, interrogators typically inventory tags from A to B followed by a re-inventory of tags from B back to A (or vice versa). Additionally, the G2X has implemented a selected flag, SL, which an interrogator may assert or deassert using a Select command. For a description of Inventoried flags S0 – S3 refer to Section 17, Ref. 1 chapter 6.3.2.2 and for a description of the Selected flag refer to Section 17, Ref. 1, chapter 6.3.2.3. For tag flags and respective persistence time refer to Section 17, Ref. 1, table 6.14. 13.4 Bit and byte ordering The transmission order for all R=>T and T=>R communications respects the following conventions: • within each message, the most-significant word is transmitted first, and • within each word, the most-significant bit (MSB) is transmitted first, whereas one word is composed of 16 bits. To represent memory addresses and mask lengths EBV-8 values are used. An extensible bit vector (EBV) is a data structure with an extensible data range. For a more detailed explanation refer to Section 17, Ref. 1, Annex A. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 19 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 13.5 Data integrity The G2X ignores invalid commands. In general, "invalid" means a command that (1) is incorrect given the current the G2X state, (2) is unsupported by the G2X, (3) has incorrect parameters, (4) has a CRC error, (5) specifies an incorrect session, or (6) is in any other way not recognized or not executable by the G2X. The actual definition of "invalid" is state-specific and defined, for each G2X state, in n Section 17, Ref. 1 Annex B and Annex C. All UCODE G2X backscatter error codes are summarized in Section 17, Ref. 1 Error codes, Annex I. For a detailed description of the individual backscatter error situations which are command specific please refer to the Section 17, Ref. 1 individual command description section 6.3.2.10. 13.6 CRC A CRC-16 is a cyclic-redundancy check that an interrogator uses when protecting certain R=>T commands, and the G2X uses when protecting certain backscattered T=>R sequences. To generate a CRC-16 an interrogator or the G2X first generates the CRC-16 precursor shown in Section 17, Ref. 1 Table 6.11, then take the ones-complement of the generated precursor to form the CRC-16. For a detailed description of the CRC-16 generation and handling rules refer to Section 17, Ref. 1, chapter 6.3.2.1. The CRC-5 is only used to protect the Query command (out of the mandatory command set). It is calculated out of X5 + X3 + 1. For a more detailed CRC-5 description refer to Section 17, Ref. 1, table 6.12. For exemplary schematic diagrams for CRC-5 and CRC-16 encoder/decoder refer to Section 17, Ref. 1, Annex F. For a CRC calculation example refer to Section 15.1, Table 27 and Table 28. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 20 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14. TAG selection, inventory and access This section contains all information including commands by which a reader selects, inventories, and accesses a G2X population An interrogator manages UCODE G2X equipped tag populations using three basic operations. Each of these operations comprises one or more commands. The operations are defined as follows Select: The process by which an interrogator selects a tag population for inventory and access. Interrogators may use one or more Select commands to select a particular tag population prior to inventory. Inventory: The process by which an interrogator identifies UCODE G2X equipped tags. An interrogator begins an inventory round by transmitting a Query command in one of four sessions. One or more G2X may reply. The interrogator detects a single G2X reply and requests the PC, EPC, and CRC-16 from the chip. An inventory round operates in one and only one session at a time. For an example of an interrogator inventorying and accessing a single G2X refer to Section 17, Ref. 1, Annex E. Access: The process by which an interrogator transacts with (reads from or writes to) individual G2X. An individual G2X must be uniquely identified prior to access. Access comprises multiple commands, some of which employ one-time-pad based cover-coding of the R=>T link. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 21 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.1 G2X Memory For the general memory layout according to the standard Section 17, Ref. 1, refer to Figure 6.17. The tag memory is logically subdivided into four distinct banks. In accordance to the standard Section 17, Ref. 1, section 6.3.2.1. The tag memory of the SL3ICS1002 G2XM is organized in following 4 memory sections: The logical address of all memory banks begin at zero (00h). Table 10. G2X memory sections Name Size Bank Reserved memory (32 bit ACCESS and 32 bit KILL password) 64 bit 00b EPC (excluding 16 bit CRC-16 and 16 bit PC) 240 bit 01b TID (including unique 32 bit serial number) 64 bit 10b User memory (G2XM only) 512 bit 11b Fig 4. G2X TID memory structure Serial Number Model Number Mask-Designer Identifier Class Identifier TID 0 31 0 11 0 11 0 7 3Fh 20h 1Fh 14h 13h 08h 07h 00h 0 6 0 4 1Fh 19h 18h 14h Version Number Sub Version Number 00000001h to FFFFFFFFh 006h E2h Whenever the 32 bit serial is exceeded the sub version is incremented by 1 Addresses 3Fh 00h Addresses Addresses Bits Bits LS Byte LSBit MSBit LSBit MSBit MS Byte LSBit MSBit LSBit MSBit 0000010b 00000b Sub Version Nr Version (Silicon) Nr Model Nr. Mask ID UCode EPC G2XM 00000b 0000011b 003h 006h UCode EPC G2XL 00000b 0000100b 004h 006h 002h 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 22 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.1.1 Memory map [1] This is the initial memory content when delivered by NXP Semiconductors [2] G2XL: HEX 3005 FB63 AC1F 3841 EC88 0467 G2XM: HEX 3005 FB63 AC1F 3681 EC88 0468 [3] only G2XM Table 11. Memory map Bank address Memory address Type Content Initial [1] Remark Bank 00 00h – 1Fh Reserved kill password: refer to Section 17, Ref. 1, chapter 6.3.2.1.1 all 00h unlocked memory 20h – 3Fh Reserved access password: refer to Section 17, Ref. 1, chapter 6.3.2.1.1 all 00h unlocked memory Bank 01 00h – 0Fh EPC CRC-16: refer to Section 17, Ref. 1, chapter 6.3.2.1.2 memory mapped calculated CRC 10h – 14h EPC Backscatter length: refer to Section 17, Ref. 1, chapter 6.3.2.1.2 00110b unlocked memory 15h EPC Reserved for future use: refer to Section 17, Ref. 1, chapter 6.3.2.1.2 0b unlocked memory 16h EPC Reserved for future use: refer to Section 17, Ref. 1, chapter 6.3.2.1.2 0b hardwired to 0 17h –1Fh EPC Numbering system indicator: refer to Section 17, Ref. 1, chapter 6.3.2.1.2 00h unlocked memory 20h - 10Fh EPC EPC: refer to Section 17, Ref. 1, chapter 6.3.2.1.2 [2] unlocked memory Bank 10 00h – 07h TID allocation class identifier: refer to Section 17, Ref. 1, chapter 6.3.2.1.3 1110 0010b locked memory 08h – 13h TID tag mask designer identifier: refer to Section 17, Ref. 1, chapter 6.3.2.1.3 0000 0000 0110b locked memory 14h – 1Fh TID tag model number: refer to Section 17, Ref. 1, chapter 6.3.2.1.3 TMNR locked memory 20h – 3Fh TID serial number: refer to [Section 17, Ref. 1, chapter 6.3.2.1.3 SNR locked memory Bank 11[3] 00h – 1FFh User user memory: refer to [Section 17, Ref. 1, chapter 6.3.2.1.4 undefined unlocked memory 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 23 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.1.1.1 User memory (only G2XM) The User Memory bank contains a sequential block of 512 bits (32 words of 16 bit) ranging from address 00h to 1Fh. The user memory can be accessed via Select, Read or Write command and it may be write locked, permanently write locked, unlocked or permanently unlocked. In addition reading of not only of the User Memory but of the whole memory including EPC and TID can be protected by using the custom ReadProtect command. 14.1.1.2 Special behavior of user memory address 1Fh WRITE or SELECT of user memory address 1Fh will falsely set an error flag. This will affect the subsequent READ or SELECT. The following commands will falsely set an internal error flag (without actually causing an error): 1) WRITE to user memory with WordPtr=1Fh 2) SELECT to user memory with compare mask ending at bitaddress 1FFh (e.g. Pointer=1FEh, length=1 or Pointer=1FDh, length=2 …) Note: The error flag is set independent of the chip state (also chips in the e.g. Ready state are affected). The falsely set error flag will affect the following sub sequential commands: A) READ command with WordCount=0 falsely responds with "memory overrun" error B) SELECT command with Length<>0 falsely assumes non existing memory location The behavior can be avoided with: • Turning off the RF carrier to reset the chip (This is what readers typically do!). • Using the READ command with WordCount<>0. • Sending other command prior to READ or SELECT (e.g. WRITE to address<>1Fh, ReqRN) or executing READ or SELECT two times. Remark: The WRITE operation itself is not affected by this problem i.e. data is written properly! With commercially available readers this behavior is typically not observed. 14.1.1.3 Supported EPC types The EPC types are defined in the EPC Tag Standards document from EPCglobal. These standards define completely that portion of EPC tag data that is standardized, including how that data is encoded on the EPC tag itself (i.e. the EPC Tag Encodings), as well as how it is encoded for use in the information systems layers of the EPC Systems Network (i.e. the EPC URI or Uniform Resource Identifier Encodings). The EPC Tag Encodings include a Header field followed by one or more Value Fields. The Header field indicates the length of the Values Fields and contains a numbering system identifier (NSI). The Value Fields contain a unique EPC Identifier and optional Filter Value when the latter is judged to be important to encode on the tag itself. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 24 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.2 Sessions, selected and inventoried flags Session, Selected and Inventory Flags are according the EPCglobal standard. For a description refer to Section 17, Ref. 1, section 6.3.2.3. 14.2.1 G2X States and slot counter For a description refer to Section 17, Ref. 1, section 6.3.2.4. 14.2.2 G2X State Diagram The tag state are according the EPCglobal standard please refer to: Section 17, Ref. 1, section 6.3.2.4 Tag states and slot counter. A detailed tag state diagram is shown in Section 17, Ref. 1, figure 6.19. Refer also to Section 17, Ref. 1, Annex B for the associated state-transition tables and to Section 17, Ref. 1, Annex C for the associated command-response tables. 14.3 Managing tag populations For a detailed description on how to manage an UCODE G2X tag populations refer to Section 17, Ref. 1, chapter 6.3.2.6. 14.4 Selecting tag populations For a detailed description of the UCODE G2X tag population selection process refer to Section 17, Ref. 1, section 6.3.2.7. 14.5 Inventorying tag populations For a detailed description on accessing individual tags based on the UCODE G2X refer to Section 17, Ref. 1, section 6.3.2.8. 14.6 Accessing individual tags For a detailed description on accessing individual tags based on the UCODE G2X refer to Section 17, Ref. 1, section 6.3.2.9. An example inventory and access of a single UCODE G2X tag is shown in Section 17, Ref. 1, Annex E.1. 14.7 Interrogator commands and tag replies For a detailed description refer to Section 17, Ref. 1, section 6.3.2.10. 14.7.1 Commands An overview of interrogator to tag commands is located in Section 17, Ref. 1, Table 6.16. Note that all mandatory commands are implemented on the G2X according to the standard. Additionally the optional command Access is supported by the G2X (for details refer to Section 14.11 “Optional Access Command”). Besides also custom commands are implemented on the G2X (for details refer to Section 14.12 “Custom Commands”. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 25 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.7.2 State transition tables The G2X responses to interrogator commands are defined by State Annex B transition tables in Section 17, Ref. 1. Following states are implemented on the G2X: • Ready, for a description refer to Section 17, Ref. 1, Annex B.1. • Arbitrate, for a description refer to Section 17, Ref. 1, Annex B.2. • Reply, for a description refer to Section 17, Ref. 1, Annex B.3. • Acknowledged, for a description refer to Section 17, Ref. 1, Annex B.4. • Open, for a description refer to Section 17, Ref. 1, Annex B.5. • Secured, for a description refer to Section 17, Ref. 1, Annex B.6. • Killed, for a description refer to Section 17, Ref. 1, Annex B.7. 14.7.3 Command response tables The G2X responses to interrogator commands are described in following Annex C sections of Section 17, Ref. 1: • Power-up, for a description refer to Section 17, Ref. 1, Annex C.1. • Query, for a description refer to Section 17, Ref. 1, Annex C.2. • QueryRep, for a description refer to Section 17, Ref. 1, Annex C.3. • QueryAdjust, for a description refer to Section 17, Ref. 1, Annex C.4. • ACK, for a description refer to Section 17, Ref. 1, Annex C.5. • NAK, for a description refer to Section 17, Ref. 1, Annex C.6. • Req_RN, for a description refer to Section 17, Ref. 1, Annex C.7. • Select, for a description refer to Section 17, Ref. 1, Annex C.8. • Read, for a description refer to Section 17, Ref. 1, Annex C.9. • Write, for a description refer to Section 17, Ref. 1, Annex C.10. • Kill, for a description refer to Section 17, Ref. 1, Annex C.11. • Lock, for a description refer to Section 17, Ref. 1, Annex C.12. • Access, for a description refer to Section 17, Ref. 1, Annex C.13. • T2 time-out, for a description refer to Section 17, Ref. 1, Annex C.17. • Invalid command, for a description refer to Section 17, Ref. 1, Annex C.18. 14.7.4 Example data-flow exchange For data flow-exchange examples refer to Section 17, Ref. 1, Annex K: • K.1 Overview of the data-flow exchange • K.2 Tag memory contents and lock-field values • K.3 Data-flow exchange and command sequence 14.8 Mandatory Select Commands Select commands select a particular UCODE G2X tag population based on user-defined criteria. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 26 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.8.1 Select For a detailed description of the mandatory Select command refer to Section 17, Ref. 1, section 6.3.2.10. 14.9 Mandatory Inventory Commands Inventory commands are used to run the collision arbitration protocol. 14.9.1 Query For a detailed description of the mandatory Query command refer to Section 17, Ref. 1, section 6.3.2.10. 14.9.2 QueryAdjust For a detailed description of the mandatory QueryAdjust command refer to Section 17, Ref. 1, section 6.3.2.10. 14.9.3 QueryRep For a detailed description of the mandatory QueryRep command refer to Section 17, Ref. 1, section 6.3.2.10. 14.9.4 ACK For a detailed description of the mandatory ACK command refer to Section 17, Ref. 1, section 6.3.2.10. 14.9.5 NAK For a detailed description of the mandatory NAK command refer to Section 17, Ref. 1, section 6.3.2.10. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 27 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.10 Mandatory Access Commands Access commands are used to read or write data from or to the G2X memory. For a detailed description of the mandatory Access command refer to Section 17, Ref. 1, section 6.3.2.10. 14.10.1 REQ_RN Access commands are used to read or write data from or to the G2X memory. For a detailed description of the mandatory Access command refer to Section 17, Ref. 1, section 6.3.2.10. 14.10.2 READ For a detailed description of the mandatory Req_RN command refer to Section 17, Ref. 1, section 6.3.2.10. 14.10.3 WRITE For a detailed description of the mandatory Write command refer to Section 17, Ref. 1, section 6.3.2.10. 14.10.4 KILL For a detailed description of the mandatory Kill command refer to Section 17, Ref. 1, section 6.3.2.10. 14.10.5 LOCK For a detailed description of the mandatory Lock command refer to Section 17, Ref. 1, section 6.3.2.10. 14.11 Optional Access Command 14.11.1 Access For a detailed description of the optional Access command refer to Section 17, Ref. 1, section 6.3.2.10. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 28 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.12 Custom Commands 14.12.1 ReadProtect The G2X ReadProtect custom command enables reliable read protection of the entire G2X memory. Executing ReadProtect from the Secured state will set the ReadProtect-bit to '1'. With the ReadProtect-Bit set the G2X will continue to work unaffected but fail its content. Following commands will be disabled: Read, Write, Kill, Lock, Access, ReadProtect, ChangeEAS, EAS Alarm and Calibrate. The G2X will only react upon an anticollision with Select, Query, QueryRep, QueryAdjust, ACK (no truncated reply), NAK, ReqRN but reply with zeros as EPC and CRC-16 content (except PC/password). ACK will return zeros except for the PC. The read protection can be removed by executing Reset ReadProtect. The ReadProtect-Bit will than be cleared. Devices whose access password is zero will ignore the command. A frame-sync must be prepended the command. After sending the ReadProtect command an interrogator shall transmit CW for the lesser of TReply or 20 ms, where TReply is the time between the interrogator's ReadProtect command and the backscattered reply. An interrogator may observe three possible responses after sending a ReadProtect, depending on the success or failure of the operation: • ReadProtect succeeds: After completing the ReadProtect the G2X shall backscatter the reply shown in Table 14 comprising a header (a 0-bit), the tag's handle, and a CRC-16 calculated over the 0-bit and handle. Immediately after this reply the G2X will render itself to this ReadProtect mode. If the interrogator observes this reply within 20 ms then the ReadProtect completed successfully. • The G2X encounters an error: The G2X will backscatter an error code during the CW period rather than the reply shown in the EPCglobal Spec (see Annex I for error-code definitions and for the reply format). • ReadProtect does not succeed: If the interrogator does not observe a reply within 20 ms then the ReadProtect did not complete successfully. The interrogator may issue a Req_RN command (containing the handle) to verify that the G2X is still in the interrogation zone, and may re-initiate the ReadProtect command. The G2X reply to the ReadProtect command will use the extended preamble shown in EPCglobal Spec (Figure 6.11 or Figure 6.15), as appropriate (i.e. a Tag shall reply as if TRext=1) regardless of the TRext value in the Query that initiated the round. Table 12. ReadProtect command Command RN CRC-16 # of bits 16 16 16 description 11100000 00000001 handle - 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 29 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL Table 13. G2X reply to a successful ReadProtect procedure Header RN CRC-16 # of bits 1 16 16 description 0 handle - Table 14. ReadProtect command-response table Starting State Condition Response Next State ready all – ready arbitrate, reply, acknowledged all – arbitrate open all - open secured valid handle & invalid access password – arbitrate valid handle & valid non zero access password Backscatter handle, when done secured invalid handle – secured killed all – killed 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 30 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.12.2 Reset ReadProtect Reset ReadProtect allows an interrogator to resets the ReadProtect-bit and re-enables reading of the G2X memory content according the EPCglobal specification. The G2X will execute Reset ReadProtect from the Open or Secured states. If a G2X in the Open or Secured states receives a Reset ReadProtect with a valid CRC-16 and a valid handle but an incorrect access password, it will not reply and transit to the Arbitrate state. If a G2X in the Open or Secured states receives a Reset ReadProtect with a valid CRC-16 and a valid handle but the ReadProtect-Bit is not set ('0'), it will not change the ReadProtect-Bit but backscatter the reply shown in Table 17. If a G2X in the Open or Secured receives a Reset ReadProtect with a valid CRC-16 but an invalid handle, or it receives a Reset ReadProtect before which the immediately preceding command was not a Req_RN, it will ignore the Reset ReadProtect and remain in its current state. A frame-sync must be prepended the Reset ReadProtect command. After sending a Reset ReadProtect an interrogator shall transmit CW for the lesser of TReply or 20 ms, where TReply is the time between the interrogator's Reset ReadProtect command and the G2X backscattered reply. An interrogator may observe three possible responses after sending a Reset ReadProtect, depending on the success or failure of the operation: • Write succeeds: After completing the Reset ReadProtect a G2X will backscatter the reply shown in Table 17 comprising a header (a 0-bit), the handle, and a CRC-16 calculated over the 0-bit and handle. If the interrogator observes this reply within 20 ms then the Reset ReadProtect completed successfully. • The G2X encounters an error: The G2X will backscatter an error code during the CW period rather than the reply shown in Table 17 (see EPCglobal Spec for error-code definitions and for the reply format). • Write does not succeed: If the interrogator does not observe a reply within 20 ms then the Reset ReadProtect did not complete successfully. The interrogator may issue a Req_RN command (containing the handle) to verify that the G2X is still in the interrogation zone, and may reissue the Reset ReadProtect command. The G2X reply to the Reset ReadProtect command will use the extended preamble shown in EPCglobal Spec (Figure 6.11 or Figure 6.15), as appropriate (i.e. a G2X will reply as if TRext=1 regardless of the TRext value in the Query that initiated the round. The Reset ReadProtect command is structured as following: • 16 bit command • Password: 32 bit Access-Password XOR with 2 times current RN16 • 16 bit handle • CRC-16 calculate over the first command-code bit to the last handle bit 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 31 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL Table 15. Reset ReadProtect command Command Password RN CRC-16 # of bits 16 32 16 16 description 11100000 00000010 (access password) 2*RN16 handle - Table 16. G2X reply to a successful Reset ReadProtect command Header RN CRC-16 # of bits 1 16 16 description 0 handle - Table 17. Reset ReadProtect command-response table Starting State Condition Response Next State ready all – ready arbitrate, reply, acknowledged all – arbitrate open ReadProtect bit is set, valid handle & valid access password Backscatter handle, when done open ReadProtect bit is set, valid handle & invalid access password – arbitrate ReadProtect bit is set, invalid handle – open ReadProtect bit is reset – open secured ReadProtect bit is set, valid handle & valid access password Backscatter handle, when done secured ReadProtect bit is set, valid handle & invalid access password – arbitrate ReadProtect bit is set, invalid handle – secured ReadProtect bit is reset – secured killed all – killed 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 32 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.12.3 ChangeEAS A G2X equipped RFID tag can be enhanced by a stand-alone operating EAS alarm feature. With an EAS-Alarm bit set to '1' the tag will reply to an EAS_Alarm command by backscattering a 64 bit alarm code without the need of a Select or Query. The EAS is a built-in solution so no connection to a backend database is required. As it is a custom command no Select or Query is required to detect the EAS state enabling fast, reliable and offline article surveillance. ChangeEAS can be executed from the Secured state only. The command will be ignored if the Access Password is zero, the command will also be ignored with an invalid CRC-16 or an invalid handle, the G2X will than remain in the current state. The CRC-16 is calculated from the first command-code bit to the last handle bit. A frame-sync must be prepended the command. The G2X reply to a successful ChangeEAS will use the extended preamble, as appropriate (i.e. a Tag shall reply as if TRext=1) regardless of the TRext value in the Query that initiated the round. After sending a ChangeEAS an interrogator shall transmit CW for less than TReply or 20 ms, where TReply is the time between the interrogator's ChangeEAS command and the G2X backscattered reply. An interrogator may observe three possible responses after sending a ChangeEAS, depending on the success or failure of the operation • Write succeeds: After completing the ChangeEAS a G2X will backscatter the reply shown in Table 20 comprising a header (a 0-bit), the handle, and a CRC-16 calculated over the 0-bit and handle. If the interrogator observes this reply within 20 ms then the ChangeEAS completed successfully. • The G2X encounters an error: The G2X will backscatter an error code during the CW period rather than the reply shown in Table 20 (see EPCglobal Spec for error-code definitions and for the reply format). • Write does not succeed: If the interrogator does not observe a reply within 20 ms then the ChangeEAS did not complete successfully. The interrogator may issue a Req_RN command (containing the handle) to verify that the G2X is still in the interrogator's field, and may reissue the ChangeEAS command. Upon receiving a valid ChangeEAS command a G2X will perform the commanded set/reset operation of the EAS_Alarm-Bit. If EAS-Bit is set, the EAS_Alarm command will be available after the next power up and reply the 64 bit EAS code upon execution. Otherwise the EAS_Alarm command will be ignored. Table 18. ChangeEAS command Command ChangeEas RN CRC-16 # of bits 16 1 16 16 description 11100000 00000011 1 ... set EAS system bit 0 ... reset EAS system bit handle 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 33 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL Table 19. G2X reply to a successful ChangeEAS command Header RN CRC-16 # of bits 1 16 16 description 0 handle - Table 20. ChangeEAS command-response table Starting State Condition Response Next State ready all – ready arbitrate, reply, acknowledged all – arbitrate open all – open secured valid handle Backscatter handle, when done secured invalid handle – secured killed all – killed Starting State Condition Response Next State 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 34 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.12.4 EAS_Alarm EAS_Alarm is a custom command causing the G2X to immediately backscatter an EAS-Alarmcode, when EAS ALARM bit is set without any delay caused by Select, Query and without the need for a backend database. The EAS feature of the G2X is available after enabling it by sending a ChangeEAS command described in Section 14.12.3 “ChangeEAS”. With an EAS-Alarm bit set to '1' the G2X will reply to an EAS_Alarm command by backscattering a fixed 64 bit alarm code. A G2X will reply to an EAS_Alarm command from the ready state only. If the EAS-Alarm bit is reset ('0') by sending a ChangeEAS command in the password protected Secure state the G2X will not reply to an EAS_Alarm command. The EAS_Alarm command is structured as following: • 16 bit command • 16 bit inverted command • DR (TRcal divide ratio) sets the T=>R link frequency as described in EPCglobal Spec. 6.3.1.2.8 and Table 6.9. • M (cycles per symbol) sets the T=>R data rate and modulation format as shown in EPCglobal Spec. Table 6.10. • TRext chooses whether the T=>R preamble is prepended with a pilot tone as described in EPCglobal Spec. 6.3.1.3. A preamble must be prepended the EAS_Alarm command according EPCglobal Spec, 6.3.1.2.8. Upon receiving an EAS_Alarm command the tag loads the CRC5 register with 01001b and backscatters the 64 bit alarm code accordingly. The reader is now able to calculate the CRC5 over the backscattered 64 bits received to verify the received code. Table 21. EAS_Alarm command Command Inv_Command DR M TRext CRC-16 # of bits 16 16 1 2 1 16 description 11100000 00000100 00011111 11111011 0: DR=8 1: DR=64/3 00: M=1 01: M=2 10: M=4 11: M=8 0: No pilot tone 1: Use pilot tone - Table 22. G2X reply to a successful EAS_Alarm command Header EAS Code # of bits 1 64 description 0 CRC5 (MSB) 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 35 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL Table 23. Eas_Alarm command-response table Starting State Condition Response Next State ready EAS-bit is set and non-zero access password Backscatter Alarm code ready arbitrate, reply, acknowledged EAS-bit is set and non-zero access password – arbitrate open EAS-bit is set and non-zero access password open secured EAS-bit is set and non-zero access password secured killed EAS-bit is set and non-zero access password – killed 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 36 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 14.12.5 Calibrate After execution of the custom Calibrate command the G2X will continuously backscatter the user memory content in an infinite loop. The G2XL will continuously backscatter zeros. This command can be used for frequency spectrum measurements. Calibrate can only be executed from the Secure state with an non-zero Access Password set otherwise the command will be ignored. The Calibrate command includes a CRC-16 calculated over the whole command, the handle and a prepended frame-sync. [1] G2XM [2] G2XL Table 24. Calibrate command Command RN16 CRC-16 # of bits 16 16 16 description 11100000 00000101 handle - Table 25. G2X reply to a successful Calibrate command Header Infinite repeat # of bits 1 512 (looped) description 0 User memory data[1] zeros[2] Table 26. Calibrate command-response table Starting State Condition Response Next State ready all – ready arbitrate, reply, acknowledged all – arbitrate secured nonzero access password Backscatter infinite _ access password is zero – secured killed all – killed 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 37 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 15. Support information 15.1 CRC Calculation EXAMPLE Old RN = 3D5Bh Table 27. Practical example of CRC calculation for a 'Req_RN' command by the reader CRC Calculated @ Reader Cmd Code for Req_RN F F F F 1 F F F E 1 F F F C 0 E F D 9 0 C F 9 3 0 8 F 0 7 0 0 E 2 F 0 1 C 5 E 1 2 8 9 9 First Byte of RN 0 5 1 3 A 0 A 2 7 4 1 4 4 E 8 1 9 9 F 1 1 3 3 E 2 1 7 7 E 5 0 E F C A 1 D F 9 4 Second Byte of RN 0 A F 0 9 1 5 E 1 2 0 B C 2 4 1 7 8 4 8 1 E 0 B 1 0 D 1 4 3 1 A 2 8 6 1 4 5 0 C -> ones complement: B A F 3 => Command-Sequence: C1 3D 5B BA F3 hex 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 38 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL Table 28. Practical example of CRC calculation for a 'Req_RN' command by the reader CRC Calculated @ Tag Cmd Code for Req_RN F F F F 1 F F F E 1 F F F C 0 E F D 9 0 C F 9 3 0 8 F 0 7 0 0 E 2 F 0 1 C 5 E 1 2 8 9 9 First Byte of RN 0 5 1 3 A 0 A 2 7 4 1 4 4 E 8 1 9 9 F 1 1 3 3 E 2 1 7 7 E 5 0 E F C A 1 D F 9 4 Second Byte of RN 0 A F 0 9 1 5 E 1 2 0 B C 2 4 1 7 8 4 8 1 E 0 B 1 0 D 1 4 3 1 A 2 8 6 1 4 5 0 C First Byte of CRC 1 9 A 3 9 0 2 4 5 3 1 5 8 8 7 1 A 1 2 F 1 4 2 5 E 0 8 4 B C 1 0 9 7 8 0 1 2 F 0 Second Byte of CRC 1 3 5 C 1 1 7 B A 3 1 E 7 6 7 1 C E C E 0 8 D B D 0 0 B 5 B 1 0 6 9 7 1 1 D 0 F -> Residue OK 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 39 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 16. Abbreviations Table 29. Abbreviations Acronym Description CRC Cyclic redundancy check CW Continuos wave EEPROM Electrically Erasable Programmable Read Only Memory EPC Electronic Product Code (containing Header, Domain Manager, Object Class and Serial Number) FM0 Bi phase space modulation G2 Generation 2 HBM Human Body Model IC Integrated Circuit LSB Least Significant Byte/Bit MSB Most Significant Byte/Bit NRZ Non-Return to Zero coding RF Radio Frequency RTF Reader Talks First Tari Type A Reference Interval (ISO 18000-6) UHF Ultra High Frequency Xxb Value in binary notation xxhex Value in hexadecimal notation 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 40 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 17. References [1] EPCglobal: EPC Radio-Frequency Identity Protocols Class-1 Generation-2 UHF RFID Protocol for Communications at 860 MHz – 960 MHz, Version 1.1.0 (December 17, 2005) [2] EPCglobal: EPC Tag Data Standards [3] EPCglobal (2004): FMCG RFID Physical Requirements Document (draft) [4] EPCglobal (2004): Class-1 Generation-2 UHF RFID Implementation Reference (draft) [5] European Telecommunications Standards Institute (ETSI), EN 302 208: Electromagnetic compatibility and radio spectrum matters (ERM) – Radio-frequency identification equipment operating in the band 865 MHz to 868 MHz with power levels up to 2 W, Part 1 – Technical characteristics and test methods [6] European Telecommunications Standards Institute (ETSI), EN 302 208: Electromagnetic compatibility and radio spectrum matters (ERM) – Radio-frequency identification equipment operating in the band 865 MHz to 868 MHz with power levels up to 2 W, Part 2 – Harmonized EN under article 3.2 of the R&TTE directive [7] [CEPT1]: CEPT REC 70-03 Annex 1 [8] [ETSI1]: ETSI EN 330 220-1, 2 [9] [ETSI3]: ETSI EN 302 208-1, 2 V<1.1.1> (2004-09-Electromagnetic compatibility And Radio spectrum Matters (ERM) Radio Frequency Identification Equipment operating in the band 865 - MHz to 868 MHz with power levels up to 2 W Part 1: Technical characteristics and test methods. [10] [FCC1]: FCC 47 Part 15 Section 247 [11] ISO/IEC Directives, Part 2: Rules for the structure and drafting of International Standards [12] ISO/IEC 3309: Information technology – Telecommunications and information exchange between systems – High-level data link control (HDLC) procedures – Frame structure [13] ISO/IEC 15961: Information technology, Automatic identification and data capture – Radio frequency identification (RFID) for item management – Data protocol: application interface [14] ISO/IEC 15962: Information technology, Automatic identification and data capture techniques – Radio frequency identification (RFID) for item management – Data protocol: data encoding rules and logical memory functions [15] ISO/IEC 15963: Information technology — Radio frequency identification for item management — Unique identification for RF tags [16] ISO/IEC 18000-1: Information technology — Radio frequency identification for item management — Part 1: Reference architecture and definition of parameters to be standardized [17] ISO/IEC 18000-6: Information technology automatic identification and data capture techniques — Radio frequency identification for item management air interface — Part 6: Parameters for air interface communications at 860–960 MHz [18] ISO/IEC 19762: Information technology AIDC techniques – Harmonized vocabulary – Part 3: radio-frequency identification (RFID) 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 41 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL [19] U.S. Code of Federal Regulations (CFR), Title 47, Chapter I, Part 15: Radio-frequency devices, U.S. Federal Communications Commission. [20] Data sheet - Delivery type description – General specification for 8” wafer on UV-tape with electronic fail die marking, BL-ID document number: 1093**1 1. ** ... document version number 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 42 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 18. Revision history Table 30. Revision history Document ID Release date Data sheet status Change notice Supersedes SL3ICS1002_1202 v.3.8 20131111 Product data sheet - SL3ICS1002_1202 v.3.7 Modifications: • Update of the delivery form (TSSOP package due to DOD removed) SL3ICS1002_1202 v.3.7 20121009 Product data sheet - 139036 Modifications: • Update of the delivery form 139036 20110310 Product data sheet 139035 Modifications: • Table 4 “TSSOP8 Marking”: added • Section 14.1.1.2 “Special behavior of user memory address 1Fh”: added 139035 20091102 Product data sheet 139034 Modifications: • Type SOT1122 added • Figure 2 “Wafer layout and pinning information”: correction of drawing 139034 20090721 Product data sheet 139033 Modifications: • Table 11 “TSSOP8 characteristics” andTable 7 “Package interface characteristics” :removed “Memory characteristics” 139033 20090605 Product data sheet - 139032 139132 Modifications: • This data sheet is a combination of data sheets SL3ICS1002 and SL3ICS1202 • New type FCS2 Aluminum, SOT1040AB2 added • Section 8.1.6 “Fail die identification”: added • Section 11 “Packing information”: edited 139032 20080716 Product data sheet 139031 Modifications: • rephrasing of Section 2 “Features and benefits” on page 2 • added “calibrate command” in Section 2 “Features and benefits” on page 2 • redesign of Figure 1 “Block diagram of G2X IC” on page 4 • merging of Fig. 2 Pinning and Fig. 3 Wafer layout - see Figure 2 “Wafer layout and pinning information” on page 5 • added type “FCS2 Polymer Strap - SOT1040AA1” in Section 4 “Ordering information”, Section 6 “Wafer layout and pinning information”, Section 7 “Package outline”, Section 8 “Mechanical specification”, Section 9 “Limiting values”, Section 10 “Characteristics” • added Section 11 “Handling information for Flip Chip Strap (FCS2, SOT1040)” on page 19 • added Section 11 “Packing information” on page 12 • added Table 8 “Symbol description” on page 14 • correction of Table 11 “Memory map” on page 22 • removed “ongoing” in 32 bit ongoing in Section 2.1 and Table 10 “G2X memory sections” 139031 20080428 Product data sheet 139030 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 43 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL Modifications: • update of Table 1 “Ordering information” on page 3 • added Section 7 “Package outline” on page 6 • added Section 8.1.7 “Map file distribution” on page 9 • added Table 9 “Limiting values TSSOP8 [1][2]” on page 14 • added room temperature in Table 11 “Memory characteristics” on page 15 • added Section 10.2 “TSSOP8 characteristics” on page 17 • update of the “EPCglobal compliance and interoperability certification” in Section 12.4 “Air interface standards” on page 15 • correction of “(excluding 16 bit CRC-16 and 16 bit PC) in Table 10 “G2X memory sections” on page 21 • correction of Initials in “tag mask designer” in Table 11 “Memory map” on page 22 • removed the sentence “The ChangeEAS custom command will toggle the state of the EAS-Alarm bit located in the EEprom” in Section 14.12.3 “ChangeEAS” on page 32. • added description of ChangeEAS in Table 18 “ChangeEAS command” on page 32 139030 20071221 Product data sheet - 139011 Modifications: • change of product status • general update 139011 20070910 Objective data sheet - 139010 Modifications: • removed double section Change EAS, EAS Alarm, Chapter 12.11.7 • changed “Reader” to “Tag” 139010 20070612 Objective data sheet - - • initial version Table 30. Revision history …continued Document ID Release date Data sheet status Change notice Supersedes 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 44 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 19. Legal information 19.1 Data sheet status [1] Please consult the most recently issued document before initiating or completing a design. [2] The term ‘short data sheet’ is explained in section “Definitions”. [3] The product status of device(s) described in this document may have changed since this document was published and may differ in case of multiple devices. The latest product status information is available on the Internet at URL http://www.nxp.com. 19.2 Definitions Draft — The document is a draft version only. The content is still under internal review and subject to formal approval, which may result in modifications or additions. NXP Semiconductors does not give any representations or warranties as to the accuracy or completeness of information included herein and shall have no liability for the consequences of use of such information. Short data sheet — A short data sheet is an extract from a full data sheet with the same product type number(s) and title. A short data sheet is intended for quick reference only and should not be relied upon to contain detailed and full information. For detailed and full information see the relevant full data sheet, which is available on request via the local NXP Semiconductors sales office. In case of any inconsistency or conflict with the short data sheet, the full data sheet shall prevail. Product specification — The information and data provided in a Product data sheet shall define the specification of the product as agreed between NXP Semiconductors and its customer, unless NXP Semiconductors and customer have explicitly agreed otherwise in writing. In no event however, shall an agreement be valid in which the NXP Semiconductors product is deemed to offer functions and qualities beyond those described in the Product data sheet. 19.3 Disclaimers Limited warranty and liability — Information in this document is believed to be accurate and reliable. However, NXP Semiconductors does not give any representations or warranties, expressed or implied, as to the accuracy or completeness of such information and shall have no liability for the consequences of use of such information. NXP Semiconductors takes no responsibility for the content in this document if provided by an information source outside of NXP Semiconductors. In no event shall NXP Semiconductors be liable for any indirect, incidental, punitive, special or consequential damages (including - without limitation - lost profits, lost savings, business interruption, costs related to the removal or replacement of any products or rework charges) whether or not such damages are based on tort (including negligence), warranty, breach of contract or any other legal theory. Notwithstanding any damages that customer might incur for any reason whatsoever, NXP Semiconductors’ aggregate and cumulative liability towards customer for the products described herein shall be limited in accordance with the Terms and conditions of commercial sale of NXP Semiconductors. Right to make changes — NXP Semiconductors reserves the right to make changes to information published in this document, including without limitation specifications and product descriptions, at any time and without notice. This document supersedes and replaces all information supplied prior to the publication hereof. Suitability for use — NXP Semiconductors products are not designed, authorized or warranted to be suitable for use in life support, life-critical or safety-critical systems or equipment, nor in applications where failure or malfunction of an NXP Semiconductors product can reasonably be expected to result in personal injury, death or severe property or environmental damage. NXP Semiconductors and its suppliers accept no liability for inclusion and/or use of NXP Semiconductors products in such equipment or applications and therefore such inclusion and/or use is at the customer’s own risk. Applications — Applications that are described herein for any of these products are for illustrative purposes only. NXP Semiconductors makes no representation or warranty that such applications will be suitable for the specified use without further testing or modification. Customers are responsible for the design and operation of their applications and products using NXP Semiconductors products, and NXP Semiconductors accepts no liability for any assistance with applications or customer product design. It is customer’s sole responsibility to determine whether the NXP Semiconductors product is suitable and fit for the customer’s applications and products planned, as well as for the planned application and use of customer’s third party customer(s). Customers should provide appropriate design and operating safeguards to minimize the risks associated with their applications and products. NXP Semiconductors does not accept any liability related to any default, damage, costs or problem which is based on any weakness or default in the customer’s applications or products, or the application or use by customer’s third party customer(s). Customer is responsible for doing all necessary testing for the customer’s applications and products using NXP Semiconductors products in order to avoid a default of the applications and the products or of the application or use by customer’s third party customer(s). NXP does not accept any liability in this respect. Limiting values — Stress above one or more limiting values (as defined in the Absolute Maximum Ratings System of IEC 60134) will cause permanent damage to the device. Limiting values are stress ratings only and (proper) operation of the device at these or any other conditions above those given in the Recommended operating conditions section (if present) or the Characteristics sections of this document is not warranted. Constant or repeated exposure to limiting values will permanently and irreversibly affect the quality and reliability of the device. Terms and conditions of commercial sale — NXP Semiconductors products are sold subject to the general terms and conditions of commercial sale, as published at http://www.nxp.com/profile/terms, unless otherwise agreed in a valid written individual agreement. In case an individual agreement is concluded only the terms and conditions of the respective agreement shall apply. NXP Semiconductors hereby expressly objects to applying the customer’s general terms and conditions with regard to the purchase of NXP Semiconductors products by customer. No offer to sell or license — Nothing in this document may be interpreted or construed as an offer to sell products that is open for acceptance or the grant, conveyance or implication of any license under any copyrights, patents or other industrial or intellectual property rights. Document status[1][2] Product status[3] Definition Objective [short] data sheet Development This document contains data from the objective specification for product development. Preliminary [short] data sheet Qualification This document contains data from the preliminary specification. Product [short] data sheet Production This document contains the product specification. 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 45 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL Export control — This document as well as the item(s) described herein may be subject to export control regulations. Export might require a prior authorization from competent authorities. Quick reference data — The Quick reference data is an extract of the product data given in the Limiting values and Characteristics sections of this document, and as such is not complete, exhaustive or legally binding. Non-automotive qualified products — Unless this data sheet expressly states that this specific NXP Semiconductors product is automotive qualified, the product is not suitable for automotive use. It is neither qualified nor tested in accordance with automotive testing or application requirements. NXP Semiconductors accepts no liability for inclusion and/or use of non-automotive qualified products in automotive equipment or applications. In the event that customer uses the product for design-in and use in automotive applications to automotive specifications and standards, customer (a) shall use the product without NXP Semiconductors’ warranty of the product for such automotive applications, use and specifications, and (b) whenever customer uses the product for automotive applications beyond NXP Semiconductors’ specifications such use shall be solely at customer’s own risk, and (c) customer fully indemnifies NXP Semiconductors for any liability, damages or failed product claims resulting from customer design and use of the product for automotive applications beyond NXP Semiconductors’ standard warranty and NXP Semiconductors’ product specifications. Translations — A non-English (translated) version of a document is for reference only. The English version shall prevail in case of any discrepancy between the translated and English versions. 19.4 Trademarks Notice: All referenced brands, product names, service names and trademarks are the property of their respective owners. UCODE — is a trademark of NXP B.V. 20. Contact information For more information, please visit: http://www.nxp.com For sales office addresses, please send an email to: salesaddresses@nxp.com 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 46 of 48 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 21. Tables Table 1. Ordering information G2XM . . . . . . . . . . . . . . . .3 Table 2. Ordering information G2XL. . . . . . . . . . . . . . . . .3 Table 3. Pin description of SOT1122 . . . . . . . . . . . . . . . .7 Table 4. SOT1122 Marking. . . . . . . . . . . . . . . . . . . . . . . .7 Table 5. Limiting values[1][2] . . . . . . . . . . . . . . . . . . . . . .10 Table 6. Wafer characteristics . . . . . . . . . . . . . . . . . . . . 11 Table 7. Package interface characteristics. . . . . . . . . . . 11 Table 8. Symbol description . . . . . . . . . . . . . . . . . . . . . .14 Table 9. Operating distances for UCODE G2X based tags and labels in released frequency bands . .14 Table 10. G2X memory sections . . . . . . . . . . . . . . . . . . .21 Table 11. Memory map. . . . . . . . . . . . . . . . . . . . . . . . . . .22 Table 12. ReadProtect command. . . . . . . . . . . . . . . . . . .28 Table 13. G2X reply to a successful ReadProtect procedure . . . . . . . . . . . . . . . . . . . . . . . . . . . . .29 Table 14. ReadProtect command-response table . . . . . .29 Table 15. Reset ReadProtect command . . . . . . . . . . . . .31 Table 16. G2X reply to a successful Reset ReadProtect command . . . . . . . . . . . . . . . . . . . . . . . . . . . . .31 Table 17. Reset ReadProtect command-response table 31 Table 18. ChangeEAS command . . . . . . . . . . . . . . . . . . 32 Table 19. G2X reply to a successful ChangeEAS command . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 Table 20. ChangeEAS command-response table . . . . . . 33 Table 21. EAS_Alarm command . . . . . . . . . . . . . . . . . . . 34 Table 22. G2X reply to a successful EAS_Alarm command . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 Table 23. Eas_Alarm command-response table. . . . . . . 35 Table 24. Calibrate command . . . . . . . . . . . . . . . . . . . . . 36 Table 25. G2X reply to a successful Calibrate command 36 Table 26. Calibrate command-response table . . . . . . . . . 36 Table 27. Practical example of CRC calculation for a 'Req_RN' command by the reader . . . . . . . . . 37 Table 28. Practical example of CRC calculation for a 'Req_RN' command by the reader. . . . . . . . . . 38 Table 29. Abbreviations . . . . . . . . . . . . . . . . . . . . . . . . . 39 Table 30. Revision history . . . . . . . . . . . . . . . . . . . . . . . . 42 22. Figures Fig 1. Block diagram of G2X IC . . . . . . . . . . . . . . . . . . . .4 Fig 2. Wafer layout and pinning information . . . . . . . . . .5 Fig 3. Package outline SOT1122 . . . . . . . . . . . . . . . . . . .6 Fig 4. G2X TID memory structure . . . . . . . . . . . . . . . . .21 139037 All information provided in this document is subject to legal disclaimers. © NXP B.V. 2013. All rights reserved. Product data sheet COMPANY PUBLIC Rev. 3.8 — 11 November 2013 139038 47 of 48 continued >> NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL 23. Contents 1 General description . . . . . . . . . . . . . . . . . . . . . . 1 2 Features and benefits . . . . . . . . . . . . . . . . . . . . 2 2.1 Key features . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 2.2 Key benefits . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 2.3 Custom commands. . . . . . . . . . . . . . . . . . . . . . 2 3 Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 4 Ordering information. . . . . . . . . . . . . . . . . . . . . 3 5 Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . 4 6 Wafer layout and pinning information . . . . . . . 5 6.1 Wafer layout . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 7 Package outline . . . . . . . . . . . . . . . . . . . . . . . . . 6 8 Mechanical specification . . . . . . . . . . . . . . . . . 8 8.1 Wafer specification . . . . . . . . . . . . . . . . . . . . . . 8 8.1.1 Wafer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 8.1.2 Wafer backside . . . . . . . . . . . . . . . . . . . . . . . . . 8 8.1.3 Chip dimensions . . . . . . . . . . . . . . . . . . . . . . . . 8 8.1.4 Passivation on front . . . . . . . . . . . . . . . . . . . . . 8 8.1.5 Au bump . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 8.1.6 Fail die identification . . . . . . . . . . . . . . . . . . . . 9 8.1.7 Map file distribution. . . . . . . . . . . . . . . . . . . . . . 9 9 Limiting values. . . . . . . . . . . . . . . . . . . . . . . . . 10 10 Characteristics. . . . . . . . . . . . . . . . . . . . . . . . . 11 10.1 Wafer characteristics . . . . . . . . . . . . . . . . . . . 11 10.2 Package characteristics . . . . . . . . . . . . . . . . . 11 11 Packing information . . . . . . . . . . . . . . . . . . . . 12 11.1 Wafer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 11.2 SOT1122 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 12 Functional description . . . . . . . . . . . . . . . . . . 13 12.1 Power transfer . . . . . . . . . . . . . . . . . . . . . . . . 13 12.2 Data transfer . . . . . . . . . . . . . . . . . . . . . . . . . . 13 12.2.1 Reader to G2X Link . . . . . . . . . . . . . . . . . . . . 13 12.2.2 G2X to reader Link . . . . . . . . . . . . . . . . . . . . . 13 12.3 Operating distances . . . . . . . . . . . . . . . . . . . . 14 12.4 Air interface standards . . . . . . . . . . . . . . . . . . 15 13 Physical layer and signaling. . . . . . . . . . . . . . 16 13.1 Reader to G2X communication . . . . . . . . . . . 16 13.1.1 Physical layer . . . . . . . . . . . . . . . . . . . . . . . . . 16 13.1.2 Modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 13.1.3 Data encoding. . . . . . . . . . . . . . . . . . . . . . . . . 16 13.1.4 Data rates . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 13.1.5 RF envelope for R=>T . . . . . . . . . . . . . . . . . . 16 13.1.6 Interrogator power-up/down waveform. . . . . . 16 13.1.7 Preamble and frame-sync . . . . . . . . . . . . . . . 16 13.2 G2X to reader communication . . . . . . . . . . . . 17 13.2.1 Modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 13.2.2 Data encoding . . . . . . . . . . . . . . . . . . . . . . . . 17 13.2.2.1 FM0 baseband . . . . . . . . . . . . . . . . . . . . . . . . 17 13.2.2.2 FM0 Preamble . . . . . . . . . . . . . . . . . . . . . . . . 17 13.2.2.3 Miller-modulated sub carrier . . . . . . . . . . . . . 17 13.2.2.4 Miller sub carrier preamble . . . . . . . . . . . . . . 17 13.2.3 Data rates . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 13.3 Link timing . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 13.3.1 Regeneration time . . . . . . . . . . . . . . . . . . . . . 18 13.3.2 Start-up time. . . . . . . . . . . . . . . . . . . . . . . . . . 18 13.3.3 Persistence time . . . . . . . . . . . . . . . . . . . . . . 18 13.4 Bit and byte ordering . . . . . . . . . . . . . . . . . . . 18 13.5 Data integrity . . . . . . . . . . . . . . . . . . . . . . . . . 19 13.6 CRC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 14 TAG selection, inventory and access . . . . . . 20 14.1 G2X Memory . . . . . . . . . . . . . . . . . . . . . . . . . 21 14.1.1 Memory map . . . . . . . . . . . . . . . . . . . . . . . . . 22 14.1.1.1 User memory (only G2XM) . . . . . . . . . . . . . . 23 14.1.1.2 Special behavior of user memory address 1Fh 23 14.1.1.3 Supported EPC types . . . . . . . . . . . . . . . . . . 23 14.2 Sessions, selected and inventoried flags. . . . 24 14.2.1 G2X States and slot counter . . . . . . . . . . . . . 24 14.2.2 G2X State Diagram . . . . . . . . . . . . . . . . . . . . 24 14.3 Managing tag populations . . . . . . . . . . . . . . . 24 14.4 Selecting tag populations. . . . . . . . . . . . . . . . 24 14.5 Inventorying tag populations . . . . . . . . . . . . . 24 14.6 Accessing individual tags. . . . . . . . . . . . . . . . 24 14.7 Interrogator commands and tag replies . . . . . 24 14.7.1 Commands. . . . . . . . . . . . . . . . . . . . . . . . . . . 24 14.7.2 State transition tables. . . . . . . . . . . . . . . . . . . 25 14.7.3 Command response tables . . . . . . . . . . . . . . 25 14.7.4 Example data-flow exchange. . . . . . . . . . . . . 25 14.8 Mandatory Select Commands . . . . . . . . . . . . 25 14.8.1 Select . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 14.9 Mandatory Inventory Commands. . . . . . . . . . 26 14.9.1 Query . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 14.9.2 QueryAdjust . . . . . . . . . . . . . . . . . . . . . . . . . . 26 14.9.3 QueryRep. . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 14.9.4 ACK . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 14.9.5 NAK . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 14.10 Mandatory Access Commands . . . . . . . . . . . 27 14.10.1 REQ_RN . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 14.10.2 READ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 14.10.3 WRITE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 14.10.4 KILL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 14.10.5 LOCK . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 14.11 Optional Access Command . . . . . . . . . . . . . . 27 14.11.1 Access . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 14.12 Custom Commands . . . . . . . . . . . . . . . . . . . . 28 NXP Semiconductors SL3ICS1002/1202 UCODE G2XM and G2XL © NXP B.V. 2013. All rights reserved. For more information, please visit: http://www.nxp.com For sales office addresses, please send an email to: salesaddresses@nxp.com Date of release: 11 November 2013 139038 Please be aware that important notices concerning this document and the product(s) described herein, have been included in section ‘Legal information’. 14.12.1 ReadProtect . . . . . . . . . . . . . . . . . . . . . . . . . . 28 14.12.2 Reset ReadProtect . . . . . . . . . . . . . . . . . . . . . 30 14.12.3 ChangeEAS . . . . . . . . . . . . . . . . . . . . . . . . . . 32 14.12.4 EAS_Alarm . . . . . . . . . . . . . . . . . . . . . . . . . . 34 14.12.5 Calibrate . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 15 Support information . . . . . . . . . . . . . . . . . . . . 37 15.1 CRC Calculation EXAMPLE . . . . . . . . . . . . . . 37 16 Abbreviations. . . . . . . . . . . . . . . . . . . . . . . . . . 39 17 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40 18 Revision history. . . . . . . . . . . . . . . . . . . . . . . . 42 19 Legal information. . . . . . . . . . . . . . . . . . . . . . . 44 19.1 Data sheet status . . . . . . . . . . . . . . . . . . . . . . 44 19.2 Definitions. . . . . . . . . . . . . . . . . . . . . . . . . . . . 44 19.3 Disclaimers . . . . . . . . . . . . . . . . . . . . . . . . . . . 44 19.4 Trademarks. . . . . . . . . . . . . . . . . . . . . . . . . . . 45 20 Contact information. . . . . . . . . . . . . . . . . . . . . 45 21 Tables . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 22 Figures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 23 Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47 1. General description The LPC11U3x are an ARM Cortex-M0 based, low-cost 32-bit MCU family, designed for 8/16-bit microcontroller applications, offering performance, low power, simple instruction set and memory addressing together with reduced code size compared to existing 8/16-bit architectures. The LPC11U3x operate at CPU frequencies of up to 50 MHz. Equipped with a highly flexible and configurable full-speed USB 2.0 device controller, the LPC11U3x brings unparalleled design flexibility and seamless integration to today’s demanding connectivity solutions. The peripheral complement of the LPC11U3x includes up to 128 kB of flash memory, up to 12 kB of SRAM data memory and 4 kB EEPROM, one Fast-mode Plus I2C-bus interface, one RS-485/EIA-485 USART with support for synchronous mode and smart card interface, two SSP interfaces, four general purpose counter/timers, a 10-bit ADC, and up to 54 general purpose I/O pins. The I/O Handler is a software library-supported hardware engine that can be used to add performance, connectivity and flexibility to system designs. It is available on the LPC11U37HFBD64/401. The I/O Handler can emulate serial interfaces such as UART, I2C, and I2S with no or very low additional CPU load and can off-load the CPU by performing processing-intensive functions like DMA transfers in hardware. Software libraries for multiple I/O Handler applications are available on http://www.LPCware.com. For additional documentation related to the LPC11U3x parts, see Section 15 “References”. 2. Features and benefits System: ARM Cortex-M0 processor, running at frequencies of up to 50 MHz. ARM Cortex-M0 built-in Nested Vectored Interrupt Controller (NVIC). Non-Maskable Interrupt (NMI) input selectable from several input sources. System tick timer. Memory: Up to 128 kB on-chip flash program memory with sector (4 kB) and page erase (256 byte) access. 4 kB on-chip EEPROM data memory; byte erasable and byte programmable; on-chip API support. Up to 12 kB SRAM data memory. LPC11U3x 32-bit ARM Cortex-M0 microcontroller; up to 128 kB flash; up to 12 kB SRAM and 4 kB EEPROM; USB device; USART Rev. 2.2 — 11 March 2014 Product data sheet LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 2 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller 16 kB boot ROM. In-System Programming (ISP) and In-Application Programming (IAP) via on-chip bootloader software. ROM-based USB drivers. Flash updates via USB supported. ROM-based 32-bit integer division routines. Debug options: Standard JTAG (Joint Test Action Group) test interface for BSDL (Boundary Scan Description Language). Serial Wire Debug. Digital peripherals: Up to 54 General Purpose I/O (GPIO) pins with configurable pull-up/pull-down resistors, repeater mode, and open-drain mode. Up to 8 GPIO pins can be selected as edge and level sensitive interrupt sources. Two GPIO grouped interrupt modules enable an interrupt based on a programmable pattern of input states of a group of GPIO pins. High-current source output driver (20 mA) on one pin. High-current sink driver (20 mA) on true open-drain pins. Four general purpose counter/timers with a total of up to 8 capture inputs and 13 match outputs. Programmable Windowed WatchDog Timer (WWDT) with a dedicated, internal low-power WatchDog Oscillator (WDO). Analog peripherals: 10-bit ADC with input multiplexing among eight pins. Serial interfaces: USB 2.0 full-speed device controller. USART with fractional baud rate generation, internal FIFO, a full modem control handshake interface, and support for RS-485/9-bit mode and synchronous mode. USART supports an asynchronous smart card interface (ISO 7816-3). Two SSP controllers with FIFO and multi-protocol capabilities. I2C-bus interface supporting the full I2C-bus specification and Fast-mode Plus with a data rate of up to 1 Mbit/s with multiple address recognition and monitor mode. I/O Handler for hardware emulation of serial interfaces and DMA; supported through software libraries. (LPC11U37HFBD64/401 only.) Clock generation: Crystal Oscillator with an operating range of 1 MHz to 25 MHz (system oscillator). 12 MHz high-frequency Internal RC oscillator (IRC) that can optionally be used as a system clock. Internal low-power, low-frequency WatchDog Oscillator (WDO) with programmable frequency output. PLL allows CPU operation up to the maximum CPU rate with the system oscillator or the IRC as clock sources. A second, dedicated PLL is provided for USB. Clock output function with divider that can reflect the crystal oscillator, the main clock, the IRC, or the watchdog oscillator. Power control: LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 3 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller Integrated PMU (Power Management Unit) to minimize power consumption during Sleep, Deep-sleep, Power-down, and Deep power-down modes. Power profiles residing in boot ROM provide optimized performance and minimized power consumption for any given application through one simple function call. Four reduced power modes: Sleep, Deep-sleep, Power-down, and Deep power-down. Processor wake-up from Deep-sleep and Power-down modes via reset, selectable GPIO pins, watchdog interrupt, or USB port activity. Processor wake-up from Deep power-down mode using one special function pin. Power-On Reset (POR). Brownout detect with up to four separate thresholds for interrupt and forced reset. Unique device serial number for identification. Single 3.3 V power supply (1.8 V to 3.6 V). Temperature range 40 C to +85 C. Available as LQFP64, LQFP48, TFBGA48, and HVQFN33 packages. 3. Applications 4. Ordering information Consumer peripherals Handheld scanners Medical USB audio devices Industrial control Table 1. Ordering information Type number Package Name Description Version LPC11U34FHN33/311 HVQFN33 plastic thermal enhanced very thin quad flat package; no leads; 33 terminals; body 7 7 0.85 mm n/a LPC11U34FBD48/311 LQFP48 plastic low profile quad flat package; 48 leads; body 7 7 1.4 mm SOT313-2 LPC11U34FHN33/421 HVQFN33 plastic thermal enhanced very thin quad flat package; no leads; 33 terminals; body 7 7 0.85 mm n/a LPC11U34FBD48/421 LQFP48 plastic low profile quad flat package; 48 leads; body 7 7 1.4 mm SOT313-2 LPC11U35FHN33/401 HVQFN33 plastic thermal enhanced very thin quad flat package; no leads; 33 terminals; body 7 7 0.85 mm n/a LPC11U35FBD48/401 LQFP48 plastic low profile quad flat package; 48 leads; body 7 7 1.4 mm SOT313-2 LPC11U35FBD64/401 LQFP64 plastic low profile quad flat package; 64 leads; body 10 10 1.4 mm SOT314-2 LPC11U35FHI33/501 HVQFN33 plastic thermal enhanced very thin quad flat package; no leads; 33 terminals; body 5 5 0.85 mm n/a LPC11U35FET48/501 TFBGA48 plastic thin fine-pitch ball grid array package; 48 balls; body 4.5 4.5 0.7 mm SOT1155-2 LPC11U36FBD48/401 LQFP48 plastic low profile quad flat package; 48 leads; body 7 7 1.4 mm SOT313-2 LPC11U36FBD64/401 LQFP64 plastic low profile quad flat package; 64 leads; body 10 10 1.4 mm SOT314-2 LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 4 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller 4.1 Ordering options [1] For general-purpose use. [2] For I/O Handler use only. LPC11U37FBD48/401 LQFP48 plastic low profile quad flat package; 48 leads; body 7 7 1.4 mm SOT313-2 LPC11U37HFBD64/401 LQFP64 plastic low profile quad flat package; 64 leads; body 10 10 1.4 mm SOT314-2 LPC11U37FBD64/501 LQFP64 plastic low profile quad flat package; 64 leads; body 10 10 1.4 mm SOT314-2 Table 1. Ordering information …continued Type number Package Name Description Version Table 2. Ordering options Type number Flash in kB EEPROM in kB SRAM0 in kB USB SRAM in kB SRAM1 in kB Total SRAM in kB[1] I/O Handler USART I2C-bus FM+ SSP USB device ADC channels GPIO pins LPC11U34FHN33/311 40 4 8 - - 8 no 1 1 2 1 8 26 LPC11U34FBD48/311 40 4 8 - - 8 no 1 1 2 1 8 40 LPC11U34FHN33/421 48 4 8 2 - 10 no 1 1 2 1 8 26 LPC11U34FBD48/421 48 4 8 2 - 10 no 1 1 2 1 8 40 LPC11U35FHN33/401 64 4 8 2 - 10 no 1 1 2 1 8 26 LPC11U35FBD48/401 64 4 8 2 - 10 no 1 1 2 1 8 40 LPC11U35FBD64/401 64 4 8 2 - 10 no 1 1 2 1 8 54 LPC11U35FHI33/501 64 4 8 2 2[1] 12 no 1 1 2 1 8 26 LPC11U35FET48/501 64 4 8 2 2[1] 12 no 1 1 2 1 8 40 LPC11U36FBD48/401 96 4 8 2 - 10 no 1 1 2 1 8 40 LPC11U36FBD64/401 96 4 8 2 - 10 no 1 1 2 1 8 54 LPC11U37FBD48/401 128 4 8 2 - 10 no 1 1 2 1 8 40 LPC11U37HFBD64/401 128 4 8 2 2[2] 10 yes 1 1 2 1 8 54 LPC11U37FBD64/501 128 4 8 2 2[1] 12 no 1 1 2 1 8 54 LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 5 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller 5. Block diagram (1) Not available on HVQFN33 packages. (2) CT16B0_CAP1, CT16B1_CAP1 available on LQFP64 packages only; CT32B0_CAP1 available on TFBGA48, LQFP48, and LQFP64 packages only; CT32B1_CAP1 available in TFBGA48/LQFP64 packages only. (3) LPC11U37HFBD64/401 only. Fig 1. Block diagram SRAM 8/10/12 kB ARM CORTEX-M0 TEST/DEBUG INTERFACE FLASH 40/48/64/96/128 kB HIGH-SPEED GPIO AHB TO APB BRIDGE CLOCK GENERATION, POWER CONTROL, SYSTEM FUNCTIONS RESET SWD, JTAG LPC11U3x slave slave master slave slave ROM 16 kB slave AHB-LITE BUS GPIO ports 0/1 I/O IOH_[20:0] HANDLER(3) CLKOUT IRC, WDO SYSTEM OSCILLATOR POR PLL0 USB PLL BOD 10-bit ADC USART/ SMARTCARD INTERFACE AD[7:0] RXD TXD CTS, RTS, DTR SCLK GPIO INTERRUPTS 32-bit COUNTER/TIMER 0 CT32B0_MAT[3:0] CT32B0_CAP[1:0](2) 32-bit COUNTER/TIMER 1 CT32B1_MAT[3:0] CT32B1_CAP[1:0](2) DCD, DSR(1), RI(1) 16-bit COUNTER/TIMER 1 WINDOWED WATCHDOG TIMER GPIO GROUP0 INTERRUPTS CT16B1_MAT[1:0] 16-bit COUNTER/TIMER 0 CT16B0_MAT[2:0] CT16B0_CAP[1:0](2) CT16B1_CAP[1:0](2) GPIO pins GPIO pins GPIO pins GPIO GROUP1 INTERRUPTS system bus SSP0 SCK0, SSEL0, MISO0, MOSI0 SSP1 SCK1, SSEL1, MISO1, MOSI1 I2C-BUS IOCON SYSTEM CONTROL PMU SCL, SDA XTALIN XTALOUT USB DEVICE CONTROLLER USB_DP USB_DM USB_VBUS USB_FTOGGLE, USB_CONNECT 002aag345 master slave EEPROM 4 kB LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 6 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller 6. Pinning information 6.1 Pinning For parts LPC11U34FHN33/311, LPC11U34FHN33/421, LPC11U35FHN33/401, LPC11U35FHI33/501 Fig 2. Pin configuration (HVQFN33) 002aag809 Transparent top view PIO0_8/MISO0/CT16B0_MAT0 PIO0_20/CT16B1_CAP0 PIO0_2/SSEL0/CT16B0_CAP0 PIO0_9/MOSI0/CT16B0_MAT1 VDD SWCLK/PIO0_10/SCK0/CT16B0_MAT2 XTALOUT PIO0_22/AD6/CT16B1_MAT1/MISO1 XTALIN TDI/PIO0_11/AD0/CT32B0_MAT3 PIO0_1/CLKOUT/CT32B0_MAT2/USB_FTOGGLE TMS/PIO0_12/AD1/CT32B1_CAP0 RESET/PIO0_0 TDO/PIO0_13/AD2/CT32B1_MAT0 PIO1_19/DTR/SSEL1 TRST/PIO0_14/AD3/CT32B1_MAT1 PIO0_3/USB_VBUS PIO0_4/SCL PIO0_5/SDA PIO0_21/CT16B1_MAT0/MOSI1 USB_DM USB_DP PIO0_6/USB_CONNECT/SCK0 PIO0_7/CTS PIO0_19/TXD/CT32B0_MAT1 PIO0_18/RXD/CT32B0_MAT0 PIO0_17/RTS/CT32B0_CAP0/SCLK VDD PIO1_15/DCD/CT16B0_MAT2/SCK1 PIO0_23/AD7 PIO0_16/AD5/CT32B1_MAT3/WAKEUP SWDIO/PIO0_15/AD4/CT32B1_MAT2 8 17 7 18 6 19 5 20 4 21 3 22 2 23 1 24 9 10 11 12 13 14 15 16 32 31 30 29 28 27 26 25 terminal 1 index area 33 VSS LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 7 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller Fig 3. Pin configuration (TFBGA48) 002aag810 LPC11U35FET48/501 Transparent top view H G F D B E C A 1 2 3 4 5 6 7 8 ball A1 index area LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 8 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller Fig 4. Pin configuration (LQFP48) LPC11U34FBD48/311 LPC11U34FBD48/421 LPC11U35FBD48/401 LPC11U36FBD48/401 LPC11U37FBD48/401 PIO1_25/CT32B0_MAT1 PIO1_13/DTR/CT16B0_MAT0/TXD PIO1_19/DTR/SSEL1 TRST/PIO0_14/AD3/CT32B1_MAT1 RESET/PIO0_0 TDO/PIO0_13/AD2/CT32B1_MAT0 PIO0_1/CLKOUT/CT32B0_MAT2/USB_FTOGGLE TMS/PIO0_12/AD1/CT32B1_CAP0 VSS TDI/PIO0_11/AD0/CT32B0_MAT3 XTALIN PIO1_29/SCK0/CT32B0_CAP1 XTALOUT PIO0_22/AD6/CT16B1_MAT1/MISO1 VDD SWCLK/PIO0_10/SCK0/CT16B0_MAT2 PIO0_20/CT16B1_CAP0 PIO0_9/MOSI0/CT16B0_MAT1 PIO0_2/SSEL0/CT16B0_CAP0 PIO0_8/MISO0/CT16B0_MAT0 PIO1_26/CT32B0_MAT2/RXD PIO1_21/DCD/MISO1 PIO1_27/CT32B0_MAT3/TXD PIO1_31 PIO1_20/DSR/SCK1 PIO1_16/RI/CT16B0_CAP0 PIO0_3/USB_VBUS PIO0_19/TXD/CT32B0_MAT1 PIO0_4/SCL PIO0_18/RXD/CT32B0_MAT0 PIO0_5/SDA PIO0_17/RTS/CT32B0_CAP0/SCLK PIO0_21/CT16B1_MAT0/MOSI1 VDD PIO1_23/CT16B1_MAT1/SSEL1 PIO1_15/DCD/CT16B0_MAT2/SCK1 USB_DM PIO0_23/AD7 USB_DP VSS PIO1_24/CT32B0_MAT0 PIO0_16/AD5/CT32B1_MAT3/WAKEUP PIO0_6/USB_CONNECT/SCK0 SWDIO/PIO0_15/AD4/CT32B1_MAT2 PIO0_7/CTS PIO1_28/CT32B0_CAP0/SCLK PIO1_22/RI/MOSI1 PIO1_14/DSR/CT16B0_MAT1/RXD 002aag811 1 2 3 4 5 6 7 8 9 10 11 12 36 35 34 33 32 31 30 29 28 27 26 25 13 14 15 16 17 18 19 20 21 22 23 48 47 46 45 44 43 42 41 40 39 38 24 37 LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 9 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller See Table 3 for the full pin name. Fig 5. Pin configuration (LQFP64) LPC11U35FBD64/401 LPC11U36FBD64/401 LPC11U37HFBD64/401 LPC11U37FBD64/501 PIO1_0 VDD PIO1_25 PIO1_13 PIO1_19 TRST/PIO0_14 RESET/PIO0_0 TDO/PIO0_13 PIO0_1 TMS/PIO0_12 PIO1_7 PIO1_11 VSS TDI/PIO0_11 XTALIN PIO1_29 XTALOUT PIO0_22 VDD PIO1_8 PIO0_20 SWCLK/PIO0_10 PIO1_10 PIO0_9 PIO0_2 PIO0_8 PIO1_26 PIO1_21 PIO1_27 PIO1_2 PIO1_4 VDD PIO1_1 PIO1_6 PIO1_20 PIO1_16 PIO0_3 PIO0_19 PIO0_4 PIO0_18 PIO0_5 PIO0_17 PIO0_21 PIO1_12 PIO1_17 VDD PIO1_23 PIO1_15 USB_DM PIO0_23 USB_DP PIO1_9 PIO1_24 VSS PIO1_18 PIO0_16 PIO0_6 SWDIO/PIO0_15 PIO0_7 PIO1_22 PIO1_28 PIO1_3 PIO1_5 PIO1_14 002aag812 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 48 47 46 45 44 43 42 41 40 39 38 37 36 35 34 33 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 64 63 62 61 60 59 58 57 56 55 54 53 52 51 50 49 LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 10 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller 6.2 Pin description Table 3 shows all pins and their assigned digital or analog functions in order of the GPIO port number. The default function after reset is listed first. All port pins have internal pull-up resistors enabled after reset except for the true open-drain pins PIO0_4 and PIO0_5. Every port pin has a corresponding IOCON register for programming the digital or analog function, the pull-up/pull-down configuration, the repeater, and the open-drain modes. The USART, counter/timer, and SSP functions are available on more than one port pin. Table 3. Pin description Symbol Pin HVQFN33 Pin TFBGA48 Pin LQFP48 Pin LQFP64 Reset state [1] Type Description RESET/PIO0_0 2 C1 3 4 [2] I; PU I RESET — External reset input with 20 ns glitch filter. A LOW-going pulse as short as 50 ns on this pin resets the device, causing I/O ports and peripherals to take on their default states and processor execution to begin at address 0. This pin also serves as the debug select input. LOW level selects the JTAG boundary scan. HIGH level selects the ARM SWD debug mode. In deep power-down mode, this pin must be pulled HIGH externally. The RESET pin can be left unconnected or be used as a GPIO pin if an external RESET function is not needed and Deep power-down mode is not used. - I/O PIO0_0 — General purpose digital input/output pin. PIO0_1/CLKOUT/ CT32B0_MAT2/ USB_FTOGGLE 3 C2 4 5 [3] I; PU I/O PIO0_1 — General purpose digital input/output pin. A LOW level on this pin during reset starts the ISP command handler or the USB device enumeration. - O CLKOUT — Clockout pin. - O CT32B0_MAT2 — Match output 2 for 32-bit timer 0. - O USB_FTOGGLE — USB 1 ms Start-of-Frame signal. PIO0_2/SSEL0/ CT16B0_CAP0/IOH_0 8 F1 10 13 [3] I; PU I/O PIO0_2 — General purpose digital input/output pin. - I/O SSEL0 — Slave select for SSP0. - I CT16B0_CAP0 — Capture input 0 for 16-bit timer 0. - I/O IOH_0 — I/O Handler input/output 0. LPC11U37HFBD64/401 only. PIO0_3/USB_VBUS/ IOH_1 9 H2 14 19 [3] I; PU I/O PIO0_3 — General purpose digital input/output pin. A LOW level on this pin during reset starts the ISP command handler. A HIGH level during reset starts the USB device enumeration. - I USB_VBUS — Monitors the presence of USB bus power. - I/O IOH_1 — I/O Handler input/output 1. LPC11U37HFBD64/401 only. LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 11 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller PIO0_4/SCL/IOH_2 10 G3 15 20 [4] I; IA I/O PIO0_4 — General purpose digital input/output pin (open-drain). - I/O SCL — I2C-bus clock input/output (open-drain). High-current sink only if I2C Fast-mode Plus is selected in the I/O configuration register. - I/O IOH_2 — I/O Handler input/output 2. LPC11U37HFBD64/401 only. PIO0_5/SDA/IOH_3 11 H3 16 21 [4] I; IA I/O PIO0_5 — General purpose digital input/output pin (open-drain). - I/O SDA — I2C-bus data input/output (open-drain). High-current sink only if I2C Fast-mode Plus is selected in the I/O configuration register. - I/O IOH_3 — I/O Handler input/output 3. LPC11U37HFBD64/401 only. PIO0_6/USB_CONNECT/ SCK0/IOH_4 15 H6 22 29 [3] I; PU I/O PIO0_6 — General purpose digital input/output pin. - O USB_CONNECT — Signal used to switch an external 1.5 k resistor under software control. Used with the SoftConnect USB feature. - I/O SCK0 — Serial clock for SSP0. - I/O IOH_4 — I/O Handler input/output 4. LPC11U37HFBD64/401 only. PIO0_7/CTS/IOH_5 16 G7 23 30 [5] I; PU I/O PIO0_7 — General purpose digital input/output pin (high-current output driver). - I CTS — Clear To Send input for USART. - I/O IOH_5 — I/O Handler input/output 5. (LPC11U37HFBD64/401 only.) PIO0_8/MISO0/ CT16B0_MAT0/R/IOH_6 17 F8 27 36 [3] I; PU I/O PIO0_8 — General purpose digital input/output pin. - I/O MISO0 — Master In Slave Out for SSP0. - O CT16B0_MAT0 — Match output 0 for 16-bit timer 0. - - Reserved. - I/O IOH_6 — I/O Handler input/output 6. (LPC11U37HFBD64/401 only.) PIO0_9/MOSI0/ CT16B0_MAT1/R/IOH_7 18 F7 28 37 [3] I; PU I/O PIO0_9 — General purpose digital input/output pin. - I/O MOSI0 — Master Out Slave In for SSP0. - O CT16B0_MAT1 — Match output 1 for 16-bit timer 0. - - Reserved. - I/O IOH_7 — I/O Handler input/output 7. (LPC11U37HFBD64/401 only.) Table 3. Pin description Symbol Pin HVQFN33 Pin TFBGA48 Pin LQFP48 Pin LQFP64 Reset state [1] Type Description LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 12 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller SWCLK/PIO0_10/SCK0/ CT16B0_MAT2 19 E7 29 38 [3] I; PU I SWCLK — Serial wire clock and test clock TCK for JTAG interface. - I/O PIO0_10 — General purpose digital input/output pin. - O SCK0 — Serial clock for SSP0. - O CT16B0_MAT2 — Match output 2 for 16-bit timer 0. TDI/PIO0_11/AD0/ CT32B0_MAT3 21 D8 32 42 [6] I; PU I TDI — Test Data In for JTAG interface. - I/O PIO0_11 — General purpose digital input/output pin. - I AD0 — A/D converter, input 0. - O CT32B0_MAT3 — Match output 3 for 32-bit timer 0. TMS/PIO0_12/AD1/ CT32B1_CAP0 22 C7 33 44 [6] I; PU I TMS — Test Mode Select for JTAG interface. - I/O PIO_12 — General purpose digital input/output pin. - I AD1 — A/D converter, input 1. - I CT32B1_CAP0 — Capture input 0 for 32-bit timer 1. TDO/PIO0_13/AD2/ CT32B1_MAT0 23 C8 34 45 [6] I; PU O TDO — Test Data Out for JTAG interface. - I/O PIO0_13 — General purpose digital input/output pin. - I AD2 — A/D converter, input 2. - O CT32B1_MAT0 — Match output 0 for 32-bit timer 1. TRST/PIO0_14/AD3/ CT32B1_MAT1 24 B7 35 46 [6] I; PU I TRST — Test Reset for JTAG interface. - I/O PIO0_14 — General purpose digital input/output pin. - I AD3 — A/D converter, input 3. - O CT32B1_MAT1 — Match output 1 for 32-bit timer 1. SWDIO/PIO0_15/AD4/ CT32B1_MAT2 25 B6 39 52 [6] I; PU I/O SWDIO — Serial wire debug input/output. - I/O PIO0_15 — General purpose digital input/output pin. - I AD4 — A/D converter, input 4. - O CT32B1_MAT2 — Match output 2 for 32-bit timer 1. PIO0_16/AD5/ CT32B1_MAT3/IOH_8/ WAKEUP 26 A6 40 53 [6] I; PU I/O PIO0_16 — General purpose digital input/output pin. - I AD5 — A/D converter, input 5. - O CT32B1_MAT3 — Match output 3 for 32-bit timer 1. - I/O IOH_8 — I/O Handler input/output 8. (LPC11U37HFBD64/401 only.) - I WAKEUP — Deep power-down mode wake-up pin with 20 ns glitch filter. Pull this pin HIGH externally before entering Deep power-down mode, then pull LOW to exit Deep power-down mode. A LOW-going pulse as short as 50 ns wakes up the part. Table 3. Pin description Symbol Pin HVQFN33 Pin TFBGA48 Pin LQFP48 Pin LQFP64 Reset state [1] Type Description LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 13 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller PIO0_17/RTS/ CT32B0_CAP0/SCLK 30 A3 45 60 [3] I; PU I/O PIO0_17 — General purpose digital input/output pin. - O RTS — Request To Send output for USART. - I CT32B0_CAP0 — Capture input 0 for 32-bit timer 0. - I/O SCLK — Serial clock input/output for USART in synchronous mode. PIO0_18/RXD/ CT32B0_MAT0 31 B3 46 61 [3] I; PU I/O PIO0_18 — General purpose digital input/output pin. - I RXD — Receiver input for USART. Used in UART ISP mode. - O CT32B0_MAT0 — Match output 0 for 32-bit timer 0. PIO0_19/TXD/ CT32B0_MAT1 32 B2 47 62 [3] I; PU I/O PIO0_19 — General purpose digital input/output pin. - O TXD — Transmitter output for USART. Used in UART ISP mode. - O CT32B0_MAT1 — Match output 1 for 32-bit timer 0. PIO0_20/CT16B1_CAP0 7 F2 9 11 [3] I; PU I/O PIO0_20 — General purpose digital input/output pin. - I CT16B1_CAP0 — Capture input 0 for 16-bit timer 1. PIO0_21/CT16B1_MAT0/ MOSI1 12 G4 17 22 [3] I; PU I/O PIO0_21 — General purpose digital input/output pin. - O CT16B1_MAT0 — Match output 0 for 16-bit timer 1. - I/O MOSI1 — Master Out Slave In for SSP1. PIO0_22/AD6/ CT16B1_MAT1/MISO1 20 E8 30 40 [6] I; PU I/O PIO0_22 — General purpose digital input/output pin. - I AD6 — A/D converter, input 6. - O CT16B1_MAT1 — Match output 1 for 16-bit timer 1. - I/O MISO1 — Master In Slave Out for SSP1. PIO0_23/AD7/IOH_9 27 A5 42 56 [6] I; PU I/O PIO0_23 — General purpose digital input/output pin. - I AD7 — A/D converter, input 7. - I/O IOH_9 — I/O Handler input/output 9. (LPC11U37HFBD64/401 only.) PIO1_0/CT32B1_MAT0/ IOH_10 - - - 1 [3] I; PU I/O PIO1_0 — General purpose digital input/output pin. - O CT32B1_MAT0 — Match output 0 for 32-bit timer 1. - I/O IOH_10 — I/O Handler input/output 10. (LPC11U37HFBD64/401 only.) PIO1_1/CT32B1_MAT1/ IOH_11 - - - 17 [3] I; PU I/O PIO1_1 — General purpose digital input/output pin. - O CT32B1_MAT1 — Match output 1 for 32-bit timer 1. - I/O IOH_11 — I/O Handler input/output 11. (LPC11U37HFBD64/401 only.) PIO1_2/CT32B1_MAT2/ IOH_12 - - - 34 [3] I; PU I/O PIO1_2 — General purpose digital input/output pin. - O CT32B1_MAT2 — Match output 2 for 32-bit timer 1. - I/O IOH_12 — I/O Handler input/output 12. (LPC11U37HFBD64/401 only.) Table 3. Pin description Symbol Pin HVQFN33 Pin TFBGA48 Pin LQFP48 Pin LQFP64 Reset state [1] Type Description LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 14 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller PIO1_3/CT32B1_MAT3/ IOH_13 - - - 50 [3] I; PU I/O PIO1_3 — General purpose digital input/output pin. - O CT32B1_MAT3 — Match output 3 for 32-bit timer 1. - I/O IOH_13 — I/O Handler input/output 13. (LPC11U37HFBD64/401 only.) PIO1_4/CT32B1_CAP0/ IOH_14 - - - 16 [3] I; PU I/O PIO1_4 — General purpose digital input/output pin. - I CT32B1_CAP0 — Capture input 0 for 32-bit timer 1. - I/O IOH_14 — I/O Handler input/output 14. (LPC11U37HFBD64/401 only.) PIO1_5/CT32B1_CAP1 /IOH_15 - H8 - 32 [3] I; PU I/O PIO1_5 — General purpose digital input/output pin. - I CT32B1_CAP1 — Capture input 1 for 32-bit timer 1. - I/O IOH_15 — I/O Handler input/output 15. (LPC11U37HFBD64/401 only.) PIO1_6/IOH_16 - - - 64 [3] I; PU I/O PIO1_6 — General purpose digital input/output pin. - I/O IOH_16 — I/O Handler input/output 16. (LPC11U37HFBD64/401 only.) PIO1_7/IOH_17 - - - 6 [3] I; PU I/O PIO1_7 — General purpose digital input/output pin. - I/O IOH_17 — I/O Handler input/output 17. (LPC11U37HFBD64/401 only.) PIO1_8/IOH_18 - - - 39 [3] I; PU I/O PIO1_8 — General purpose digital input/output pin. - I/O IOH_18 — I/O Handler input/output 18. (LPC11U37HFBD64/401 only.) PIO1_9 - - - 55 [3] I; PU I/O PIO1_9 — General purpose digital input/output pin. PIO1_10 - - - 12 [3] I; PU I/O PIO1_10 — General purpose digital input/output pin. PIO1_11 - - - 43 [3] I; PU I/O PIO1_11 — General purpose digital input/output pin. PIO1_12 - - - 59 [3] I; PU I/O PIO1_12 — General purpose digital input/output pin. PIO1_13/DTR/ CT16B0_MAT0/TXD - B8 36 47 [3] I; PU I/O PIO1_13 — General purpose digital input/output pin. - O DTR — Data Terminal Ready output for USART. - O CT16B0_MAT0 — Match output 0 for 16-bit timer 0. - O TXD — Transmitter output for USART. PIO1_14/DSR/ CT16B0_MAT1/RXD - A8 37 49 [3] I; PU I/O PIO1_14 — General purpose digital input/output pin. - I DSR — Data Set Ready input for USART. - O CT16B0_MAT1 — Match output 1 for 16-bit timer 0. - I RXD — Receiver input for USART. PIO1_15/DCD/ CT16B0_MAT2/SCK1 28 A4 43 57 [3] I; PU I/O PIO1_15 — General purpose digital input/output pin. I DCD — Data Carrier Detect input for USART. - O CT16B0_MAT2 — Match output 2 for 16-bit timer 0. - I/O SCK1 — Serial clock for SSP1. Table 3. Pin description Symbol Pin HVQFN33 Pin TFBGA48 Pin LQFP48 Pin LQFP64 Reset state [1] Type Description LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 15 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller PIO1_16/RI/ CT16B0_CAP0 - A2 48 63 [3] I; PU I/O PIO1_16 — General purpose digital input/output pin. - I RI — Ring Indicator input for USART. - I CT16B0_CAP0 — Capture input 0 for 16-bit timer 0. PIO1_17/CT16B0_CAP1/ RXD - - - 23 [3] I; PU I/O PIO1_17 — General purpose digital input/output pin. - I CT16B0_CAP1 — Capture input 1 for 16-bit timer 0. - I RXD — Receiver input for USART. PIO1_18/CT16B1_CAP1/ TXD - - - 28 [3] I; PU I/O PIO1_18 — General purpose digital input/output pin. - I CT16B1_CAP1 — Capture input 1 for 16-bit timer 1. - O TXD — Transmitter output for USART. PIO1_19/DTR/SSEL1 1 B1 2 3 [3] I; PU I/O PIO1_19 — General purpose digital input/output pin. - O DTR — Data Terminal Ready output for USART. - I/O SSEL1 — Slave select for SSP1. PIO1_20/DSR/SCK1 - H1 13 18 [3] I; PU I/O PIO1_20 — General purpose digital input/output pin. - I DSR — Data Set Ready input for USART. - I/O SCK1 — Serial clock for SSP1. PIO1_21/DCD/MISO1 - G8 26 35 [3] I; PU I/O PIO1_21 — General purpose digital input/output pin. - I DCD — Data Carrier Detect input for USART. - I/O MISO1 — Master In Slave Out for SSP1. PIO1_22/RI/MOSI1 - A7 38 51 [3] I; PU I/O PIO1_22 — General purpose digital input/output pin. - I RI — Ring Indicator input for USART. - I/O MOSI1 — Master Out Slave In for SSP1. PIO1_23/CT16B1_MAT1/ SSEL1 - H4 18 24 [3] I; PU I/O PIO1_23 — General purpose digital input/output pin. - O CT16B1_MAT1 — Match output 1 for 16-bit timer 1. - I/O SSEL1 — Slave select for SSP1. PIO1_24/CT32B0_MAT0 - G6 21 27 [3] I; PU I/O PIO1_24 — General purpose digital input/output pin. - O CT32B0_MAT0 — Match output 0 for 32-bit timer 0. PIO1_25/CT32B0_MAT1 - A1 1 2 [3] I; PU I/O PIO1_25 — General purpose digital input/output pin. - O CT32B0_MAT1 — Match output 1 for 32-bit timer 0. PIO1_26/CT32B0_MAT2/ RXD/IOH_19 - G2 11 14 [3] I; PU I/O PIO1_26 — General purpose digital input/output pin. - O CT32B0_MAT2 — Match output 2 for 32-bit timer 0. - I RXD — Receiver input for USART. - I/O IOH_19 — I/O Handler input/output 19. (LPC11U37HFBD64/401 only.) PIO1_27/CT32B0_MAT3/ TXD/IOH_20 - G1 12 15 [3] I; PU I/O PIO1_27 — General purpose digital input/output pin. - O CT32B0_MAT3 — Match output 3 for 32-bit timer 0. - O TXD — Transmitter output for USART. - I/O IOH_20 — I/O Handler input/output 20. (LPC11U37HFBD64/401 only.) Table 3. Pin description Symbol Pin HVQFN33 Pin TFBGA48 Pin LQFP48 Pin LQFP64 Reset state [1] Type Description LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 16 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller [1] Pin state at reset for default function: I = Input; O = Output; PU = internal pull-up enabled; IA = inactive, no pull-up/down enabled; F = floating; If the pins are not used, tie floating pins to ground or power to minimize power consumption. [2] 5 V tolerant pad. RESET functionality is not available in Deep power-down mode. Use the WAKEUP pin to reset the chip and wake up from Deep power-down mode. An external pull-up resistor is required on this pin for the Deep power-down mode. See Figure 32 for the reset pad configuration. [3] 5 V tolerant pad providing digital I/O functions with configurable pull-up/pull-down resistors and configurable hysteresis (see Figure 31). [4] I2C-bus pin compliant with the I2C-bus specification for I2C standard mode, I2C Fast-mode, and I2C Fast-mode Plus. The pin requires an external pull-up to provide output functionality. When power is switched off, this pin is floating and does not disturb the I2C lines. Open-drain configuration applies to all functions on this pin. [5] 5 V tolerant pad providing digital I/O functions with configurable pull-up/pull-down resistors and configurable hysteresis (see Figure 31); includes high-current output driver. [6] 5 V tolerant pad providing digital I/O functions with configurable pull-up/pull-down resistors, configurable hysteresis, and analog input. When configured as a ADC input, digital section of the pad is disabled and the pin is not 5 V tolerant (see Figure 31); includes digital input glitch filter. [7] Pad provides USB functions. It is designed in accordance with the USB specification, revision 2.0 (Full-speed and Low-speed mode only). This pad is not 5 V tolerant. [8] When the system oscillator is not used, connect XTALIN and XTALOUT as follows: XTALIN can be left floating or can be grounded (grounding is preferred to reduce susceptibility to noise). Leave XTALOUT floating. PIO1_28/CT32B0_CAP0/ SCLK - H7 24 31 [3] I; PU I/O PIO1_28 — General purpose digital input/output pin. - I CT32B0_CAP0 — Capture input 0 for 32-bit timer 0. - I/O SCLK — Serial clock input/output for USART in synchronous mode. PIO1_29/SCK0/ CT32B0_CAP1 - D7 31 41 [3] I; PU I/O PIO1_29 — General purpose digital input/output pin. - I/O SCK0 — Serial clock for SSP0. - I CT32B0_CAP1 — Capture input 1 for 32-bit timer 0. PIO1_31 - - 25 - [3] I; PU I/O PIO1_31 — General purpose digital input/output pin. USB_DM 13 G5 19 25 [7] F - USB_DM — USB bidirectional D line. USB_DP 14 H5 20 26 [7] F - USB_DP — USB bidirectional D+ line. XTALIN 4 D1 6 8 [8] - - Input to the oscillator circuit and internal clock generator circuits. Input voltage must not exceed 1.8 V. XTALOUT 5 E1 7 9 [8] - - Output from the oscillator amplifier. VDD 6; 29 B4; E2 8; 44 10; 33; 48; 58 - - Supply voltage to the internal regulator, the external rail, and the ADC. Also used as the ADC reference voltage. VSS 33 B5; D2 5; 41 7; 54 - - Ground. Table 3. Pin description Symbol Pin HVQFN33 Pin TFBGA48 Pin LQFP48 Pin LQFP64 Reset state [1] Type Description LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 17 of 77 NXP Semiconductors LPC11U3x 32-bit ARM Cortex-M0 microcontroller 7. Functional description 7.1 On-chip flash programming memory The LPC11U3x contain up to 128 kB on-chip flash program memory. The flash can be programmed using In-System Programming (ISP) or In-Application Programming (IAP) via the on-chip boot loader software. The flash memory is divided into 4 kB sectors with each sector consisting of 16 pages. Individual pages can be erased using the IAP erase page command. 7.2 EEPROM The LPC11U3x contain 4 kB of on-chip byte-erasable and byte-programmable EEPROM data memory. The EEPROM can be programmed using In-Application Programming (IAP) via the on-chip boot loader software. 7.3 SRAM The LPC11U3x contain a total of 8 kB, 10 kB, or 12 kB on-chip static RAM memory. On the LPC11U37HFBD64/401, the 2 kB SRAM1 region at location 0x2000 0000 to 0x2000 07FFF is used for the I/O Handler software library. Do not use this memory location for data or other user code. 7.4 On-chip ROM The on-chip ROM contains the boot loader and the following Application Programming Interfaces (APIs): • In-System Programming (ISP) and In-Application Programming (IAP) support for flash including IAP erase page command. • IAP support for EEPROM • USB API • Power profiles for configuring power consumption and PLL settings • 32-bit integer division routines 7.5 Memory map The LPC11U3x incorporates several distinct memory regions, shown in the following figures. Figure 6 shows the overall map of the entire address space from the user program viewpoint following reset. The interrupt vector area supports address remapping. The AHB (Advanced High-performance Bus) peripheral area is 2 MB in size and is divided to allow for up to 128 peripherals. The APB (Advanced Peripheral Bus) peripheral area is 512 kB in size and is divided to allow for up to 32 peripherals. Each peripheral of either type is allocated 16 kB of space. This addressing scheme allows simplifying the address decoding for each peripheral. LPC11U3X All information provided in this document is subject to legal disclaimers. © NXP Semiconductors N.V. 2014. All rights reserved. Product data sheet Rev. 2.2 — 11 March 2014 18 of 77 NXP Semiconductors LPC11U3x 32-b